Short-circuit protection of the power switch

Abstract

A control system for a power switch including a channel. The system includes a short-circuit protection circuit configured to change the control terminal of the power switch from one potential level to another potential level so that the charge carrier concentration in the channel is increased in response to detection of an overvoltage applied to the power switch by an overvoltage detection circuit while the control terminal of the power switch is at a second potential level.

Description

Technical Field

[0001] The present disclosure generally relates to semiconductor switches that switch power in applications including, for example, switching power converters and inverters.

Background Art

[0002] Electronic devices use power to operate. There are many application contexts where providing power to an electronic device requires relatively high voltage and / or current to be switched. If the power switch is not properly designed to withstand high voltages or to conduct large currents, the power switch can be damaged. Further, even a properly designed power switch can be damaged under inappropriate operating conditions including short circuits, electrostatic discharge events, power surges, lightning strikes, etc. Depending on the context, a failure of the power switch can not only cause a malfunction of the device including the power switch, but also cause a malfunction of other equipment.

[0003] The safe operating area (SOA) of a power switch is a definition of the current and voltage states in which the power switch can be assumed to operate without self-damage and degradation. In practice, manufacturers and suppliers present in the data sheet the safe operating area for different power switches for both the forward bias (i.e., while the power switch is on) state and the reverse bias (i.e., while the power switch is off or switching off) state.

[0004] In addition to safety, it is further desirable for power to be switched efficiently. The two main causes of power loss in a semiconductor switch are conduction loss and switching loss. Conduction loss occurs when the power switch is in the conducting state and due to the inherent forward voltage drop across the channel of the power switch. For example, the conduction loss can be reduced by reducing the duty cycle (i.e., the relative duration during which the power switch is in the conducting state) and / or by reducing the forward voltage drop across the channel of the power switch. Generally, the duty cycle is determined by the operating context, and reducing the duty cycle to reduce conduction loss is often not practical. For example, the forward voltage drop can be reduced by increasing the effective size of the channel (i.e., the dimensions of one channel) (or by conducting current in parallel through multiple channels), or by increasing the number of charge carriers in the channel. For example, through careful design of the power switch and / or by biasing the control terminal of the switch, the number of mobile charge carriers can be increased to increase the number of mobile charge carriers in the channel in the on state.

[0005] By way of example, many modern enhancement-mode IGBT power switches can be biased in the on state with a gate-to-emitter voltage of about +15 volts. In many applications, this is sufficient to draw enough carriers into the channel so that the conduction loss is acceptable. However, in other applications (e.g., applications that use a relatively low switching frequency), the conduction loss occupies an unacceptably large portion of the total losses. In these other applications, the gate-to-emitter voltage can be increased, for example, up to about +25 volts so that more charge carriers are drawn into the channel.

[0006] Switching losses occur when the power switch switches from an off state to an on state, and vice versa. During switching, both the voltage across the power switch and the current through the power switch transition between their respective steady-state values in the off and on states. The product of this voltage and current during the transition is the power lost due to switching. For example, by lowering the switching frequency (i.e., how often the switch transitions occur) and / or by switching more rapidly between the on and off states, the switching losses can be reduced. In many instances, the switching frequency is determined by the operating context, and it may not be practical to lower the switching frequency to reduce switching losses. By moving charge carriers more rapidly within and out of the switch channel, e.g., by increasing the attractive / repulsive forces that move charge carriers in / out of the channel, through careful design of the power switch and / or by biasing the control terminal of the switch, the duration of the switching can be shortened.

[0007] The desires for both safe and efficient switching are often mutually conflicting, and design trade - offs often have to be made. For example, increasing the number of charge carriers in a channel increases the likelihood that the current and voltage states in a power switch will move outside the safe operating range in the event of a short - circuit condition. There is necessarily a delay in detecting a short - circuit and switching the power switch off after detection. During this delay, the voltage across the power switch can rise and the current through the power switch can increase to a high level such that the transistor (e.g., in the case of a MOSFET) saturates or (e.g., in the case of an IGBT) desaturates. During (de)saturation, the current conducted by the power switch can become high enough to damage or fail the power switch. To prevent this, driver circuits for the gate or other control terminals of the power switch often include a (de)saturation protection function. For example, the voltage across the power switch can be measured to detect (de)saturation.

[0008] To reduce the delay in responding to a short - circuit condition, control and drive circuits including (de)saturation protection are generally directly coupled to the power switch. In some examples, particularly in applications where the power switch is an IGBT, such driver circuits can be implemented in an application - specific integrated circuit (ASIC) designed to drive a power switch with specific characteristics.

SUMMARY OF THE INVENTION

[0009] Like reference numerals in the various drawings indicate like elements.

BRIEF DESCRIPTION OF THE DRAWINGS

[0010]

Figure 1

Figure 2

Figure 3

Figure 4

DETAILED DESCRIPTION OF THE INVENTION

[0011] For purposes of explanation, the detailed description is presented from the perspective of an insulated gate bipolar transistor (IGBT) power switch. However, the corresponding teachings can be applied to various different enhancement or depletion type devices that involve electron or hole charge carriers (e.g., bipolar junction transistor (BJT), metal-oxide-semiconductor field-effect transistor (MOSFET), high-electron-mobility transistor (HEMT), etc.). The device can be realized in silicon, silicon carbide, gallium nitride, or other semiconductor materials.

[0012] FIG. 1 is a schematic diagram of various waveforms 100, 105, 110, 112 during response to detection of a short circuit during driving of an IGBT power switch. For example, the waveforms shown can occur when the IGBT power switch is biased by a gate-to-emitter voltage designed to draw a relatively large number of charge carriers into the IGBT channel, e.g., a gate-to-emitter voltage of about +25 volts in modern IGBT devices.

[0013] In particular, waveform 100 is a reference voltage V as a function of time Drepresents, and waveform 105 is the actual (internal) gate-to-emitter voltage V of the gate as a function of time GE represents, and waveform 110 is the collector current I as a function of time C represents, and waveform 112 is the collector-to-emitter voltage V as a function of time CE represents. The waveform 100 of the reference voltage V D represents the ideal voltage at the gate of the IGBT power switch, while the gate-to-emitter voltage V GE represents the actual voltage at the gate of the IGBT power switch. In a real device, the gate-to-emitter voltage V GE differs from the reference voltage V D . The exact nature of the difference depends on the nature of the circuit implementation. The collector current I C in waveform 110, and the collector-to-emitter voltage V CE in waveform 112 are the responses of the driven system to the gate-to-emitter voltage V GE in waveform 105.

[0014] The time scales of waveforms 100, 105, 110, and 112 are the same, and for a typical IGBT power switch, the time spread is generally between 2 microseconds and 10 microseconds. The voltage scales of waveforms 100 and 105 are approximately the same, and for a typical IGBT power switch, the voltage spread is generally between -15 volts and 25 volts. For a typical IGBT power switch, the current scale of waveform 110 is generally between 2 kiloamperes and 30 kiloamperes. For a typical IGBT power switch, the voltage scale of waveform 112 is generally between 600 volts and 6500 volts.

[0015] At the beginning of the period shown, the IGBT is in the conducting state. The reference voltage V D represented by waveform 100 and the gate-to-emitter voltage V GEBoth are substantially at their respective highest levels 115, 120. For the period shown, the gate and other capacitances (including gate-emitter capacitance, gate-collector capacitance, and parasitics) are substantially fully charged, and the reference voltage and the actual gate voltage do not substantially change. Further, the collector current I represented by waveform 110 C is at level 125, and the collector-to-emitter voltage V CE is at level 127. Generally, the magnitude of the current at level 125 is mainly determined by system regulation. The magnitude of the voltage at level 127 is mainly determined by the device characteristics of the IGBT, and the applied gate voltage and other parameters (such as temperature). In other words, the forward voltage drop of the IGBT is generally negligible compared to the voltage across the load. For the period shown, the power demand of the load is constant, and the collector current is in a steady state over time.

[0016] Levels 115, 120 are selected to make the voltage drop through the IGBT channel in the on state relatively small. In any case, when the IGBT is in the on state and the reference voltage V D and the actual gate-to-emitter voltage V GE are at levels lower than 115, 120, even if the voltage drop through the IGBT remains negligible compared to the voltage drop through the load, the IGBT control and drive circuit uses a gate voltage that further reduces the forward voltage drop of the channel and reduces conduction losses to drive the IGBT.

[0017] At time point T0, a short-circuit state occurs outside the power switch. The collector current I represented by waveform 110 Cbegins to increase rapidly. After the delay, in response to the detection of the short - circuit state, at time T1, the over - current protection function in the IGBT control and drive circuit is triggered. In FIG. 1, the magnitude of the delay is different in the period from T0 to T1. The short - circuit state can be detected by various different methods. For example, the rate of change of the voltage across the IGBT and / or the collector current can be compared with their respective threshold values to detect the short - circuit state.

[0018] Regardless of how the short - circuit state is detected, the collector current I C continues to increase during this delay. In some examples, the increase can be large enough such that the collector current I C increases to a level where the IGBT power switch cannot be turned off. In other words, directly turning off the IGBT power switch using such a large collector current I C will damage the IGBT.

[0019] Therefore, instead of directly turning off the IGBT power switch, the IGBT control and drive circuit initiates a two - step process to turn off the IGBT. In the first step, the IGBT control and drive circuit does not attempt to drive the IGBT using the voltage that switches it to the off state. Rather, the IGBT control and drive circuit first uses a gate - to - emitter voltage V GE level 135 to drive the IGBT in order to reduce the number - and thus the concentration - of charge carriers in the channel. Generally, the concertation of charge carriers is sufficient to keep the IGBT in the on state. In other words, the IGBT control and drive circuit drives the IGBT using a reference voltage V D level 130. By reducing the number of charge carriers in the channel, even for a desaturated IGBT, with a smaller collector current I C level, the IGBT is not damaged. In fact, many gate - driver control devices for IGBT power switches already use a gate - to - emitter voltage V GEincludes a sufficient desaturation protection function to protect the IGBT power switch when it is low. A low gate-to-emitter voltage V GE is used so that the IGBT power switch can be switched to the off state by a safe method.

[0020] In the context shown, voltage level 130 is lower than voltage level 115. In a typical IGBT device, voltage level 130 can be, for example, between 12 volts and 17 volts, while voltage level 115 can be, for example, between 20 volts and 30 volts. However, in some examples, voltage level 130 can be, for example, between 5 volts and 12 volts, resulting in a reduced charge carrier concentration without the power switch remaining in a fully on state. In the illustrated embodiment, the ideal reference voltage V D between levels 115 and 130 in waveform 100 is shown as an ideal step. In the actual gate-to-emitter voltage V GE of the real world, there are non-ideal situations. In particular, due to the gate and other capacitances, the actual gate-to-emitter voltage V GE represented by waveform 105 decreases more slowly than the reference voltage V D of waveform 100.

[0021] After the IGBT control and drive circuit starts driving the IGBT power switch using the reference voltage V D level 130, the collector current I C continues to increase over a period of time, and the IGBT power switch finally becomes desaturated. In the illustrated embodiment, at T2, desaturation of the IGBT power switch occurs and the collector-to-emitter voltage V CE rises rapidly as shown in waveform 112. In a further illustrated embodiment, the collector current I C reaches its maximum value at a point near T2.

[0022] As described above, assuming that the IGBT control and drive circuit continues to directly drive the IGBT to the off state, the increasing collector current I Crises to a level that may damage the IGBT. Instead, the IGBT control and drive circuit drives the IGBT power switch using a sufficient voltage level to reduce the number of charge carriers in the channel and can maintain the IGBT in the on state. Over time (in this example, during the period from T2 to T3), the collector current drops to a level from which the IGBT power switch is driven off. At time T3, the IGBT is desaturated but can be safely driven to the off state.

[0023] Note that the time when the gate-to-emitter voltage V GE reaches approximately level 135 is not necessarily the same as the time when the IGBT power switch can be safely driven to the off state. In other words, in other embodiments, the gate-to-emitter voltage V GE approaches level 135 before or after time T3.

[0024] Regarding the collector current I C at T1, even after the IGBT control and drive circuit indicates that the IGBT is driven using the reference voltage V D level 130, the collector current I C initially continues to increase rapidly. As the IGBT becomes desaturated, the collector-to-emitter voltage V CE rises. However, at some point, the collector current I C reaches a peak and, as the gate-to-emitter voltage V GE and the number of charge carriers in the channel decrease, the collector current I C begins to decrease further. The collector-to-emitter voltage V CE generally reaches a peak shortly after the collector current I C and then begins to decrease. At T3, the collector-to-emitter voltage V CE reaches a level from which the IGBT can be safely driven to the off state. In response, at T3, the IGBT control and drive circuit uses a reference voltage V DDrive the IGBT using level 145. In the waveform 100 shown, level 145 is negative. For example, level 145 can be between -3 volts and -20 volts.

[0025] Collector current I C decreases relatively rapidly during this transition and decreases to approximately zero when the IGBT reaches the off state. The collector-to-emitter voltage V CE rises during this transition and can overshoot the level 155 of the switched voltage, as shown. Nevertheless, this overshoot remains within the safe operating range of the IGBT power switch. Eventually, the collector-to-emitter voltage V CE stabilizes at the level 155 of the switched voltage.

[0026] Figure 2 is the state transition diagram 200 of the control and drive circuit for the power switch. For purposes of explanation, the state transition diagram 200 omits many aspects of the operation of the control and drive circuit. For example, the active clamp function generally continues to operate while the switch is transitioning to the off state (e.g., during transitions 229, 213 in the state transition diagram 200). As another example, multi-stage drive can be used to transition the switch to the on state (e.g., during transition 229 in the state transition diagram 200). As yet another different example, the start-up of the device is generally omitted from the state transition diagram 200. As yet another different example, the states shown may not be exclusive or may include more than one state. For example, there may be multiple reduced gate voltage states in which the control and drive circuit drives the IGBT power switch using different reduced reference gate voltages. As another example, there may be multiple active clamp states in which the gate-to-emitter voltage V GE is different. Further, there may be additional transitions between such "enhanced" and "reduced" clamped states. The state transition diagram 200 is thus to be interpreted as illustrating only a part of the operation of the control and drive circuit.

[0027] The state transition diagram 200 includes a switch-on state 205, a reduced gate voltage state 210, an active clamp state 215, a switch-off state / anomaly state 220, and a switch-off state / no anomaly state 225.

[0028] During normal operation, the power switch transitions along transition 229 between the switch-on state 205 and the switch-off state / no anomaly state 225 according to the operating context of the power switch. The switch-off state / no anomaly state 225 causes one or more reflexive transitions 227 that represent continuous monitoring of one or more states indicating that the IGBT power switch is to be transitioned to the switch-on state 205. The switch-on state 205 causes one or more reflexive transitions 228 that represent continuous monitoring of one or more states indicating that the IGBT power switch is to be transitioned to the switch-off state / no anomaly state 225. The states are diverse and can include, for example, a feedback signal reaching a certain level, a request signal from another circuit, a start command, or a resume command. For the sake of simplicity, despite these multiple possibilities, only one reflexive transition 227, 228 is shown for each state 205, 225.

[0029] Turning to short-circuit detection and protection, the switch-on state 205 causes reflexive transition 207 and state transition 209. The reflexive transition 207 represents continuous monitoring for detecting a short-circuit state external to the IGBT power switch. As described above, such a short-circuit can be detected, for example, based on a relatively high rate of change of the collector current through the IGBT power switch. The short-circuit detection monitoring can include comparing a parameter with a threshold level indicating a short-circuit. The state transition 209 is triggered by the detection of a short-circuit and causes the control and drive circuit to transition to the reduced gate voltage state 210. In the reduced gate voltage state 210, the control and drive circuit drives the IGBT power switch using a reduced reference gate voltage. In the context of the waveforms shown in FIG. 1, the state transition 209 corresponds to the transition of the reference voltage V D occurring at T1.

[0030] The reduced gate voltage state 210 causes reflective transitions 212 and two state transitions 213, 214. The reflective transition 212 monitors the voltage V applied to the IGBT power switch. CE The monitoring indicates continuous monitoring of the voltage V. CE The monitoring may include comparing the voltage V with a threshold level indicating a harmful overvoltage applied to the IGBT power switch, and comparing the voltage V with a threshold value indicating that the IGBT power switch can be switched off without leaving the safe operating range. As will be described in detail below, the V CE monitoring can be implemented using active or passive components. For example, the response of the control and drive circuit to the voltage V crossing the threshold can be triggered by a comparator or by a passive component such as a diode. The state transition 213 is triggered by the voltage V applied to the IGBT power switch indicating that the IGBT power switch is saturated, and transitions the control and drive circuit to the state 220 where the IGBT power switch is driven off. In the context of the waveforms shown in FIG. 1, the state transition 213 corresponds to the transition of the reference voltage V occurring at T3. In other embodiments, the collector current I CE can be monitored to detect when the IGBT power switch is transitioned to the off state. CE CE D C D C C The state transition 214 is triggered when the voltage V applied to the IGBT power switch rises to a threshold level indicating a harmful overvoltage applied to the IGBT power switch. The state transition 214 transitions the control and drive circuit to the active clamp state 215. Generally, an active clamp is configured to slow down the transition of the IGBT power switch to the off state by slowing down the depletion of charge carriers from the channel during the transition. The active clamp can be implemented in many different ways, but various approaches generally slow down the discharge of the gate of the IGBT power switch. Such a transition is shown below in FIG. 3.

[0031] CE CE ​

[0032] The active clamp state 215 causes a reflective transition 217 and a state transition 219. The reflective transition 217 indicates continuous monitoring of the voltage V applied to the IGBT power switch. The monitoring may include comparing the voltage V with a threshold level indicating that the voltage V has dropped to a level within the safe operating range of the IGBT power switch. The state transition 219 is triggered by the voltage V applied to the IGBT power switch dropping to this threshold level and returns the control and drive circuit to the reduced gate voltage state 210. CE of the voltage V applied to the IGBT power switch. The monitoring may include comparing the voltage V with a threshold level indicating that the voltage V has dropped to a level within the safe operating range of the IGBT power switch. The state transition 219 is triggered by the voltage V applied to the IGBT power switch dropping to this threshold level and returns the control and drive circuit to the reduced gate voltage state 210. CE with the voltage V CE applied to the IGBT power switch. The monitoring may include comparing the voltage V with a threshold level indicating that the voltage V has dropped to a level within the safe operating range of the IGBT power switch. The state transition 219 is triggered by the voltage V applied to the IGBT power switch dropping to this threshold level and returns the control and drive circuit to the reduced gate voltage state 210. CE applied to the IGBT power switch dropping to this threshold level and returns the control and drive circuit to the reduced gate voltage state 210.

[0033] In some embodiments, the function for triggering the active clamp of the IGBT control and drive circuit is effective in both the switch-on state 205 and the active clamp state 215. Thus, the same function already present in the IGBT control and drive circuit can be used in both states 205, 215. In some embodiments, the gate-to-emitter voltage V in the active clamp state 215 is the same as in the switch-on state 205. However, this is not necessarily the case, and the magnitude of the gate-to-emitter voltage V by which the IGBT control and drive circuit drives the power switch does not have to be the same as the gate-to-emitter voltage V in the switch-on state 205. For example, in some embodiments, the gate-to-emitter voltage V in the active clamp state 215 may be slightly lower than the gate-to-emitter voltage V in the switch-on state 205. Considering such a potential difference, the switch-on state 205 and the active clamp state 215 are shown separately. Further, note that the active clamp function of the IGBT control and drive circuit is effective in both the switch-on state 205 and the active clamp state 215. GE applied to the IGBT power switch is the same as in the switch-on state 205. However, this is not necessarily the case, and the magnitude of the gate-to-emitter voltage V by which the IGBT control and drive circuit drives the power switch does not have to be the same as the gate-to-emitter voltage V in the switch-on state 205. For example, in some embodiments, the gate-to-emitter voltage V in the active clamp state 215 may be slightly lower than the gate-to-emitter voltage V in the switch-on state 205. Considering such a potential difference, the switch-on state 205 and the active clamp state 215 are shown separately. Further, note that the active clamp function of the IGBT control and drive circuit is effective in both the switch-on state 205 and the active clamp state 215. GE by which the IGBT control and drive circuit drives the power switch does not have to be the same as the gate-to-emitter voltage V in the switch-on state 205. For example, in some embodiments, the gate-to-emitter voltage V in the active clamp state 215 may be slightly lower than the gate-to-emitter voltage V in the switch-on state 205. Considering such a potential difference, the switch-on state 205 and the active clamp state 215 are shown separately. Further, note that the active clamp function of the IGBT control and drive circuit is effective in both the switch-on state 205 and the active clamp state 215. GE in the switch-on state 205. For example, in some embodiments, the gate-to-emitter voltage V in the active clamp state 215 may be slightly lower than the gate-to-emitter voltage V in the switch-on state 205. Considering such a potential difference, the switch-on state 205 and the active clamp state 215 are shown separately. Further, note that the active clamp function of the IGBT control and drive circuit is effective in both the switch-on state 205 and the active clamp state 215. GE applied to the IGBT power switch may be slightly lower than the gate-to-emitter voltage V in the switch-on state 205. Considering such a potential difference, the switch-on state 205 and the active clamp state 215 are shown separately. Further, note that the active clamp function of the IGBT control and drive circuit is effective in both the switch-on state 205 and the active clamp state 215. GE in the switch-on state 205. Considering such a potential difference, the switch-on state 205 and the active clamp state 215 are shown separately. Further, note that the active clamp function of the IGBT control and drive circuit is effective in both the switch-on state 205 and the active clamp state 215.

[0034] The off state / anomaly state 220 causes a reflex transition 222 and a state transition 224. The reflex transition 222 indicates continuous monitoring for reset of the anomaly state. In the anomaly state, the IGBT control and drive circuit cannot drive the IGBT power switch to the on state. Reset can occur, for example, from a human operator, a delay circuit, or another circuit indicating that driving the IGBT power switch to the on state is once again permitted. The state transition 224 is triggered by such an indication and transitions the IGBT control and drive circuit to the off state / anomaly-free state 225.

[0035] Figure 3 is a schematic diagram of various waveforms 300, 305, 310, 315 during response to detection of a short circuit during driving of an IGBT power switch. In particular, waveform 300 represents the reference voltage V D as a function of time, waveform 305 represents the gate-to-emitter voltage V GE as a function of time, waveform 310 represents the collector current I C as a function of time, and waveform 315 represents the collector-to-emitter voltage V CE as a function of time. The scale is the same as the scale in Figure 1.

[0036] At the beginning of the period shown, the IGBT is in the conducting state. Both the reference voltage V D represented by waveform 300 and the gate-to-emitter voltage V GE represented by waveform 305 are at their respective substantially highest levels 320, 325. Further, the collector current I C represented by waveform 310 is at level 330 and the collector-to-emitter voltage V CE is at level 335. Generally, levels 320, 325 are selected to make the voltage drop across the IGBT relatively small and to reduce conduction losses.

[0037] At time T0, a short circuit occurs outside the power switch. The collector current represented by waveform 310 begins to increase rapidly. The overcurrent protection function in the IGBT control device is triggered in response to the detection of the short circuit state. As described above, after the short circuit state is detected, the overcurrent protection function in the IGBT control device starts a process to switch the IGBT off at time T1. The duration of the delay (i.e., the period between T0 and T1) is the period required to start short circuit protection. Again, the IGBT control and drive circuit does not initially drive the IGBT using the voltage that switches the IGBT to the off state. Rather, the IGBT control and drive circuit first uses a reference voltage V D level 320 to drive the IGBT, which reduces the charge carrier concentration compared to when it is driven using reference voltage V D level 340. Generally, this relatively low charge carrier concentration may be sufficient to keep the IGBT in the on state. Similar to the case of FIG. 1, for a typical IGBT, the voltage level 320 can be, for example, between 20 volts and 30 volts. And the voltage level 340 can be, for example, between 7 volts and 17 volts.

[0038] The IGBT control device and related circuits detect a high level of collector-to-emitter voltage V CE and respond at time T3 by raising the gate-to-emitter voltage V GE represented by waveform 305. The increase can be achieved using an active clamp function. For example, as shown in FIG. 4 below, any gate clamps can be removed, and the active clamp function can raise the voltage applied to the gate. In the illustrated embodiment, the gate-to-emitter voltage V GE rises to the same level 320 as in the on state by a generally step-like transition at time T3 on the time scale shown. This is not necessarily always the case. For example, in other embodiments, the gate-to-emitter voltage V GEmay be raised to a level between level 340 and level 320. As another example, in some implementations, the gate-to-emitter voltage V GE The rise in, for example, the collector-to-emitter voltage V CE or the collector-to-emitter voltage V CE may be more gradual, with the rate of change adjusted based on the rate of change of

[0039] Gate-to-emitter voltage V GE Regardless of the details of the rise in C In other words, the rate of decrease of the collector-to-emitter voltage V CE The rise in is proportional to the stray inductance of the commutation loop and the rate of change of the current through the IGBT power switch. By drawing more charge carriers into the channel of the IGBT power switch, the collector current I C Decrease rate of collector-emitter voltage V CE However, IGBTs require a relatively low and therefore safe level of collector-to-emitter voltage V CE There is no risk of going outside the safe operating range due to

[0040] At time T4, the collector-to-emitter voltage V CE has been reduced to such an extent that the IGBT power switches are unlikely to be damaged. The IGBT controller and associated circuitry are CE and--in response to a decrease--reducing the charge carrier concentration in the channel to provide a gate-to-emitter voltage V GE The gate voltage may again be reduced to a voltage level that results in a larger forward voltage drop across the channel than when driven using level 325 .

[0041] IGBT control devices and related circuits can detect the level of the collector-to-emitter voltage V CE in several different ways. For example, in some embodiments, a transient voltage suppressor in an active clamp circuit coupled to the collector of an IGBT power switch can passively monitor the collector-to-emitter voltage V CE . When the collector-to-emitter voltage V CE rises above the breakdown voltage of the transient voltage suppressor diode associated with the IGBT power switch, the active clamp circuit can use the current drawn from the collector to raise the gate bias, and additional charge carriers can be drawn into the channel of the IGBT power switch. As the collector-to-emitter voltage V CE drops below the threshold, the feedback from the collector of the IGBT power switch stops and the gate voltage drops. Other approaches are possible.

[0042] In the illustrated embodiment, the IGBT control and drive circuit drives the IGBT using the same reference voltage V D level 340 during the period from T1 to T3. However, this is not necessarily the case, and other reference voltage V D levels below level 320 or above level 320 can be used. The gate-to-emitter voltage V GE represented by waveform 305 drops more slowly than the reference voltage V D represented by waveform 300.

[0043] In some examples, the desaturation of the IGBT power switch is detected and the collector current I C reaches a level from which the IGBT can be safely driven into the off state. In some embodiments, this is a steady-state desaturated state. In response, the IGBT control and drive circuit drives the IGBT using a gate-to-emitter voltage V GE level 375 that is low enough (generally negative) to switch the IGBT into the off state.

[0044] However, in the illustrated embodiment, before the collector current I C reaches such a level, the collector-to-emitter voltage V CE represented by waveform 315 starts to increase again, and the IGBT is in a state where there is a risk of going out of the safe operating range again. The IGBT control device and the related circuit detect the increase in the collector-to-emitter voltage V CE and respond by enabling the reference voltage V D represented by waveform 300 to be increased at time t5. Again, an approximately step-like transition to level 320 is shown, but it is not essential. More charge carriers are drawn into the channel of the IGBT, and the rate of decrease of the collector current I C becomes smaller. Further, the collector-to-emitter voltage V CE represented by waveform 315 starts to decrease.

[0045] At time T6, the collector-to-emitter voltage V CE represented by waveform 315 has decreased to a level where the IGBT is less likely to go out of the safe operating range again. The IGBT control and drive circuit can monitor the collector-to-emitter voltage V CE and, in response to the decrease, reduce the charge carrier concentration in the channel and lower the gate-to-emitter voltage V GE to a voltage level that results in a larger channel forward voltage drop in the channel compared to when driven using level 325. The gate-to-emitter voltage V GE represented by waveform 305 further decreases.

[0046] As described above, it is assumed that the gate-to-emitter voltage V GE approaches level 355. Further, for a longer period (in this example, the period between T6 and T7), the collector current I CThere is a level, and it reaches a level from which the IGBT can be safely driven to the off state. In some examples, this level is a steady state level determined by both the applied gate voltage, the level 360 of the switched voltage, the temperature, and other factors. In any case, the desaturation protection circuit can detect the desaturation of the IGBT power switch, and in response at T7, the IGBT control and drive circuit applies a gate-to-emitter voltage V GE level 365 to drive the IGBT.

[0047] Again, the gate-to-emitter voltage V represented by waveform 305 GE drops more slowly than the reference voltage V in waveform 300 D . The collector current I C drops relatively rapidly during this transition and drops to approximately zero as the IGBT reaches the off state. The collector-to-emitter voltage V CE can transiently overshoot the level 360 of the switched voltage due to the rapid change in the collector current I C . Eventually, the collector-to-emitter voltage V CE stabilizes at the level 360 of the switched voltage.

[0048] Figure 4 is a schematic diagram of a circuit 400 involved in responding to the detection of a short circuit during the driving of an IGBT power switch. The circuit 400 includes the IGBT power switch 405 itself, and a driver circuit 410, an IGBT control device 415, and a short circuit detection and protection circuit 420.

[0049] The IGBT power switch 405 is shown as an n-channel device and includes a collector coupled to node 407, an auxiliary emitter (i.e., the bond wire between the auxiliary emitter and the main emitter) coupled to the internal parasitic inductance 425, and a main emitter node 409. Typically, the IGBT power switch 405 can withstand relatively high voltages, such as voltages between, for example, 600 volts and 6500 volts. The IGBT power switch 405 can be part of any of many different devices in different operating contexts. For example, the IGBT power switch 405 is typically part of a half-bridge topology as part of a phase leg in an inverter, or part of a motor drive, or part of a switching power converter.

[0050] The driver circuit 410 is coupled to the gate of the IGBT power switch 405. In the illustrated embodiment, the driver circuit includes a pull-up transistor 430, a pull-down transistor 435, and associated resistors that function as gate resistors for the IGBT power switch 405. Other embodiments are possible, including, for example, using additional voltage levels, transistors, and one gate resistor. For example, in some embodiments, two or more pull-up transistors may be provided to bias the gate of the IGBT power switch 405 to different levels (e.g., levels 320, 340 (FIG. 3)). In response to an on signal, the pull-down transistor 435 is switched off and the pull-up transistor 430 is switched to a conducting state, forming a conduction path between the power supply voltage 412 and the gate of the IGBT power switch 405. In the illustrated embodiment, the power supply voltage 412 has a fixed level with respect to the auxiliary emitter node 411. The level of the power supply voltage 412 is selected to keep the voltage drop across the IGBT power switch 405 relatively small. In the context of FIGS. 1 and 3, the power supply voltages 412 are levels 115 and 320.

[0051] In response to the off signal, the pull-up transistor 430 is switched off, and the pull-down transistor 435 is switched to the conducting state, forming a conduction path between the power supply voltage 414 and the gate of the IGBT power switch 405. In the illustrated embodiment, the voltage supplied by the power supply voltage 414 has a negative level with respect to the auxiliary emitter node 411. In the context of FIGS. 1 and 3, the power supply voltage 414 is at levels 145, 150, and levels 375, 365. Note that the IGBT control device 415 is configured to ensure that both transistors 430 and 435 are not switched to the conducting state simultaneously.

[0052] The IGBT control device 415 is a control device configured to control the switching of the IGBT power switch 405. The IGBT control device 415 includes a V CE detection terminal 440 and is involved in the active clamp function 445. V CE The detection terminal 440 is coupled across the IGBT power switch 405 to detect the collector-to-emitter voltage V CE .

[0053] The active clamp function 445 is configured to slow down the transition of the IGBT power switch 405 to the off state when a collector-emitter overvoltage occurs, and to suppress the transient voltage generated due to a rapid change in the collector current. In the illustrated embodiment, the active clamp function 445 includes a direct current path for charging the gate of the IGBT power switch 405 using the voltage at the collector of the IGBT power switch 405, and functions internal to the IGBT control device 415. The illustrated embodiment of the direct current path includes a Zener diode 450 (or similar device) and a diode 455 for charging the gate of the IGBT power switch 405 using the voltage at the collector of the IGBT power switch 405, and functions internal to the control device 415 of the IGBT. For example, in some embodiments, the active clamp function 445 may include a circuit for reducing the flow of current through the pull-down transistor 435 in response to detection of a collector overvoltage. By reducing this flow of current, the gate of the IGBT power switch 405 can be more rapidly charged via the Zener diode 450 (or similar device) and the diode 455.

[0054] Furthermore, in the illustrated embodiment, the active clamp function 445 includes an output terminal 460 coupled to control a switch in the short-circuit detection and protection circuit 420. During active clamping, the clamp circuit 475 is disabled and the gate voltage is allowed to increase.

[0055] The schematically illustrated embodiment of the short-circuit detection and protection circuit 420 includes a comparator 470, a clamp circuit 475, and a switch 480. The comparator 470 is coupled to compare the voltage across the inductor 425 to a reference voltage V R to detect a short-circuit condition. The voltage across the inductor 425 indicates the time rate of change of the collector current I C . In a practical embodiment, the detection of a short-circuit condition generally involves other components and functions. For example, a blanking circuit may limit the collector current IC It is used to eliminate the high rate of change.

[0056] Furthermore, for the purpose of explanation, a battery is shown as providing a reference voltage V with respect to the reference voltage 409. This is not necessarily the case. Generally, the reference voltage V R is set by a voltage divider or other voltage source. In some embodiments, the reference voltage V R can be set by the user, for example, based on a safe operating range, or other characteristics of the IGBT power switch 405, and / or the operating context of the IGBT power switch 405. R

[0057] The clamp circuit 475 is configured to reversibly clamp the gate voltage of the IGBT power switch 405 to a maximum value. For example, the clamp circuit 475 can clamp the voltage applied to the gate of the IGBT power switch 405 to levels 130, 340 (FIGS. 1, 3). Thus, the clamp circuit 475 forms a dynamic voltage suppressor and, in the shown embodiment, includes a Zener diode (or similar device) and a transistor switch. In the shown embodiment, the maximum value of the gate voltage of the IGBT power switch 405 is set by the breakdown voltage of the Zener diode (or similar device) and the voltage drop through the transistor switch. The transistor switch can switch to a conducting state and a non-conducting state to reversibly clamp and unclamp the gate voltage.

[0058] The switch 480 can be switched by the IGBT control device 415 to enable or prevent the comparator 470 from using the clamp circuit 475 to clamp the gate voltage of the IGBT power switch 405.

[0059] During operation, the IGBT control device 415 can control the transistors 430, 435 such that the IGBT power switch 405 is in an on state. The short-circuit detection and protection circuit 420 compares the rate of change of the collector current I C with a threshold reference voltage VR If a short circuit is detected, the comparator 470 compares - switching off the pull-up transistor 430, and - Biasing the clamp circuit 475 to clamp the gate voltage of the IGBT power switch 405 Such a signal may be output.

[0060] In the context of state transition diagram 200 (FIG. 2), this corresponds to state transition 209. If the clamp continues, the actual gate-to-emitter voltage V of the IGBT power switch 405 GE eventually drops to a clamped level (i.e., below the voltage level provided by the power supply voltage 412) and the number of charge carriers present in the channel of the IGBT power switch 405 decreases. When the IGBT power switch 405 reaches a desaturation level, the IGBT power switch 405 may be safely driven to an off state and the IGBT controller 415 may control the transistors 430, 435 to transition the IGBT power switch 405 to an off state. In the context of state transition diagram 200 (FIG. 2), this corresponds to state transition 213.

[0061] In some examples, the IGBT controller 415 may be configured to operate at a V CE 4. The active clamp function 445 detects an overvoltage via detection terminal 440. In these cases, the active clamp function 445 outputs a signal through terminal 460 to open switch 480, preventing comparator 470 from clamping the gate voltage of the IGBT power switch 405. Without clamping, the gate of the IGBT power switch 405 would rise due to the charge provided by active clamp function 445 (e.g., via a Zener diode 450 (or similar device) and resistor 455). In the context of state transition diagram 200 (FIG. 2), this corresponds to state transition 214.

[0062] When desaturation is detected, and the collector current I CNow, when it drops to a certain level, a level from which the IGBT power switch 405 can be safely driven to the off state, the IGBT control device 415 can control the transistors 430 and 435 to transition the IGBT power switch 405 to the off state so as to correspond to the state transition 213. When an overvoltage occurs again across the IGBT power switch 405, the active clamp function 445 resumes the active clamp so as to correspond to the state transition 214.

[0063] The transition between clamping the gate voltage of the IGBT power switch 405 and the increased gate voltage due to the active clamp can continue until the collector current I drops to a certain level, a level from which the IGBT power switch 405 can be safely driven to the off state so as to correspond to the state transition 213. C can decrease.

[0064] Although many embodiments have been described, it is understood that various modifications can be made. Therefore, other embodiments are within the scope of the claims described below.

[0065] Furthermore, the present invention can be further realized by one or more of the following embodiments.

[0066] Embodiment 1. A control system for a power switch including a channel, the system comprising a short - circuit detection circuit configured to detect an external short - circuit, an over - voltage detection circuit configured to detect an over - voltage applied to the power switch, and a short - circuit protection circuit. The short - circuit protection circuit is configured to change the control terminal of the power switch from a first potential level that provides a charge - carrier concentration in the channel of the power switch to a second potential level in response to detection of an external short - circuit by the short - circuit detection circuit while the power switch is in a first on - state, wherein the second potential level reduces the charge - carrier concentration in the channel. The short - circuit protection circuit is further configured to change the control terminal of the power switch from the second potential level to a third potential level that increases the charge - carrier concentration in the channel in response to detection of an over - voltage applied to the power switch by the over - voltage detection circuit while the control terminal of the power switch is at the second potential level.

[0067] Embodiment 2. The control system of Embodiment 1, wherein the short - circuit detection circuit is configured to detect an external short - circuit state based on a time - rate of change of current in the power switch.

[0068] Embodiment 3. The control system of Embodiment 1 or Embodiment 2, wherein the short - circuit protection circuit includes an active clamp circuit configured to decelerate the transition of the power switch to the off - state in response to detection of an over - voltage.

[0069] Embodiment 4. The control system of Embodiment 3, wherein the active clamp circuit is configured to charge the control terminal of the power switch using a voltage switched by the power switch.

[0070] Embodiment 5. The control system according to any one of Embodiments 1 to 4, wherein the short - circuit protection circuit includes a switchable clamp configured to clamp the control terminal of the power switch to the second potential level.

[0071] Embodiment 6. The control system according to Embodiment 5, wherein the short-circuit protection circuit includes an active clamp circuit configured to decelerate the transition of the power switch, and the short-circuit protection circuit is configured to terminate or reduce the clamp of the control terminal of the power switch in order to decelerate the transition.

[0072] Embodiment 7. The control system according to any one of Embodiments 1 to 6, wherein the short-circuit protection circuit is configured to clamp the control terminal of the power switch in response to the short-circuit detection circuit detecting an external short circuit.

[0073] Embodiment 8. The control system according to Embodiment 7, wherein the short-circuit protection circuit includes an active clamp circuit configured to decelerate the transition of the power switch, and the short-circuit protection circuit is configured to terminate or reduce the clamp of the control terminal of the power switch in order to decelerate the transition.

[0074] Embodiment 9. The third potential level is the first potential level. The control system according to any one of Embodiments 1 to 8.

[0075] Embodiment 10. The control system according to any one of Embodiments 1 to 9, wherein the power switch is an IGBT power switch, the voltage level of the first potential level is between 20 volts and 30 volts, and the voltage level of the second potential level is between 7 volts and 17 volts.

[0076] Embodiment 11. The control system according to any one of Embodiments 1 to 10, wherein when the control terminal is at the second potential level, the power switch remains in the second on state.

[0077] Embodiment 12. The control system according to any one of Embodiments 1 to 11, wherein the short-circuit protection circuit is further configured to change the control terminal of the power switch circuit to a fourth potential level that reduces the charge carrier concentration in the channel and switches the power switch to the off state in response to the detection of the desaturation of the power switch.

[0078] Embodiment 13. A method of controlling a power switch, the method comprising biasing the power switch in an on state to provide a first carrier concentration in a channel of the power switch; detecting an external short circuit that increases the flow of current through the power switch during the biasing of the power switch to provide the first carrier concentration; in response to the detection of the external short circuit, biasing the power switch to provide a second carrier concentration in the channel of the power switch, wherein the second carrier concentration is less than the first carrier concentration; detecting an overvoltage applied to the power switch during the biasing of the power switch to provide the second carrier concentration; and in response to the detection of the overvoltage, biasing the power switch to provide a third carrier concentration in the channel of the power switch, wherein the third carrier concentration is greater than the second carrier concentration.

[0079] Embodiment 14. The method according to Embodiment 13, further comprising detecting that the voltage applied to the power switch has dropped to a threshold level during the biasing of the power switch to provide the third carrier concentration; and biasing the power switch to provide a fourth carrier concentration in the channel of the power switch, wherein the fourth carrier concentration is less than the third carrier concentration.

[0080] Embodiment 15. The method according to Embodiment 14, further comprising repeatedly biasing the power switch to provide a higher carrier concentration in response to the detection of the overvoltage, and then to provide a lower carrier concentration in response to detecting that the voltage applied to the power switch has dropped to a threshold level.

[0081] Embodiment 16. The method according to any one of Embodiments 13 to 15, wherein the potential level for providing the first carrier concentration is the same as the potential level switch for providing the third carrier concentration, and the potential levels for providing the second carrier concentration and the fourth carrier concentration are the same.

[0082] Embodiment 17. The method according to any one of Embodiments 13 to 16, wherein detecting an external short circuit includes detecting that a rate of change over time of a current flowing through the power switch is greater than a threshold value.

[0083] Embodiment 18. Biasing the power switch to provide a second carrier concentration includes clamping the control terminal of the power switch to a voltage level, and biasing the power switch to provide a third carrier concentration includes unclamping the control terminal of the power switch from the voltage level. The method according to any one of Embodiments 13 to 17.

[0084] Embodiment 19. Biasing the power switch to provide a third carrier concentration includes biasing the control terminal of the power switch using a voltage switched by the power switch. The method according to any one of Embodiments 13 to 18.

[0085] Embodiment 20. The power switch is an IGBT power switch, the first carrier concentration is provided by a gate-to-emitter voltage between 20 volts and 30 volts, and the second carrier concentration is provided by a gate-to-emitter voltage between 7 volts and 17 volts. The method according to any one of Embodiments 13 to 19.

[0086] Embodiment 21. The method according to any one of Embodiments 13 to 20, wherein the switch remains in the on state at the second carrier concentration and the third carrier concentration.

[0087] Embodiment 22. The method according to any one of Embodiments 13 to 21, further comprising detecting saturation of the power switch and biasing the power switch to the off state in response to detecting saturation of the power switch.

Claims

1. A control system for a power switch including a channel, wherein the control system a short-circuit detection circuit configured to detect an external short circuit; an overvoltage detection circuit configured to detect an overvoltage applied to the power switch; a short-circuit protection circuit, while the power switch is in a first on state, in response to detection of an external short circuit by the short-circuit detection circuit, changing a control terminal of the power switch from a first potential level that provides a charge carrier concentration in the channel of the power switch to a second potential level, wherein the second potential level reduces the charge carrier concentration in the channel; while the control terminal of the power switch is at the second potential level, in response to detection of an overvoltage applied to the power switch by the overvoltage detection circuit, changing the control terminal of the power switch from the second potential level to a third potential level that increases the charge carrier concentration in the channel; the short-circuit protection circuit configured to do the above; A control system comprising the above.

2. The control system according to claim 1, wherein the short-circuit detection circuit is configured to detect a state of the external short circuit based on a rate of change of current in the power switch. The control system according to claim 1.

3. The control system according to claim 1, wherein the short-circuit protection circuit includes an active clamp circuit configured to decelerate a transition of the power switch to an off state in response to detection of the overvoltage. The control system according to claim 1.

4. The control system according to claim 3, wherein the active clamp circuit is configured to charge a control terminal of the power switch using a voltage switched by the power switch. The control system according to claim 3.

5. The control system according to claim 1, wherein the short-circuit protection circuit includes a switchable clamp configured to clamp the control terminal of the power switch at the second potential level. The control system according to claim 1.

6. The control system according to claim 5, wherein the short-circuit protection circuit includes an active clamp circuit configured to decelerate a transition of the power switch, and the short-circuit protection circuit is configured to end or reduce a clamp of the control terminal of the power switch to decelerate the transition. The control system according to claim 5.

7. The short-circuit protection circuit is configured to clamp the control terminal of the power switch in response to the short-circuit detection circuit detecting the external short circuit. The control system according to claim 1.

8. The short-circuit protection circuit includes an active clamp circuit configured to decelerate the transition of the power switch. The short-circuit protection circuit is configured to end or reduce the clamp of the control terminal of the power switch in order to decelerate the transition. The control system according to claim 7.

9. The third potential level is the first potential level. The control system according to claim 1.

10. The power switch is an IGBT power switch. The voltage level of the first potential level is between 20 volts and 30 volts. The voltage level of the second potential level is between 7 volts and 17 volts. The control system according to claim 1.

11. When the control terminal is at the second potential level, the power switch remains in the second on state. The control system according to claim 1.

12. The short-circuit protection circuit is further configured to change the control terminal of the circuit of the power switch to a fourth potential level to reduce the charge carrier concentration in the channel and switch the power switch to an off state in response to detection of desaturation of the power switch. The control system according to claim 1.

13. A method of controlling a power switch, the method comprising: Biasing the power switch in an on state to provide a first carrier concentration in the channel of the power switch; Detecting an external short circuit that increases the flow of current through the power switch while biasing the power switch to provide the first carrier concentration; Biasing the power switch to provide a second carrier concentration in the channel of the power switch in response to detection of the external short circuit, the second carrier concentration being less than the first carrier concentration; Detecting an overvoltage applied to the power switch while biasing the power switch to provide the second carrier concentration. Biasing the power switch to provide a third carrier concentration within the channel of the power switch in response to detection of the overvoltage, the third carrier concentration being greater than the second carrier concentration. A method comprising. **Claim 14** During biasing of the power switch to provide the third carrier concentration, detecting that the voltage across the power switch has dropped to a threshold level. Biasing the power switch to provide a fourth carrier concentration within the channel of the power switch, the fourth carrier concentration being less than the third carrier concentration. The method according to claim 13, further comprising. **Claim 15** First, in response to detection of an overvoltage, providing a higher carrier concentration, and then, in response to detecting that the voltage across the power switch has dropped to the threshold level, repeatedly biasing the power switch to provide a lower carrier concentration. The method according to claim 14. **Claim 16** The potential level for providing the first carrier concentration is the same as the potential level switch for providing the third carrier concentration. The potential levels for providing the second carrier concentration and the fourth carrier concentration are the same. The method according to claim 14. **Claim 17** Detecting the external short circuit includes detecting that the time rate of change of the current flowing through the power switch is greater than a threshold value. The method according to claim 14. **Claim 18** Biasing the power switch to provide the second carrier concentration includes clamping the control terminal of the power switch to a voltage level. Biasing the power switch to provide the third carrier concentration includes unclamping the control terminal of the power switch from the voltage level. The method according to claim 14. **Claim 19** Biasing the power switch to provide the third carrier concentration includes biasing the control terminal of the power switch using the voltage switched by the power switch. The method according to claim 14. **Claim 20** The power switch is an IGBT power switch. The first carrier concentration is provided by a gate-to-emitter voltage between 20 volts and 30 volts, The second carrier concentration is provided by a gate-to-emitter voltage between 7 volts and 17 volts, The method according to claim 14.

21. The power switch remains in the on state at the second carrier concentration and the third carrier concentration, The method according to claim 13.

22. Detecting desaturation of the power switch, Biasing the power switch to an off state in response to the detection of the desaturation of the power switch, The method according to claim 13, further comprising.

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