Drive control device for synchronous motor

Abstract

The present invention relates to a drive control device for a synchronous motor. To provide a drive controller of a synchronous motor capable of suppressing torque ripples of sixth order, twelfth order, etc., which cannot be suppressed by a basic drive controller.SOLUTION: The drive control device of the synchronous motor comprises at least a means for generating a final current command value for suppressing a specified torque ripple and a current control means for performing discrete time control of a stator current so as to follow the final current command value, wherein a discrete time compensating current controller arranged in parallel to a PI current controller and having a pole on a unit circle with a zero point as an origin and multiplying a control gain and a control period by a control deviation is added to the discrete time current controller being a central device of the current control means.SELECTED DRAWING: Figure 5

Description

[Technical Field]

[0001] The present invention relates to a drive control device for a synchronous motor, and in particular to a drive control device for a synchronous motor that can suppress 6th, 12th, and other torque ripples that cannot be fully suppressed by a basic drive control device. [Background technology]

[0002] In the present invention, frequency and angular frequency are used synonymously, as is widely used today among those skilled in the art. Similarly, velocity and angular velocity are used synonymously. The units of frequency, angular frequency, velocity, and angular velocity are [rad / s] unless otherwise specified.

[0003] In the present invention, the unit of time is the second "s." Further, the unit of the control period Ts of the discrete time control is the second.

[0004] As is well known to those skilled in the art, in the synchronous motor of the present invention, there is a one-to-one correspondence between the rotor electrical speed ω2n and the mechanical speed ω2m. Therefore, in the present invention, unless otherwise specified, the rotor speed is expressed using the "electrical speed."

[0005] High-performance control of synchronous motors can be achieved by the so-called vector control method. When using basic vector control, even if the power converter has ideal characteristics, the generated torque has periodic ripples (pulsations) according to the rotor speed due to the spatial distortion characteristics of synchronous motors, represented by non-sinusoidal speed electromotive force.

[0006] Traditionally, compensation for ripples in the generated torque has been achieved by superimposing an appropriate compensation signal on the initial torque command value or the corresponding initial current command value. This method can be expected to provide effective compensation if the current control system has a reasonable degree of tracking ability for the compensation signal. For reference, Figure 8 shows the torque ripple compensation method used in Patent Document (2) listed below.

[0007] Torque ripple compensation methods for synchronous motors can be classified from various viewpoints. However, considering the basic principles of ripple compensation described above, it is appropriate to classify them based on the method of generating the compensation signal and the method of controlling the stator current (see Patent Documents (1)-(2) and Non-Patent Documents (1)-(9) listed below).

[0008] The methods of generating the compensation signal are classified into those that do not require a sensor such as a torque sensor for detecting the torque ripple equivalent signal (hereinafter referred to as sensorless) (Patent documents (1)-(2), Non-patent documents (1)-(3), (9)), and those that require the use of a torque sensor or the like (hereinafter referred to as sensor-use) (Non-patent documents (4)-(8)).

[0009] The former type of sensorless motors is further subdivided into those that estimate a torque ripple equivalent signal online (i.e., estimate it in real time) based on a mathematical model (see Patent Document (2) and Non-Patent Document (9)), and those that generate a torque ripple equivalent signal using data acquired offline in advance (see Patent Document (1) and Non-Patent Document (9)).

[0010] Stator current control methods for torque ripple compensation include those that use only a basic current controller, such as a PI current controller, and those that use a current controller that is designed to have high tracking ability to the torque ripple in addition to the basic current controller.

[0011] Shinnaka has proposed a method of using a high-order current controller in combination with a basic current controller, such as a PI current controller, and a current controller that pursues high tracking performance for torque ripple (see patent document (2) and non-patent document (9)). Nakamura et al. have also proposed the use of a feedforward controller that utilizes a motor inverse model (non-patent documents (7) and (8)).

[0012] Shinnaka's high-order current controller method performs feedback current control using a third-order, fifth-order, or both high-order current controller. For reference, the third-order, fifth-order, or both continuous-time high-order current controller Gcnt(s) is shown below, with the electrical speed ω2n and the time differential operator s.

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[0013] Feedback current control generally exhibits high robustness against motor parameter variations. However, high tracking and stability are mutually exclusive, and pursuing high tracking can compromise stability. In a high-order current controller (third-order, fifth-order, or both), it is desirable to simultaneously change not only the electrical speed ω2n of the denominator polynomial constituting the high-order current controller but also the gain d of the numerator polynomial in response to the motor electrical speed ω2n. If the gain change of the numerator polynomial in equations (1) and (2) is halted, the operating range within which reasonable performance is obtained is restricted. According to patent document (2) and non-patent document (9), in practice, the operating range within which reasonable performance is obtained is restricted, and this is at the expense of halting the gain change of the numerator polynomial. In a continuous-time high-order current controller that simultaneously suppresses sixth- and twelfth-order torque ripples as shown in equation (3), it is difficult to maintain the gain d of the numerator polynomial unchanged regardless of the electrical speed.

[0014] As shown in the second equation of each of equations (1)-(3), the continuous-time high-order current controller can be interpreted as a "controller arranged in series" that is a continuous-time current controller that pursues high tracking ability to torque ripple of 6th order, 12th order, or both, in comparison with a basic current controller represented by a PI current controller.

[0015] Instead, Nakamura et al.'s feedforward control does not impair stability. However, this control method is highly sensitive to motor parameter fluctuations, and if the parameters used differ from the actual parameters, the control performance may be degraded contrary to expectations. [Prior art documents] [Patent documents]

[0016] [Patent Document 1] Kazuo Shimane: "Elevator Control Device", Unexamined Patent Publication, JP 2001-31339 (July 27, 1999) [Patent Document 2] Shinnaka Shinji: "Drive control device for synchronous motor", Published Patent Application No. 2012-100510 (October 31, 2010) [Non-patent literature]

[0017] [Non-Patent Document 1] Yoshihisa Hojo, Yoichi Omori, Shigenori Hagiwara, Takashi Kosaka, and Nobuyuki Matsui, "Torque Ripple Reduction Control of IPMSM with Concentrated Windings," Proceedings of the 2004 Industrial Applications Division Conference of the Institute of Electrical Engineers of Japan, I, pp. 499-502 (August 2004) [Non-patent document 2] K. Yoshimototo and Y. Kitajima: “A Novel Harmonic Current Control for IPMSMs”, Proc. of the 2005 International Power Electronics Conference (IPEC-Niigata 2005), pp. 2042-2048 (2005-4) [Non-patent document 3] N. Nakao and K. Akatsu: “A New Control Method for Torque Ripple Compensation of Permanent Magnet Motors”, Proc. of the 2010 International Power Electronics Conference (IPEC - Sapporo 2010), pp. 1421 - 11427 (2010 - 6)

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[0018] The present invention was made against the above background, and its objectives can be summarized as follows: To provide a drive control device for a synchronous motor that simultaneously has high stability and high tracking capability over a wide operating range by using a discrete-time current controller in which the current controller gain to be changed is only a gain directly related to the rotor speed (electrical speed or equivalent mechanical speed) over a range from low to high speeds, in response to a final current command value whose frequency component changes according to the rotor speed. Furthermore, to provide a drive control device for a synchronous motor with non-sinusoidal speed electromotive force characteristics that simultaneously has high stability and high tracking capability over a wide operating range, and that is capable of efficiently reducing torque ripple, particularly that including sixth-order, twelfth-order, or both. [Means for solving the problem]

[0019] In order to achieve the above object, the invention of claim 1 comprises: final current command value generating means for generating a compensation signal for compensating for an initial torque command value or an initial current command value, and using the generated compensation signal to compensate for the initial torque command value or the initial current command value, thereby generating a final current command value; and current control means including a discrete-time current controller for capturing the stator current as a vector signal on a dq synchronous coordinate system of two orthogonal axes with the rotor N-pole phase as the d-axis phase, and for discrete-time controlling the stator current so as to follow the final current command value. A drive control device for a synchronous motor including at least: when "Ts" is a control period of discrete time control, "z-1" is a delay operator, "ω0" is a frequency of a component to be compensated included in the compensation signal, the product signal of the control period of discrete time control and the frequency of the compensation signal is "ω-0", and the gain is "Kh", the discrete time current controller is arranged in parallel with a basic current controller represented by a PI current controller and selects one of the following discrete time compensation current controllers Gh(z-1) having the following control period "Ts":

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[0020] The invention of claim 2 is a drive control device for a synchronous motor according to claim 1, wherein when a rotor electrical speed is "ω2n", the product signal "ω-0" of the discrete time compensation current controller, which is arranged in parallel with the basic current controller and has a control period "Ts", is expressed as one of the following equations (5):

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[0021] The effect of the invention of claim 1 will be explained. Since it is believed to be useful in explaining the effect of the present invention, the common basic principle of the torque ripple compensation methods used in patent documents (1), (2), non-patent documents (1)-(9), etc. will be explained first. When a torque command value τ* is given to a vector-controlled synchronous motor, the synchronous motor generates a torque τ. The generated torque τ at this time is the sum of a fundamental component τf resulting from the fundamental wave component of the rotor magnetic flux and speed electromotive force, and a ripple component τh resulting from this harmonic component. That is,

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[0022] Now, let us assume that the ripple component τh can be obtained somehow. The torque command value τ* is synthesized as shown in the following equation (7) using the basic component command value τf* corresponding to the basic component τf and the compensation signal τh^≒τh for canceling and compensating for the ripple component τh.

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[0023] Figure 1(b) shows an example of the generated torque when the base command value τf* is constant. The torque command value τ* shown on the left side of Figure 1(b) is synthesized by superimposing (in this case, subtracting) a compensation signal τh for ripple cancellation compensation on the constant base command value τf*. By including the compensation signal τh in the torque command value τ*, a constant generated torque τ is obtained. This is the basic principle of torque ripple compensation.

[0024] This principle involves obtaining a ripple τh contained in the generated torque τ and using a signal equivalent to the obtained ripple to generate a compensation signal for canceling the ripple. Therefore, in order to compensate for the appearance of the ripple in the generated torque based on this principle, it is a prerequisite that at least the ripple τh contained in the generated torque τ can be obtained. Regarding obtaining the ripple τh contained in the generated torque τ, a signal equivalent to the torque ripple may be generated using data acquired offline, as in Patent Document (1), or a signal equivalent to the torque ripple may be estimated online based on a mathematical model, as in Patent Document (2).

[0025] In the present invention, a series of steps including generating a signal equivalent to the torque ripple τh, and further compensating an initial torque command value or an initial current command value substantially equivalent thereto using the generated torque ripple equivalent signal to generate a final current command value is referred to as a "final current command value generating means." In the present invention, it is assumed that the final current command value generating means utilizes known means. In other words, the final current command value generating means is not the gist of the present invention, and therefore further explanation will be omitted.

[0026] The gist of the invention of claim 1 lies in the current control means. The current control means of the present invention is configured by arranging one of the "discrete-time compensation current controllers Gh(z-1) having Ts" shown in equation (4) in parallel with a basic current controller GPI(z-1) represented by a PI current controller. For example, when equation (4a) in equation (4) is used, the discrete-time current controller Gcnt(z-1), which is the main device constituting the current control means, is written as follows:

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[0027] The frequency ω0 of the component to be compensated for contained in the compensation signal is an integer multiple of the rotor electrical speed ω2n. Therefore, the compensation signal ω0 also changes in response to changes in the rotor electrical speed. Consequently, the product signal ω-0 shown at the end of the right-hand side of equation (9) also changes in response to the electrical speed. As equation (9) clearly shows, only the product signal inside the cosine function changes in response to the electrical speed. In principle, there is no need to change any of the other controller coefficients. Consequently, according to the invention of claim 1, it is possible to configure a current control system that exhibits high tracking capability while maintaining high stability for current command values ​​having frequency components according to speed over a very wide range, from low speeds to speeds exceeding the rated speed.

[0028] In equation (9), the discrete-time compensation current controller is expressed as a single expression. This is to simplify the explanation of the effect. The same effect can be obtained when using multiple discrete-time compensation current controllers using different product signals ω-0 (see equations (10) to (12), Figure 5, and related explanations below). Similarly, in equation (9), equation (4a) is used as the discrete-time compensation current controller, but the same effect can be obtained when equation (4b) is used.

[0029] The effect of the invention of claim 2 will be explained. In many synchronous motors, the main component of torque ripple is sixth order, twelfth order, or both. As a result, the main component of the final current command value for compensating for torque ripple is sixth order, twelfth order, or both. In accordance with the invention of claim 2, the discrete-time current controller Gcnt(z-1), which is the main device constituting the current control means for compensating for torque ripple of sixth order, twelfth order, or both sixth and twelfth order components, is shown below using equation (5a) in equation (5).

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[0030] According to the invention of claim 2, the coefficients of the denominator polynomial of the discrete-time compensation current controller are changed according to 6 or 12 times the rotor electrical speed, or both. As a result, according to the invention of claim 2, it is possible to independently and simply configure current control means that tracks the main component of the final current command value. This in turn makes it possible to simply configure current control means that contributes to reducing the main component of torque ripple. Note that, although equation (5a) is used as the discrete-time compensation current controller in equations (10) to (12), the same effect can be obtained by using equation (5b). [Brief explanation of the drawings]

[0031] [Figure 1] "Graph showing the general relationship between torque command value and generated torque" [Figure 2] "Block diagram showing the basic configuration of a drive control device in one embodiment" [Figure 3] "Block diagram showing the basic configuration of a final current command value generator in one embodiment" [Figure 4] "Overall block configuration of a discrete-time current controller in one embodiment" [Figure 5] "Block configuration of the q-axis discrete-time current controller in the discrete-time current controller in one embodiment" [Figure 6] "Block configuration for a discrete-time compensation current controller for reducing the sixth-order torque ripple component in the discrete-time current controller for the q axis in one embodiment" [Figure 7] "Block configuration for a discrete-time compensation current controller for reducing the 12th-order torque ripple component in the q-axis discrete-time current controller in one embodiment" [Figure 8] "Block diagram showing the basic configuration of a conventional drive control device" DETAILED DESCRIPTION OF THE INVENTION

[0032] An embodiment of the present invention will be described in detail below with reference to the accompanying drawings. Figure 2 shows an example of a drive control device of the present invention applied to a synchronous motor. Reference numeral 1 denotes a synchronous motor, 2 denotes a power converter, 3 denotes a current detector, 4a and 4b denote a three-phase to two-phase converter and a two-phase to three-phase converter, respectively, 5a and 5b denote vector rotators, 6 denotes a discrete-time current controller using the discrete-time compensation current controller of the present invention, 7 denotes a phase detector, 8 denotes a speed detector, 9 denotes a cosine / sine signal generator, and 10 denotes a final current command value generator that realizes a final current command value generating means. In Figure 2, the drive control device is comprised of devices 2 through 10, excluding the motor 1. For clarity, a 2x1 vector signal is represented by a single thick signal line in this diagram. This is also followed in the block diagrams that follow. In this figure, the suffixes r (dq synchronous coordinate system), s (αβ fixed coordinate system), and t (uvw coordinate system) are added to the vector signals to clearly indicate the coordinate systems in which the stator current and stator voltage, which are vector signals, are defined.

[0033] The three-phase stator currents obtained in a discrete time manner by the current detector 3 are converted into two-phase currents in an αβ fixed coordinate system by a three-phase to two-phase converter 4a, and then converted into two-phase currents in a dq synchronous coordinate system by a vector rotator 5a. The converted currents are sent to a discrete-time current controller 6. The discrete-time current controller 6 generates two-phase voltage command values ​​in the dq synchronous coordinate system so that the two-phase currents in the dq synchronous coordinate system follow the final current command values ​​of each phase. The two-phase voltage command values ​​in the dq synchronous coordinate system are sent to a vector rotator 5b. 5b converts the voltage command values ​​in the dq synchronous coordinate system into two-phase voltage command values ​​in the αβ fixed coordinate system and sends them to a two-phase to three-phase converter 4b. 4b converts the two-phase voltage command values ​​into three-phase voltage command values ​​and outputs them as final command values ​​to the power converter 2. The power converter 2 generates voltages in a discrete time manner according to the command values ​​and applies them to the synchronous motor 1 to drive it.

[0034] The final current command value generator 10 in the embodiment of FIG. 2 receives an initial torque command value τf* and a rotor phase θα as input signals, and outputs a final d-axis current command value id* and a q-axis current command value Iq* to the discrete-time current controller 6. As shown in the diagram, the internal configuration of the final current command value generator 10 is such that the d-axis current command value is forcibly set to zero. On the other hand, the q-axis current command value is obtained as the output of a q-axis current command value generator 10a that processes the initial torque command value τf*, which is an input signal. Both the d-axis current command value and the q-axis current command value are input to the discrete-time current controller 6 according to the present invention.

[0035] Figure 3, based on Non-Patent Document (9), shows the internal configuration of the q-axis current command generator 10a, a key component of the final current command generator 10. In the figure, Φ and Np represent the rotor magnetic flux intensity (fundamental component) and the number of pole pairs specific to the synchronous motor. The relative speed electromotive forces of the 5th, 7th, 11th, and 13th harmonic components relative to the fundamental component of the speed electromotive force are expressed as w5, w7, w11, and w13, respectively (expressed as relative magnetic flux intensities: w5 / 5, w7 / 7, w11 / 11, and w13 / 13). The relative speed electromotive forces have already been obtained through preliminary experiments, etc. The q-axis current command generator 10a in Figure 3 generates a compensation signal iqh* from the initial current command iqf* and adds the compensation signal iqh* to the initial current command iqf* to obtain the final current command iq*. The relative speed electromotive force and the relative magnetic flux strength have already been explained in detail in Non-Patent Document (9) well known to those skilled in the art, and therefore further explanation will be omitted.

[0036] As is already clear from the above explanation, in the drive control device made up of the devices 2 to 11, the devices 2 to 9 realize current control means that captures the stator current as a vector signal on a dq synchronous coordinate system of two orthogonal axes with the rotor N-pole phase as the d-axis phase, and controls it to follow the final current command value. In addition, final current command value generator 10 constitutes final current command value generation means that generates the final current command value.

[0037] The core of the present invention lies in the discrete-time current controller 6, which is one of the components of the drive control device. While the figure illustrates the case of torque control mode, the configuration of the discrete-time current controller 6 according to the present invention is the same in other control modes such as speed control mode. Below, we will explain an embodiment of the discrete-time current controller 6, which does not need to be changed depending on the control mode. [Example]

[0038] FIG. 4 shows an example of the configuration of a discrete-time current controller based on the inventions of claims 1 and 2. Naturally, separate discrete-time current controllers are required for the d-axis and the q-axis. The same discrete-time current controller can be used for the d-axis and the q-axis. As a specific example of a discrete-time current controller, one of equations (10) to (11) based on equations (4a) and (5a) can be used. Which of equations (10) to (11) should be used depends on the strength of the harmonic components of the rotor magnetic flux of the synchronous motor to be controlled (or the strength of the harmonic components of the speed electromotive force), and cannot be generalized. If you want to reduce the torque ripple of the 6th order component, use equation (10); if you want to reduce the torque ripple of the 12th order component, use equation (11); and if you want to reduce the torque ripple of both the 6th order and the 12th order components simultaneously, use equation (12). Equation (12) can be used.

[0039] 6a in Figure 5 shows an example of the internal configuration of a discrete-time current controller (for q-axis current control) that assumes the simultaneous reduction of the sixth- and twelfth-order torque ripple components. Please refer to equation (12) to see the state of the discrete-time compensation current controller arranged in parallel with the PI current controller (basic current controller). The PI current controller (basic current controller) arranged in the upper part of the figure uses, without modification, a method known to those skilled in the art and used in non-patent document (9) and elsewhere. The second and third stages of the figure are discrete-time compensation current controllers according to the present invention, which are responsible for suppressing the sixth-order and twelfth-order torque ripple components in parallel, respectively.

[0040] 6a-6 in Figure 6 shows an example of the configuration of a discrete-time compensation current controller (for q-axis current control) that suppresses the sixth-order component of torque ripple, which is used in 6a in Figure 5. Please note that the control deviation of the q-axis current, "iq*-iq," is input as the difference between the direct value and the value delayed by two control periods after static coefficient processing, that is, as the numerator polynomial "1-(z-2)" in equation (12). [Example]

[0041] Instead of formulas (10) and (11), any of the following formulas based on formulas (4b) and (5b) may be adopted.

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[0042] The only difference between the above equations (13) to (15) and the previously explained equations (10) to (11) is the numerator polynomial of the discrete-time compensation current controller. Formally, the numerator polynomial is replaced by "1-(z-2)" to "1-(z-1)." Therefore, the parallel arrangement shown in Figure 5 also holds. However, the configurations of the individual discrete-time compensation current controllers are not identical. To illustrate this, Figure 7 shows an example configuration of a discrete-time compensation current controller for suppressing the 12th-order torque ripple component. Note that the control deviation of the q-axis current, "iq*-iq," is input as the difference between the direct value and the value delayed by one control cycle after static coefficient processing, i.e., as the numerator polynomial "1-(z-1)" in equation (15). [Example]

[0043] In the first embodiment, a discrete-time compensation current controller based on equations (10) to (11) was described. Alternatively, in the second embodiment, a discrete-time compensation current controller based on equations (13) to (15) was described. Alternatively, it is possible to use the discrete-time compensation current controller based on equations (10) to (11) for suppressing sixth-order torque ripple, and the discrete-time compensation current controller based on equations (13) to (15) for suppressing twelfth-order torque ripple. The reverse is also possible. [Example]

[0044] Similarly, it is also possible to use a discrete-time compensation current controller based on equations (10) to (11) for q-axis current control, and alternatively use a discrete-time compensation current controller based on equations (13) to (15) for d-axis current control, or vice versa. [Example]

[0045] The following describes the gain design of the controller. The gains of the discrete-time current controllers proposed in this invention can be designed independently of each other. Regarding the gain design of the PI current controller, which is a fundamental wave current controller, the gain design method detailed in Non-Patent Document (9) can be used without modification. In this case, there is no need to consider the existence of the discrete-time compensation current controller.

[0046] The design of the 6th-order gain Kh6 of the discrete-time compensation current controller does not need to take into account the presence of a PI current controller, which is a fundamental wave current controller, or a 12th-order discrete-time compensation current controller. Similarly, the design of the 12th-order gain Kh12 of the discrete-time compensation current controller does not need to take into account the presence of a PI current controller, which is a fundamental wave current controller, or a 6th-order discrete-time compensation current controller.

[0047] Although it depends on the characteristics of the synchronous motor to be controlled, the effective gain design of the discrete time compensation current controller is generally the following value when the control period is set to Ts=0.0001 [s], for example.

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[0048] When using any of the formulas (13) to (15) based on formulas (4b) and (5b), the gain may become too high during high-speed rotation. In this case, the gain may be attenuated and corrected as follows according to the rotor speed:

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[0049] The present invention is particularly suited to synchronous motor applications requiring high quality torque generation. [Explanation of symbols]

[0050] 1 synchronous motor 2. Power Converter 3 Current detector 4a 3-phase to 2-phase converter 4b 2-phase to 3-phase converter 5a Vector Rotator 5b Vector Rotator 6 Discrete-time current controller 6a q-axis discrete time current controller 7 Phase Detector 8 Speed ​​Detector 9 cosine sine signal generator 10 Final current command generator 10a q-axis current command generator

Claims

1. a final current command value generating means for generating a compensation signal for compensating for an initial torque command value or an initial current command value, and for generating a final current command value by compensating the initial torque command value or the initial current command value using the generated compensation signal; current control means including a discrete-time current controller that captures the stator current as a vector signal on a dq synchronous coordinate system of two orthogonal axes with the rotor N-pole phase as the d-axis phase, and controls the stator current in discrete time so that it follows the final current command value; A drive control device for a synchronous motor comprising at least When "Ts" is the control period of the discrete time control, "z-1" is the delay operator, "ω0" is the frequency of the component to be compensated contained in the compensation signal, the product signal of the control period of the discrete time control and the frequency of the compensation signal is "ω-0", and the gain is "Kh", The discrete-time current controller selects one of the following discrete-time compensation current controllers having the following control period "Ts" arranged in parallel with the basic current controller represented by the PI current controller: A drive control device for a synchronous motor, comprising:

2. When the rotor electrical speed is "ω2n", the product signal "ω-0" of the discrete time compensation current controller having a control period "Ts" arranged in parallel with the basic current controller is expressed by the following equation: or the following equation 2. A drive control device for a synchronous motor according to claim 1, wherein the rotational speed is varied in accordance with six or twelve times the rotor electrical speed "ω2n", or both of them.

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