Semiconductor devices, communication devices, imaging systems and radar devices

Abstract

To provide a technology advantageous for communication using terahertz waves. [Solution] A semiconductor device comprising: a reference signal oscillator that generates a reference signal; a first injection-locked oscillator and a second injection-locked oscillator that oscillate at a frequency in the terahertz band in synchronization with the reference signal; a first modulator connected to the first injection-locked oscillator; a second modulator connected to the second injection-locked oscillator; and a signal transmission unit that transmits the signal output from the first modulator and the signal output from the second modulator as electromagnetic waves, wherein the phase of the signal supplied from the first injection-locked oscillator to the first modulator is shifted from the phase of the signal supplied from the second injection-locked oscillator to the second modulator.

Description

[Technical Field]

[0001] The present invention relates to a semiconductor device, a communication device, an imaging system, and a radar device. [Background technology]

[0002] Development of oscillators and detectors for wireless communication systems using terahertz waves is underway. Patent Document 1 discloses a phased array radio that transmits and receives signals at frequencies of 300 GHz or higher. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Japanese Patent Publication No. 2023-010120 Summary of the Invention [Problem to be solved by the invention]

[0004] Generally, as the frequency increases, the wavelength becomes shorter, and so the size of the antenna becomes smaller. In a phased array antenna, the antenna spacing must be equal to or less than the wavelength to suppress grating lobes. However, the circuit scale becomes larger due to increased high-frequency loss. Furthermore, complex multiplexing processes such as quadrature-phase modulation (QAM modulation) are used in communication systems. In order to achieve smaller antennas and array configurations, the oscillator circuits and modulator circuits must also be made smaller.

[0005] An object of the present invention is to provide a technique that is advantageous for communication using terahertz waves. [Means for solving the problem]

[0006] In view of the above problems, a semiconductor device according to an embodiment of the present invention is a semiconductor device comprising: a reference signal oscillator that generates a reference signal; a first injection-locked oscillator and a second injection-locked oscillator that oscillate at a frequency in the terahertz band in synchronization with the reference signal; a first modulator connected to the first injection-locked oscillator; a second modulator connected to the second injection-locked oscillator; and a signal transmission unit that transmits a signal output from the first modulator and a signal output from the second modulator as electromagnetic waves, wherein the phase of the signal supplied from the first injection-locked oscillator to the first modulator is shifted from the phase of the signal supplied from the second injection-locked oscillator to the second modulator. [Effects of the Invention]

[0007] According to the present invention, it is possible to provide a technique that is advantageous for communication using terahertz waves. [Brief explanation of the drawings]

[0008] [Figure 1] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 2] FIG. 4 is a diagram showing the injection locking characteristics of the RTD oscillator of the present embodiment. [Figure 3] FIG. 2 is a diagram showing a more detailed configuration example of the semiconductor device of the present embodiment. [Figure 4] FIG. 2 is a diagram showing an example of the configuration of a phase adjuster according to the present embodiment. [Figure 5] FIG. 2 is a diagram showing an example of the configuration of a filter circuit according to the present embodiment. [Figure 6] FIG. 2 is a diagram showing an example of an equivalent circuit of the semiconductor device according to the embodiment. [Figure 7] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 8] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 9] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 10] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 11]FIG. 2 is a diagram showing an example of the configuration of a modulation / demodulation unit including a variable impedance element according to the present embodiment. [Figure 12] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 13] FIG. 1 is a diagram showing an example of the configuration of a communication system and an imaging system using the semiconductor device of the present embodiment. [Figure 14] FIG. 2 is a diagram showing an example of an equivalent circuit of the semiconductor device according to the embodiment. [Figure 15] 1A to 1C are diagrams showing application examples of the semiconductor device of the present embodiment. [Figure 16] 1A to 1C are diagrams showing application examples of the semiconductor device of the present embodiment. DETAILED DESCRIPTION OF THE INVENTION

[0009] Hereinafter, embodiments will be described in detail with reference to the accompanying drawings. Note that the following embodiments do not limit the scope of the invention claimed. Although multiple features are described in the embodiments, not all of these multiple features are necessarily essential to the invention, and multiple features may be combined arbitrarily. Furthermore, in the accompanying drawings, the same reference numerals are used to designate the same or similar components, and redundant explanations will be omitted.

[0010] Semiconductor devices according to embodiments of the present disclosure will be described with reference to FIGS. 1 to 16. In the following description, a case where the semiconductor device is used as a transmitter of a communication system (communication device) will be described. However, this is not limiting, and the semiconductor device of this embodiment can also be used as a receiver of a communication system (communication device). Furthermore, in this disclosure, terahertz waves refer to electromagnetic waves in a frequency range of 100 GHz or more and 10 THz or less, more specifically, 300 GHz or more and 3 THz or less. In the description of each embodiment, description of configurations that are the same as those in other embodiments may be omitted. Furthermore, the embodiments can be modified or combined as appropriate.

[0011] A semiconductor device 100 according to a first embodiment will be described with reference to FIGS. 1(a) to 6. FIG. 1(a) is a diagram illustrating an outline of the configuration of the semiconductor device 100 according to this embodiment. FIG. 1(b) is a schematic cross-sectional view of the RTD oscillator 102a of the semiconductor device 100 taken along dashed line A-A' in FIG. 1(a). FIG. 1(c) is a conceptual diagram illustrating the wiring of functional blocks of the semiconductor device 100. As shown in FIG. 1(a), the semiconductor device 100 includes a reference oscillator 101, RTD oscillators 102a and 102b, mixers 103a and 103b, a combiner 104, an antenna 105, and a phase adjuster 106. The semiconductor device 100 is configured to emit terahertz waves TW from the antenna 105. The terahertz waves TW are radio signals (RF signals) generated within the semiconductor device 100 and wirelessly communicated with a receiver (not shown).

[0012] The semiconductor device 100 includes a semiconductor structure for generating terahertz waves TW. More specifically, the semiconductor device 100 includes RTD oscillators 102a and 102b, each of which uses a resonant tunneling diode (RTD) as an oscillator and is combined with a resonant structure. The RTD oscillators in the RTD oscillators 102a and 102b are injection-locked oscillators that oscillate at a terahertz frequency in synchronization with a reference signal supplied from a reference oscillator 101 and output a signal. In other words, the RTDs in the RTD oscillators 102a and 102b are injection-locked oscillators that upconvert at least one of the frequency components of the reference signal supplied from the reference oscillator 101 to a high-frequency component in the terahertz band and output the upconverted signal. The RTD oscillator 102a includes two RTD units 110a and 110b, each of which includes an RTD as an oscillator. Similarly, the RTD oscillator 102b is provided with two RTD units 110a and 110b, each including an RTD as an oscillator. The RTD oscillator 102a and the RTD oscillator 102b may have the same configuration, so the following description will focus on the RTD oscillator 102a, but the RTD oscillator 102b may also have a similar configuration. Furthermore, the terahertz band signal oscillated by the RTD, which is the oscillator in the RTD oscillator 102a or 102b, is referred to as the oscillation signal.

[0013] A reference signal is supplied to the RTD oscillator 102a and the RTD oscillator 102b from a (single) reference oscillator 101, which constitutes a reference signal oscillator that generates a reference signal. The reference signal is output from the reference oscillator 101 and branched into two systems by a branching unit 108. A phase delay of π / 2 (90°) is added to one of the two branched reference signals by a phase adjuster 106, which adjusts the phase of the reference signal. Therefore, the phase of the reference signal supplied to the RTD units 110a and 110b arranged in the RTD oscillator 102a and the RTD units 110a and 110b arranged in the RTD oscillator 102b is shifted by π / 2. The RTDs that are the oscillators of the RTD oscillators 102a and 102b oscillate in the terahertz band and output oscillation signals. The oscillation signal is input to mixers 103a and 103b as local oscillation (LO) signals. Mixers 103a and 103b are each a modulation / demodulation unit, and in this embodiment will be described as a modulation unit that performs modulation. However, this is not limited thereto, and when semiconductor device 100 functions as a receiver as described above, mixers 103a and 103b function as demodulation units. Semiconductor device 100 includes mixer 103a, which is a modulation unit connected to an RTD, which is an oscillator arranged in RTD oscillating unit 102a, which is an injection-locked oscillator. It can also be said that semiconductor device 100 includes mixer 103a, which is a modulation unit connected to an RTD, which is an oscillator arranged in RTD oscillating unit 102a, which is an injection-locked oscillator.

[0014] Data signals 109a and 109b supplied from a baseband circuit (not shown) are modulated by mixing them with an LO signal by a mixer. Data signals 109a and 109b can be different signals depending on the data supplied to the baseband circuit. Here, the data signals are intermediate frequency (IF) signals. Two modulated signals modulated by two mixers 103a and 103b are combined into one signal by a combiner 104. The combined signal is radiated into space by an antenna 105 connected to the combiner 104 as a terahertz wave TW consisting of the modulated signal. The combiner 104 and the antenna 105 form a signal transmission unit for transmitting the modulated signals output from mixers 103a and 103b into space as electromagnetic waves. The arrangement and order of the components in this configuration are one aspect for realizing the present disclosure, and the connection configuration to which the present disclosure can be applied is not limited to this embodiment.

[0015] As shown in FIGS. 1(a) and 1(b), the RTD oscillator 102a includes RTD sections 110a and 110b, each of which includes an RTD, a high-frequency oscillator. The RTD sections 110a and 110b each include a resonant conductor 111 and a GND layer 150 that form a resonant structure for oscillation at a terahertz frequency, along with the RTD, which is an oscillator. A reference signal is supplied to the RTD oscillator 102a from a reference oscillator 101. The oscillation frequency of the RTD sections 110a and 110b is determined by the resonant structure and the inductance and capacitance of the element itself. While the RTD oscillator 102a can self-oscillate at a self-oscillation frequency on its own, the reference oscillator 101 is used as a master signal generator, and the RTD oscillator 102a operates as a slave. As a result, the RTD oscillating unit 102a oscillates in a state where it is injection locked to the oscillation signal of the reference oscillator 101, so the oscillation frequency and phase noise of the slave RTD oscillating unit 102a follow the accuracy of the master reference oscillator 101. In this way, this method makes it possible to stabilize the oscillation frequency of the RTD oscillating unit 102a, which is the LO, and reduce the phase noise. The reference oscillator 101 is a wave source for synchronizing the timing of the oscillation at the terahertz frequency of the RTD oscillating unit 102a. In this way, it can be said that not only the RTD but also the RTD units 110a and 110b that form a resonant structure with the RTD are injection locked oscillators. Therefore, the reference oscillator 101 oscillates at the oscillation frequency f THz In this embodiment, the frequency of the reference signal is equal to the oscillation frequency f of the oscillation signal. THz However, the frequency of the reference signal is not limited to this.

[0016] The power injected from the reference oscillator 101 to the RTD oscillation unit 102a is the output P RTDIt may be equal to or greater than ((3 / 16)cosωτ×ΔIΔV). Here, ω is the angular frequency of the oscillation signal in the terahertz band, and τ is the carrier transit time in the semiconductor layers that make up the RTD units 110a and 110b. Also, ΔI and ΔV are the current difference and voltage difference, respectively, between the current peak and current valley in the negative resistance region of the RTD. There is no limit to the number of RTDs arranged in the RTD oscillating unit 102a, and the sum of the power for the number of RTDs arranged may be injected from the reference oscillator 101 to the RTD oscillating unit 102a. Therefore, the output of the reference oscillator 101 is (number of RTDs arranged in the RTD oscillating unit 102a)×P RTD In addition, the output from the reference oscillator 101 may be greater than the transmission loss from the reference oscillator 101 to each RTD in the RTD oscillating unit 102a. RTD Even if the input signal is smaller than r, locking, which will be described later, is possible. Specifically, even if the input signal is a small signal, about one thousandth of the output of one RTD, the RTD oscillator 102a may synchronize with the signal supplied from the reference oscillator 101. For example, if r=5×10 -4 Injection locking with a small signal of about V is possible. inj / V osc The injection ratio r is expressed as the voltage amplitude (V inj ) and the voltage amplitude of the oscillation signal (V osc ) is the voltage amplitude ratio. It is expected that the effect of reducing phase noise by injection locking will be weakened if the injection ratio r is small, but it is possible to adjust it to the degree of phase noise required for the semiconductor device 100. Therefore, in a circuit to which the present disclosure is applied, P RTD Even when using a reference oscillator 101 having an output smaller than ((3 / 16)cosωτ×ΔIΔV), it is possible to synchronize the RTD oscillation unit 102a while maintaining the frequency accuracy output by the reference oscillator 101. Therefore, the signal strength output by the reference oscillator 101 is P RTD These settings can be made without being largely dependent on the value of . These settings are similar to those for the RTD oscillator 102b.

[0017] The RTD sections 110a and 110b arranged in the RTD oscillator section 102a are supplied with a bias voltage from a bias section (not shown). The bias section supplies power to a bias pattern 112 and supplies the bias voltage to the resonant conductor 111 through a bias supply via 113. The bias pattern 112 is arranged in a dielectric section 151 arranged between the resonant conductor 111 and a GND layer 150. The bias section supplies the power required to drive the RTD sections 110a and 110b and adjusts the bias voltage applied to the RTD sections 110a and 110b. The bias voltage is selected from a voltage that falls within the negative differential resistance region of the RTD and is applied. An AC shunt 115 may also be connected to the bias pattern 112. The AC shunt 115 connects a resistance and a capacitance in series and, by connecting to GND, acts as a filter that shorts out signals in a specific frequency band. The AC shunt 115 can be appropriately arranged in various locations on the bias pattern 112 or in the bias section to suppress parasitic oscillation, which is oscillation at a frequency other than a predetermined frequency in the RTD oscillating section 102a. The AC shunt 115 may be any filter that suppresses parasitic oscillation, and may also have other configurations, such as a DC shunt. In addition, the bias power applied from the bias section to the RTD sections 110a and 110b to drive the RTDs is direct current. Therefore, the bias section and bias pattern 112 may be directly connected, or may be connected via a low-pass filter (LPF) or the like.

[0018] FIG. 1(b) is a schematic diagram of the cross section A-A' of the RTD oscillator 102a in FIG. 1(a). The resonant structure of the RTD oscillator 102a is a microstrip resonator and includes a GND layer 150, which is a wide lower conductor; a resonant conductor 111, which is a line-shaped upper conductor; and a dielectric portion 151 located between the lower and upper conductors. The GND layer 150 is also called a reflector layer. The RTD section 110a is electrically connected between the upper conductor (resonant conductor 111) and the lower conductor (GND layer 150). The RTD section 110a includes a lower electrode 152 in contact with the GND layer 150, which is the lower conductor; an upper electrode 154 in contact with the resonant conductor 111, which is the upper conductor; and an RTD layer 153 disposed between the lower electrode 152 and the upper electrode 154 and forming an RTD. The lower electrode 152 and the upper electrode 154 are formed using conductive materials, such as metal layers, high-carrier-concentration semiconductor layers, and vias, depending on the applicable process. An insulating layer such as a passivation layer 155 for protecting devices such as the RTD section 110a may be disposed on the resonant conductor 111 and the dielectric section 151. As described above, in the dielectric section 151, the bias pattern 112 is disposed between the resonant conductor 111, which is an upper conductor, and the GND layer 150, which is a lower conductor. The bias pattern 112 is connected to the resonant conductor 111 using a via 113 for bias supply. These structures are formed on a semiconductor substrate 156 using various semiconductor processes.

[0019] In the resonator conductor 111, which is the upper conductor, the line length (resonator length) indicated by the arrow 116 in FIG. 1(a) is set to a value equal to or greater than the wavelength λ of the signal oscillated by the RTD oscillating section 102a. eff For λ eff The resonant conductor 111 may be designed based on this line length. This causes the resonant conductor 111 to resonate at a frequency in the terahertz band, and a resonant electric field is generated in the resonant conductor 111 as a standing wave. The resonant electric field is generated in the direction along the arrow 116, with both ends of the resonant conductor 111 forming antinodes (points of maximum amplitude) of the electric field and the center of the resonant conductor 111 forming nodes (points of zero amplitude). In addition, λeff Two RTD units 110a and 110b are arranged on a resonant conductor 111 of 1 / 2 at positions facing each other with a node of the resonant electric field at the center. In this configuration, the two RTD units 110a and 110b oscillate in a push-pull mode. The push-pull mode is an oscillation mode in which the two RTD units 110a and 110b arranged on the resonant conductor 111 oscillate in sync with their phases reversed (out of phase).

[0020] Furthermore, the thickness of the dielectric portion 151 constituting the RTD oscillation portion 102a is λ eff / 10 or more and λ eff The resonant conductor 111 may have an open stub 114 extending in the direction of the arrow 117 shown in FIG. 1(a). One end of the open stub 114 is connected to the center of the resonant conductor 111, and the other end is open (released). The length of the open stub 114 in the direction of the arrow 117 is λ eff / 4, which plays a role in fixing the node of the standing wave oscillating in the resonant conductor 111 to the center of the resonant conductor. In addition, the width of the via 113 for bias supply is smaller than the width of the arrow 116 of the resonant conductor 111.

[0021] Furthermore, the width of the portion (connection portion) of the bias pattern 112 connected to the bias supply via 113 is smaller (thinner) than the width of the arrow 116 of the resonant conductor 111. These widths may be 1 / 10 or less (λ / 10 or less) of the effective wavelength λ of the terahertz-band signal standing in the resonant conductor 111. This is because arranging the bias supply via 113 and the bias pattern 112 in dimensions and positions that do not interfere with the resonant electric field of the resonant conductor 111 is suitable for improving resonance efficiency. Additionally, the bias supply via 113 may be arranged in the center, which is a node of the standing wave of the resonant conductor 111. By positioning the connection point of the bias supply via 113 at the node of the standing wave in the resonant conductor 111, the impedance of the oscillation signal at the connection point is maximized. Therefore, loss due to leakage of the oscillation signal in the resonant conductor 111 from the bias supply via 113 to the bias pattern 112 can be suppressed. The bias section is adjusted appropriately to efficiently synchronize the reference oscillator 101 with the RTD oscillator sections 102a and 102b.

[0022] The RTD oscillator 102a and the RTD oscillator 102b generate oscillation signals synchronized with a reference signal generated by a reference oscillator 101 and phase-adjusted by a phase adjuster 106. Therefore, the LO signals output by the RTD oscillator 102a and the RTD oscillator 102b have a 90° phase difference due to the phase adjustment of the reference signal by the phase adjuster 106. The oscillation signals generated by the RTD oscillators 102a and 102b are supplied as local oscillation signals (LO signals) to the mixers 103a and 103b, but they have a 90° phase difference. Therefore, the signal on the mixer 103a side may be called an In-Phase (I) signal, and the signal on the mixer 103b side may be called a Quadrature-Phase (Q) signal. In this embodiment, to perform 16-quadrature amplitude modulation (16QAM), the data signal 109a supplied to the mixer 103a is modulated by four-level amplitude modulation using the oscillation signal for the I signal. On the other hand, mixer 103b performs modulation by four-level amplitude modulation using the oscillation signal for the Q signal and data signal 109b. The modulated signal for the I signal (RF signal) and the modulated signal for the Q signal (RF signal) are combined in combiner 104 to form a signal with a total of 16 phase and amplitude combinations (4 values ​​x 4 values). The combined modulated signal is radiated and output from antenna 105 as an RF signal.

[0023] The conceptual diagram in FIG. 1(c) is a diagram that schematically illustrates the connections of the above-mentioned functional blocks. The semiconductor device 100 includes a reference oscillator 101, which is a reference signal generator that generates a reference signal, and RTD oscillators 102a and 102b, which are injection-locked oscillators connected to the reference oscillator 101. The reference oscillator 101 is further connected to a branching unit 108 that branches to input a reference signal to the two injection-locked oscillators, and a phase shifter (PS) 106 that adjusts the phases of the two injection-locked oscillators. The reference oscillator 101 also includes signal lines for inputting data signals 109a and 109b to mixers 103a and 103b, which are modulation units that receive the oscillation signals of the two injection-locked oscillators as LO signals. The modulated signal (RF signal) modulated by the modulation unit is then combined by a combining unit 104 that combines multiple RF signals. The combined RF signal is then radiated into space by an antenna 105 that transmits the RF signal into space. A semiconductor device 100 can be configured using such circuit blocks. In this embodiment, the phase adjuster 106 is arranged between the reference oscillator 101, which is a reference signal oscillator, and the RTD oscillator 102b, which is an injection-locked oscillator. However, this is not limiting, and the phase adjuster 106 may be arranged between the reference oscillator 101, which is a reference signal oscillator, and the RTD oscillator 102a, which is an injection-locked oscillator. Furthermore, for example, the phase adjuster 106 may be arranged between the reference oscillator 101 and the RTD oscillator 102a and between the reference oscillator 101 and the RTD oscillator 102b. The phase adjuster 106 may be arranged in any manner that generates a predetermined phase difference (e.g., 90°) between the LO signal output from the RTD oscillator 102a and the LO signal output from the RTD oscillator 102b.

[0024] Next, using Figures 2(a) and 2(b), we will explain the influence of the bias voltage supplied to the RTD units 110a and 110b during injection locking performed by the reference oscillator 101 on the RTD units 110a and 110b, which are oscillators. Figure 2(a) is a graph with the bias voltage applied to the RTD unit 110a on the horizontal axis and the frequency oscillated by the RTD oscillator unit 102a on the vertical axis. The process of inputting a reference signal from the reference oscillator 101 to the RTD units 110a and 110b to stabilize the frequency of the RTD oscillator unit 102a is called injection locking. In this embodiment, the RTD oscillator unit 102a uses the harmonic components of the signal output by the reference oscillator 101. The solid line 201 in the graph plots the correlation between the oscillation frequency and bias voltage in the RTD oscillator unit 102a when there is an input from the reference oscillator 101, i.e., when injection locking is performed. In this embodiment, a reference signal of 0.25 THz, which is a subharmonic frequency half that of 0.5 THz, is supplied from the reference oscillator 101 to the RTD oscillating unit 102a. The dashed line 202 in the graph plots the correlation between the oscillation frequency and bias voltage in the RTD oscillating unit 102a when injection locking is not achieved. The frequency described in this embodiment is an example, and the frequency of the signal output by the reference oscillator 101 may be the same as that of the RTD oscillating unit 102a or may have a harmonic component.

[0025] First, regarding the self-oscillation frequency of the RTD unit 110a, the dashed line 202 in Figure 2(a) indicates a frequency of 0.5 THz when the bias voltage is 0.7 V. It can be seen from the graph that the self-oscillation frequency decreases as the bias voltage decreases and increases as the bias voltage increases. Next, we examine the solid line 201, which indicates injection locking. When the bias voltage is 0.7 V, the frequency is the same as the dashed line 202 (0.5 THz) when injection locking is not performed. However, even when the bias voltage is decreased to 0.69 V, the frequency remains at 0.5 THz. Similarly, even when the bias voltage is increased to 0.71 V, the frequency remains at 0.5 THz. In this way, the oscillation frequency of the RTD oscillation unit 102a is locked by injection locking from the reference oscillator 101. This locking is called "locking." Conversely, the change from a locked state to an unlocked state is called "unlocking." In the graph of Figure 2(a), the locked region is the bias voltage region between 0.69V and 0.71V, which is called the locking range. Within the locking range, the RTD oscillator 102a, which is the injection-locked oscillator, has its oscillation frequency fixed (locked) depending on the frequency of the reference oscillator 101. It is also known that the stronger the signal from the reference oscillator 101 input to the RTD units 110a and 110b, the wider the bias voltage range becomes; conversely, the weaker the signal, the narrower the voltage range becomes. Figure 2(a) illustrates an example in which a 0.01V change causes unlocking. However, the bias voltage value and change amount are not limited to this; they can also be smaller or larger (e.g., 0.1V). These values ​​can be set arbitrarily depending on the RTD, oscillator, and bias unit configuration used.

[0026] FIG. 2(b) is a graph showing the bias voltage applied to the RTD unit 110a on the horizontal axis and the phase difference between the injected reference signal of the reference oscillator 101 and the oscillation signal of the RTD oscillation unit 102a on the vertical axis when injection locking is achieved. When the bias voltage is 0.7V, the phase difference is 0°. This indicates that the injection locking frequency and the self-oscillation frequency match (reference signal frequency f1 × N = oscillation frequency f2 of the RTD oscillation unit 102), so the phases of the signal from the reference oscillator 101 and the signal from the RTD oscillation unit 102a match. When the bias voltage is reduced, the phase difference gradually shifts toward -90°, and when the bias voltage is increased, the phase difference gradually shifts toward +90°. This phenomenon is understood to occur because the frequency difference between the reference signal from the reference oscillator 101 and the self-oscillation signal from the RTD oscillation unit 102a appears as a phase difference when locked, due to a frequency difference between the two signals. This phase difference can be calculated theoretically as φ=sin -1 (Q√(P o / P i )×(f1−f0) / (f1)), where φ is the oscillation phase of the RTD oscillator 102a, and P i is the oscillation power of the RTD oscillator 102a. o is the oscillation power of the harmonic components of the reference signal input from the reference oscillator 101 and synchronized, and Q is the Q value of the RTD oscillation unit 102a. The sharper the frequency spectrum, the higher the Q value, and it is used as an output index of the oscillation circuit. f1 is the self-oscillation frequency, and f0 is the frequency oscillated by injection locking, i.e., the oscillation frequency of the RTD unit 110a. According to a theoretical formula, this phase difference varies within the range of -90° to +90°. In other words, the phase difference within the locking range varies within the range of -90° to +90°, and the width of the locking range varies depending on the ratio of the oscillation power and injection power of the injection-locked oscillator.

[0027] Figures 3(a) to 3(c) are block diagrams showing the details of the circuit blocks of this embodiment. Figure 3(a) shows the components from the reference oscillator 101 to the outputs of the RTD oscillators 102a and 102b. Figure 3(b) shows the periphery of the mixers 103a and 103b and the input section for the data signals 109a and 109b. Figure 3(c) shows the combiner 104 and antenna 105, which are signal transmission sections.

[0028] The reference oscillator 101 outputs a reference signal at a subharmonic frequency of 0.25 THz. For example, a general phase-locked loop (PLL) oscillator circuit can be used for the reference oscillator 101. The PLL oscillator circuit is configured to oscillate, at a desired high frequency, a low-frequency oscillation signal generated by an oscillation source with a lower frequency than the reference signal, such as a high-precision low-frequency oscillation source using a crystal oscillator. The connection port 120 is a connection port between a semiconductor substrate including the reference oscillator 101 and a semiconductor substrate including the RTD oscillators 102a and 102b and their peripheral circuits. For example, the semiconductor substrate including the reference oscillator 101 and the semiconductor substrate including the RTD oscillators 102a and 102b may be at least partially stacked. The connection port 120 may be, for example, a three-terminal connection using a coplanar waveguide. In a coplanar waveguide, two parallel GND lines are arranged on either side of a single signal line, sandwiching the signal line. When the reference oscillator 101 is disposed on a semiconductor substrate separate from the RTD oscillators 102a and 102b, these three wiring patterns (GND line, signal line, GND line) are arranged in parallel and connected to a connection port 120 on the semiconductor substrate that includes the RTD oscillators 102a and 102b. The connection may be made using bumps, such as a flip chip, or may be made using wire bonding from a common interposer substrate. The connection port 120 is provided with three-terminal pads for connection, and a signal input to the substrate is transmitted and converted to a microstrip line. The reference signal is a high-frequency signal of several GHz or more, higher than the signals used in ordinary electrical circuits. Therefore, the connection at the connection port 120 is appropriately made using a circuit configuration used for microwaves, millimeter waves, or the like.

[0029] The phase of the reference signal input from connection port 120 is adjusted, for example, using a 90° hybrid coupler 121. The hybrid coupler 121 is a circuit that functions as the branching unit 108 and phase adjuster 106 in FIG. 1(a). In the hybrid coupler 121, the signal input from port 122 is transmitted to ports 124 and 125. A phase delay of 90° occurs in the path from port 122 to port 124, and a phase delay of 180° occurs in the path from port 122 to port 125. As a result, the phase difference between port 124 and port 125 becomes 90°, and the input reference signal is branched and output to two RTD oscillators 102a and 102b. The signal from port 122 can be considered to be always shorted to GND because the signal on the path from port 122 to port 123 and the signal on the path from port 122 to port 124 to port 125 to port 123 cancel each other out at port 123. Therefore, the reference signal input to hybrid coupler 121 is not output from port 123.

[0030] Here, the configuration of the branching unit 108 and the phase adjuster 106 is not limited to the hybrid coupler 121. FIGS. 4(a) and 4(b) show examples of configurations usable as the phase adjuster 106 and the branching unit 108 other than the hybrid coupler 121. FIG. 4(a) shows an adjustment circuit using line length delay, and FIG. 4(b) shows an example of an adjustment circuit using a polyphase filter. These circuits have three terminals: an input port 400, an output port 401, and an output port 402, and are designed so that when a signal input from the input port 400 is output from the two output ports 401 and 402, the phases of the signals are shifted by 90° from each other.

[0031] In the line delay type adjustment circuit of FIG. 4(a), a signal input from input port 400 is split into two, one of which passes through λ / 4 strip line 403 and is output from output port 401. The other signal passes through λ / 2 strip line 404 and is output from output port 402, where λ is the effective wavelength of the signal as it passes through this line. Because the line lengths of the two strip lines 403 and 404 differ by λ / 4, the phase of the signal at output port 402 is output 90° behind the signal at output port 401. Also, for example, it is possible for a λ / 4 strip line to be provided on one path and no strip line to be provided on the other.

[0032] The adjustment circuit in FIG. 4(b) is a combination of filter circuits called polyphase filters, which use resistors 405 and 408 and capacitances 406 and 407. The circuit from input port 400 to output port 401 is a low-pass filter (LPF) circuit, and the circuit from input port 400 to output port 402 is a high-pass filter (HPF). By utilizing the fact that the output phase of the LPF lags the input signal and the output phase of the HPF leads the input signal, a phase difference of 90° is achieved between output port 401 and output port 402. As described above, phase adjuster 106 to which the present invention is applied may be any circuit that imparts a phase difference to the signals output from the two output ports, and is not limited to this.

[0033] The RTD oscillator 102a and the RTD oscillator 102b are connected to ports 124 and 125, respectively, of the hybrid coupler 121 shown in FIG. 3(a). As described above, the RTD oscillator 102a and the RTD oscillator 102b are injection-locked oscillators that oscillate synchronously at a frequency in the terahertz band using the reference signal supplied from the reference oscillator 101 as a master signal source. Two RTD units, 110a and 110b, are connected to the resonant conductor 111, and oscillate in a push-pull mode in which the phases of the RTD units are reversed by 180°. The RTD oscillator 102b has a similar configuration, but the phase of the reference signal input from the reference oscillator 101 is delayed by 90° from that of the RTD oscillator 102a. The RTD oscillator 102b oscillates in synchronization with the reference signal, so that the phases of the RTD oscillator 102a and the RTD oscillator 102b are orthogonal (they have a phase difference of 90°), and oscillation signals having two terahertz band frequencies are generated. A connection port 126a to the mixer 103a is connected to the right end of the resonant conductor 111 in FIG. 3(a). In other words, the connection port 126a to the mixer 103a is connected to the RTD oscillator 102a, and similarly, the connection port 126b to the mixer 103b is connected to the RTD oscillator 102b.

[0034] Here, the resonant conductor 111 has a T-shape, but it may have another shape and may be configured with a resonant structure including, for example, a resistance R, an inductance L, and a capacitance C that resonates the oscillation signal. Furthermore, the number of RTDs (RTD sections 110a and 110b) connected to the resonant conductor 111 is not limited to two, and may be one, three, or more. When the number of RTDs is small, noise due to RTD variations can be minimized. Furthermore, increasing the number of RTDs can increase the power of the oscillation signal. RTDs capable of oscillating in the terahertz band by generating a negative resistance sufficient to compensate for the resonance of the resonant conductor 111 and the loss in the circuit can be appropriately arranged.

[0035] FIG. 3(b) is a block diagram including peripheral circuits of the mixers 103a and 103b. In this embodiment, a mixer configuration using single-ended mixers (diodes 132a and 132b) is used. The components around the mixer are divided into two systems, one for the I signal and one for the Q signal. In FIG. 3(b), the connection configuration of the elements on the upper side with reference numbers ending in "a" is for the I signal, and the connection configuration of the elements on the lower side with reference numbers ending in "b" is for the Q signal. Hereinafter, including the description of the embodiments described later, the elements for the I signal will be mainly described, since the elements for the Q signal may have similar configurations. In other words, when an element for the Q signal is not described, that element may have the same configuration as the element for the I signal. Furthermore, in the above-mentioned RTD oscillators 102a and 102b, etc., when an element for the Q signal is not described, that element may have the same configuration as the element for the I signal.

[0036] As shown in FIG. 3(b), the periphery of the mixer 103a includes an LO filter 127a, an IF filter 129a, a matching circuit 130a, an RF filter 131a, a diode 132a, and a matching circuit 133a. The connection ports 126a and 126b are connected to the output terminals of the RTD oscillators 102a and 102b shown in FIG. 3(a). The LO signal output by the RTD oscillator 102a connected to the connection port 126a is supplied to the LO filter 127a. The LO filter 127a may be a bandpass filter (BPF) or HPF configured to transmit the LO signal generated by the RTD oscillator 102a and block the data signal 109a. The LO filter 127a may be a filter circuit that transmits only specific frequencies, such as a stripline open stub or a sectorial radial stub, or a wiring pattern with an adjusted inductance L and capacitance C.

[0037] The data signal 109a is supplied from a baseband circuit (not shown) via an IF connection port 128a. As mentioned above, IF stands for Intermediate Frequency and is also called an intermediate frequency. IF is often used in heterodyne systems to remove noise components from the data signal to be modulated. Furthermore, when directly modulating a data signal, such as in a direct conversion system, the data signal can be used directly. The IF signal supplied from the IF connection port 128a is, for example, a 50 GHz data signal, and is passed through an IF filter 129a to be combined with an LO signal. The IF filter 129a functions as an LPF, separating high-frequency signals such as the LO signal from the IF signal line.

[0038] The signal resulting from combining the IF signal and the LO signal is transmitted to a matching circuit 130a disposed between the mixer 103a and a node connected to a signal line carrying the data signal 109a. The matching circuit 130a adjusts the impedance of the line to match the impedance between the peripheral circuit of the mixer 103a and the peripheral circuit of the RTD oscillator 102a. An RF filter 131a is disposed between the matching circuit 130a and the mixer 103a. The RF filter 131a is disposed to suppress degradation of signal quality caused by the RF signal generated by the mixer 103a being reflected back to the LO signal side. The mixer 103a is a single-ended mixer connected to GND via a single diode 132a. A signal is supplied to the anode of the diode 132a, and the cathode is connected to GND. Mixer 103a mixes the input LO signal and IF signal using the nonlinearity of the diode, and performs frequency conversion to an RF signal.

[0039] The RF signal generated by mixer 103a is supplied to mixer output terminal 134a via matching circuit 133a. Matching circuit 133a is a necessary element for adjusting the impedance between mixer 103a and a circuit connected downstream of mixer output terminal 134a, reducing reflected waves toward mixer 103a, and improving the efficiency of RF signal transmission to antenna 105. In this embodiment, a single-ended mixer is used as mixer 103a, but any other element having the configuration of a so-called mixer may be used. Mixer 103a may be configured, for example, as a single-balanced mixer using multiple diodes, a double-balanced mixer, or a bidirectional diode mixer also known as a harmonic mixer. Furthermore, mixers using transistors or the like may also be appropriately arranged, and the configuration of mixer 103a is not limited to the above. Furthermore, the matching circuits and filter circuits around mixer 103a and the connection positions with the IF signal can be swapped as appropriate depending on the circuit structure and the configuration of mixer 103a. The configuration of the Q signal is similar, but the I signal and Q signal can be set to provide different information to the values ​​of the IF signals input from two IF connection ports 128a and 128b. By independently arranging two systems of mixers 103a and 103b and modulating the different signal information of the I signal and Q signal, it is possible to simplify the modulation by the mixers and improve the quality of the modulated signal.

[0040] FIG. 3(c) shows the configuration from mixer output terminals 134a and 134b to antenna 105. The circuit in FIG. 3(c) includes a combiner 104, which is a signal transmission unit, and antenna 105. The combiner 104 is also called a multiplexing circuit. Two types of modulated RF signals output from the two mixers 103a and 103b are expressed as I and Q signals (IQ modulated signals), and are input from mixer output terminals 134a and 134b, respectively. This configuration is called an IQ mixer. The combiner 104 may be, for example, a combiner circuit using a Wilkinson coupler. The combiner 104 has a circuit configuration including a resistor 135 connecting two input terminals and two λ / 4 lines with an output branch located midway between them. The λ / 4 lines of the Wilkinson coupler may be bent as shown in FIG. 3(c) to reduce the circuit area. By using a Wilkinson coupler, the input ports of the IQ modulated signals are appropriately separated, and it is possible to prevent the IQ modulated signal from flowing back to a mixer on the opposite side from the original input. The configuration of the combiner 104 is not limited to a Wilkinson coupler, and other appropriate configurations may be used.

[0041] The antenna 105 may be a combination of a feed line 136 and a patch antenna 118. The RF signal combined by the combiner 104 is supplied to the patch antenna 118 from the feed line 136 (which may also be referred to as a feed line). The antenna 105 is a combination of a GND layer 150 (not shown in FIG. 3(c)) which is a ground conductor, the patch antenna 118, and a dielectric material therebetween. The RF signal input to the antenna 105 is radiated as a terahertz wave TW vertically upward from a surface facing the ground conductor with the patch antenna 118 as a reference. A microstrip antenna or a rectangular patch antenna may be used as the patch antenna 118. The patch antenna 118 is configured such that the effective wavelength λ of the terahertz wave TW on the patch antenna, which is the radiated RF signal, is 1 / 2 GHz. Reff λ in the direction of arrow 119 in FIG. 3(c). ReffThe length may be set to about 1 / 2. By adjusting the position of the connection between the patch antenna 118 and the feed line 136 to an appropriate position inside the patch antenna 118, the impedances of the feed line 136 and the patch antenna 118 are matched, allowing for efficient transmission of RF signals. It is also possible to insert a λ / 4 matching circuit into the feed line 136 to match the input impedance of the patch antenna 118. By integrating the circuits of Figures 3(a), 3(b), and 3(c) on a semiconductor substrate, it is possible to realize a transmission circuit that can perform QAM modulation on an LO signal having a frequency in the terahertz band and transmit it as an RF signal.

[0042] In this embodiment, the semiconductor device 100 is shown as a transmission circuit configuration, but a similar configuration can also be realized as a reception circuit. A signal received from the antenna 105 is input from the opposite side of the combiner 104, which functions as a branching section, and the RF signal and LO signal are input and demodulated by the mixers 103a and 103b, and a data signal is output from the IF connection ports 128a and 128b. The RTD oscillators 102a and 102b function as LO oscillators in the same way as in the oscillator circuit, and a configuration can be realized in which the two mixers 103a and 103b function as IQ mixers. The mixers 103a and 103b can be referred to as demodulation sections (modulation / demodulation sections including transmission), and the combiner 104 can be referred to as a branching section.

[0043] Figures 5(a) to 5(f) are examples of the above-mentioned filter circuits. Figure 5(a) is an example of a microstrip low-pass filter, and Figure 5(b) is a schematic circuit diagram of the filter in Figure 5(a). Figure 5(c) is a microstrip low-pass filter with a different shape from that in Figure 5(a). Figure 5(d) shows a filter using stubs. Figures 5(e) and 5(f) are examples of actual filter circuits.

[0044] FIG. 5(a) shows an example of a low-pass filter (LPF) whose frequency band can be designed by adjusting the width of the microstrip line. The line width between input port 410 and output port 411 is changed. Capacitance sections 412, 414, and 416, which have wide lines, can be regarded as capacitances inserted between the filter circuit and GND. Inductance sections 413 and 415, which have narrow lines, can be regarded as inductances connected in series to the input and output ports. Arrows 421a to 421e shown in FIG. 5(a) indicate the line lengths of the respective lines, λ 1eff It is set at / 8. 1eff represents the effective wavelength of the signal to be blocked by this filter. Figure 5(b) shows an equivalent circuit representation of the capacitance sections 412, 414, and 416 and the inductance sections 413 and 415 in Figure 5(a). Another possible low-pass filter pattern is shown in Figure 5(c). Figure 5(c) shows the line pattern superimposed on the equivalent circuit. Coupling of inductance component L and capacitance component C is considered at various points around the T-shaped section 418 extending from line 417. A filter circuit can be constructed by appropriately designing the line and pattern width. Figure 5(d) shows a filter configuration using open stub 419 and radial stub 420. The length, size, and position of the stubs on the microstrip line from input port 410 to output port 411 can be appropriately set to match the effective wavelength of the signal to be filtered. This allows signals with a specific frequency band to be filtered out from multiple passing signals. In the matching circuits 130a, 130b, 133a, and 133b, by appropriately changing the width of the stubs and patterns, their positions on the lines, etc., it is possible to adjust the impedance for a specific frequency passing through the matching circuit and perform impedance matching between the upstream and downstream stages of the matching circuit.

[0045] The configuration shown in FIG. 5(a) is an example of an IF filter and can be used for the IF filters 129a and 129b shown in FIG. 3(b). An IF signal is input through an input port 410 and output through an output port 411. This circuit is a low-pass filter (LPF), allowing low-frequency IF signals (several GHz to several tens of GHz) to pass while blocking high-frequency signals (LO and RF signals). This suppresses signal quality degradation and increased LO signal loss due to noise and crosstalk in the IF signal circuit. In this embodiment, the capacitance sections 412, 414, and 416 may each have a capacitance of approximately 20 to 40 fF, and the inductance sections 413 and 415 may each have a linewidth set to provide an impedance of approximately 50 to 100 [pH]. These values ​​are appropriately set to achieve a configuration that maintains low impedance in the IF signal band while achieving attenuation of less than one-hundredth (-20 dB) for terahertz-band frequencies such as LO signals. Furthermore, in order to form such a filter on the same semiconductor substrate as the RTD oscillators 102a and 102b, the size of the filter must be such that the inductance and capacitance can be formed by the semiconductor process. LOeff The line length (arrows 421a to 421e) is set to λ LOeff With the above configuration, the filter circuit shown in Fig. 5(a) can be used as an LPF that passes IF signals of 100 GHz or less and blocks LO and RF signals of several hundred GHz or more.

[0046] The filter circuit shown in Fig. 5(e) is a filter configuration according to this embodiment of the LO filter 127a shown in Fig. 3(b). The filter circuit includes a capacitive coupling unit 427, two capacitance units 428 and 430, and an inductance unit 429 disposed between the two capacitance units 428 and 430, between an input port 425 and an output port 426. The capacitive coupling unit 427 performs capacitive coupling (AC coupling) by overlapping the conductors of the microstrip line with an insulator interposed therebetween. This provides DC isolation between the input port 425 and the output port 426.

[0047] This capacitive coupling unit 427 must pass the LO signal and block the IF signal, which is in a low frequency band. The IF signal reaching the RTD oscillator 102a can adversely affect the stabilization of the oscillation of the RTD unit 110a of the RTD oscillator 102a, which is an injection-locked oscillator. Therefore, the strength of the IF signal must be lower than the strength of the oscillation signal oscillated by the RTD oscillator 102a, for example, at least 1 / 100th of that. Therefore, the capacitive coupling unit 427 and the LO filter are configured to increase the impedance in the IF signal frequency band (several GHz to several tens of GHz) by approximately 10 times, or even 100 times, the frequency band in which the RTD oscillator 102a oscillates. Specifically, the capacitance of the capacitive coupling unit 427 may be approximately 20 fF, and it is configured so that it can be considered conductive with no attenuation in the frequency band of the LO signal oscillated by the RTD oscillator 102a (RTD unit 110a). Furthermore, the capacitive coupling portion 427 can also be provided at the connection portion that connects the resonant conductor 111 and the LO filter 127a.

[0048] The capacitance section 428, inductance section 429, and capacitance section 430 of the LO filter 127a, which, together with the capacitive coupling section 427, constitute a bandpass filter (BPF) that performs LC resonance. The LO filter 127a uses a narrow-band filter that passes the LO signal's oscillation frequency of 0.5 THz. The narrow-band LO filter 127a can block multiple frequency signals, including the IF signal supplied as the data signal 109a and the RF signal generated by the mixer 103a. The LO filter 127a thus serves to reduce the strength of unwanted signals returning to the RTD oscillator. For example, by determining the line widths so that the capacitance components of the capacitance sections 428 and 430 are approximately 50 to 100 fF and the inductance component of the inductance section 429 is approximately several pH, a BPF is formed that matches the frequency of the LO signal.

[0049] FIG. 5(f) shows the configuration of the matching circuit 133a shown in FIG. 3(b) in this embodiment. In this embodiment, the matching circuit 133a also includes a wideband bandpass filter (BPF). Capacitive coupling units 433 and 435 are arranged in series between the input port 431 and the output port 432, and short stubs 434 and 436 are arranged between the two capacitive coupling units 433 and 435 and between the capacitive coupling unit 435 and the output port 432, respectively. The capacitive coupling units 433 and 435 are arranged as elements that connect the capacitance component due to capacitive coupling and the inductance component of the line in series. The short stubs 434 and 436 are used as inductance units that have an inductance component between them and GND by adjusting their lengths. Specifically, the matching circuit 133a can be designed to have an optimal inductance component while satisfying a length l<λ / 4, where λ is the effective wavelength of the RF signal passing through the matching circuit 133a. The matching circuit 133a transmits the RF signal output by the mixer 103a to the antenna 105 while filtering out unwanted signals incident from the antenna 105. The matching circuit 133a also removes unwanted harmonic components, such as intermodulation products, generated by the mixer 103a. The matching circuit 133a also removes low-frequency signals, such as IF signals. The matching circuit 133a can also reduce transmission loss by adjusting the impedance between the mixer 103a and the combiner 104. The capacitance components of the capacitive coupling units 433 and 435 are approximately 1 to 30 fF, and the inductance components of the signals flowing through the capacitive coupling units 433 and 435 are approximately several fF to several tens of fF. The inductance components of the short stubs 434 and 436 are approximately 10 to 30 pF and are designed to resonate in the RF frequency band. By connecting multiple similar components, a BPF with a wide passband of several tens of GHz or more can be configured.

[0050] An equivalent circuit of the semiconductor device 100 in this embodiment will be described with reference to FIG. 6. A bias is applied to the RTD oscillators 102a and 102b from a bias unit 600. The RTD oscillator 102a includes two RTD units 110a and 110b, the impedance of the resonant conductor 111 represented by impedance Z4, and the impedance of the RTD units 110a and 110b represented by impedance Z5. The reference oscillator 101 is disposed on the far left in FIG. 6. A transmission line 601 for a reference signal, which is an output signal of the reference oscillator 101, is connected to the reference oscillator 101 to synchronize the RTD oscillators 102a and 102b. The transmission line 601 has a transmission line impedance Z1 along with parasitic inductances L1 and L2, parasitic capacitances C1 and C2, and parasitic resistances R1 and R2.

[0051] A hybrid coupler 121 is connected between the transmission path 601 and the RTD oscillators 102a and 102b. The hybrid coupler 121 can be expressed as having an impedance Z2 connected in series to the terminal connected to the RTD oscillator 102a, and an impedance Z2 and an impedance Z3 connected in series to the terminal connected to the RTD oscillator 102b. The impedance Z3 is set so that the reference signal of the reference oscillator 101, which the hybrid coupler 121 outputs to the RTD oscillator 102b, is delayed by 90° relative to the reference signal output to the RTD oscillator 102a. The hybrid coupler 121 and the RTD oscillator 102a are connected via a capacitance C3. The hybrid coupler 121 and the RTD oscillator 102b are connected via a capacitance C4. The capacitances C3 and C4 are electrically connected by AC coupling to allow the reference signal of the reference oscillator 101 to pass. Specifically, as a bandpass filter for passing a signal of 100 to 100 GHz from reference oscillator 101, it may have a capacitance of about 10 to 100 fF.

[0052] An LO filter 127a having an impedance Z8 is connected between the output side of the RTD oscillator 102a and the IF filter 129a. The IF signal supplied from the IF connection port 128a is connected to the output side of the LO filter 127a via an IF filter 129a having an impedance Z9. Furthermore, the output side of the LO filter 127a is connected to the mixer 103a via a matching circuit 130a having an impedance Z12 and an RF filter 131a having an impedance Z13. The same is true for the output side of the RTD oscillator 102b. The IQ modulated signals modulated by the mixers 103a and 103b are combined by a Wilkinson coupler shown as combiner 104 and radiated from the antenna 105 connected via a feed line 136 shown as impedance Z18.

[0053] The frequency of the reference signal generated by the reference oscillator 101 is set to half the frequency of the LO signal oscillated by the RTD oscillator 102a, and the capacitances C3 and C4 can be configured to pass the frequency of the reference signal and block the frequency of the LO signal. Specifically, the capacitances C3 and C4 can have a capacitance of approximately tens to hundreds of fF as a bandpass filter for passing signals from the reference oscillator in the range of 100 to several hundred GHz, as described above. The capacitances C3 and C4 can be realized by using an AC coupling unit in which an insulator is arranged between the conductor pattern of the signal line connected to the output port of the hybrid coupler 121 and the resonant conductor 111 of the RTD oscillator 102a. The impedances Z8 to Z15 are circuits that serve as filters and impedance matching and are arranged on the signal lines connecting the RTD oscillators 102a and 102b to the mixers 103a and 103b. These filters are arranged to efficiently mix the RF signals output by mixers 103a and 103b, the IF data signals, and the LO signal input to the mixers from the RTD oscillators. Various filters, such as capacitance, fan-shaped radial stubs, and microstrip lines, are used to pass desired frequencies and block unwanted frequencies. Furthermore, a planar antenna such as a patch antenna is used for antenna 105, and a signal line (feed line 136) is connected at an inset from the end of the patch antenna to match impedance with the line. The RF signals output from RTD oscillators 102a and 102b and modulated by mixers 103a and 103b form standing waves on patch antenna 118 of antenna 105 and are radiated into space. A filter circuit or other suitable circuit may be arranged on the stripline connecting antenna 105 and feed line 136.

[0054] Furthermore, a filter circuit or the like may be appropriately arranged around the RTD oscillation unit 102a. Here, the wavelength of the reference signal output by the reference oscillator 101 is λ inj , the wavelength of the oscillation signal generated by the RTD oscillation unit 102a is λ RTD A λ oscillating filter used in the high frequency circuit is provided on the input side of the RTD oscillator 102a as a filter circuit.inj A short stub of λ / 4 may be provided. By providing an appropriate short stub at the input end of the RTD oscillator 102a, the frequency of the reference signal is ignored, allowing only the reference signal to be transmitted. On the other hand, the LO signal having twice the frequency of the reference signal oscillated by the RTD oscillator 102a is inj The length of / 4 is λ RTD Therefore, the LO signal is always connected to GND. RTD Therefore, it is possible to prevent the signal from returning to the reference oscillator 101 side. Furthermore, a λ inj An open stub of 1 / 4 may be arranged. The open stub operates in the opposite manner to the short stub, and functions as a filter that cannot transmit the reference signal frequency but can transmit the LO signal, which is twice the frequency of the reference signal. The stub filter configuration can be based on technology used in microwave and millimeter-wave circuits, so it can be scaled and arranged appropriately to match the terahertz band frequency.

[0055] The reference oscillator 101, mixers 103a and 103b, antenna 105, and associated peripheral circuits of the semiconductor device 100 described in this embodiment do not necessarily need to be arranged on the same semiconductor substrate. For example, a semiconductor device can be configured in which multiple components are integrated on separate semiconductor substrates. In this case, the semiconductor device 100 can be realized by connecting multiple semiconductor substrates using wire bonding, flip chip, or the like during the semiconductor substrate mounting process. The semiconductor device 100 can also be manufactured by bonding and combining semiconductor substrates in a semiconductor process, or by integrating multiple semiconductor substrates into a single package. Furthermore, the semiconductor device 100 may employ an Antenna In Package (AiP) configuration in which an antenna and semiconductor substrates are integrated into a single package.

[0056] A balun, a conversion circuit, a matching circuit, etc. may be appropriately arranged at the boundary between multiple semiconductor substrates. Furthermore, in the above-described configuration, no active elements are arranged in the path between the RTD units 110a and 110b (RTD oscillators 102a and 102b) serving as injection-locked oscillators and the mixers 103a and 103b serving as modulation units. However, this is not limiting. In addition to the above-described components, the peripheral circuits may also include amplifier circuits such as a power amplifier (PA) or a low-noise amplifier (LNA), phase adjustment circuits such as a phase shifter, and switches. The semiconductor substrate may be based on silicon or a compound semiconductor such as indium phosphide (InP) or gallium arsenide (GaAs). For example, the reference signal oscillator (reference oscillator 101) may be formed on a silicon substrate, and the subsequent stages (the antenna 105 side) from the connection port 120 shown in FIG. 3(a) onward may be formed on a compound semiconductor substrate. In this case, a silicon substrate on which the reference oscillator 101 and other components are formed and a compound semiconductor substrate on which the RTD oscillators 102a and 102b and other components are formed may be stacked to form the semiconductor device 100. However, the configuration and implementation of the semiconductor substrate of the semiconductor device 100 to which the present disclosure can be applied are not limited to those described above. In either case, it is sufficient to impart a phase difference to the output signals (oscillation signals) of the RTD oscillators 102a and 102b, which operate as injection-locked oscillators using the reference signal from the reference oscillator 101. In addition, by modulating the RTD oscillators 102 and 102 with multiple data signals according to their respective phase differences, it is possible to configure a semiconductor device 100 to which the present disclosure can be applied, which can realize a transceiver circuit using the RTD oscillators 102 and 103.

[0057] In this embodiment, the RTD oscillators 102a and 102b are used to efficiently generate terahertz wave signals in a small area. Furthermore, by locating the RTD oscillators 102a and 102b near circuits such as the mixers 103a and 103b, losses in transmission lines can be minimized. The RTD oscillators 102a and 102b synchronize the I and Q signals with a 90° phase difference, outputting the I and Q signals. Modulated signals are generated by modulating the I and Q signals with data signals, respectively. Multiple modulated signals can be combined and radiated from the antenna 105. This configuration eliminates the need for phase modulation by the mixers 103a and 103b, even for multilevel modulation such as QAM. This configuration reduces the effects of phase shifts due to mixer characteristics and transmission path characteristics, thereby improving signal quality through compactness and improved high-frequency characteristics.

[0058] Next, a semiconductor device 700 according to a second embodiment of the present disclosure will be described with reference to FIGS. 7(a) to 7(c), 11, and 14. FIG. 7(a) is a circuit block diagram of the semiconductor device 700 according to this embodiment, and FIG. 7(b) is a schematic diagram of the cross-sectional structure taken along line A-A' in FIG. 7(a). FIG. 7(c) shows a conceptual diagram of the circuit block of the semiconductor device 700. Unlike the first embodiment described above, the semiconductor device 700 does not have a mixer. Also, the configuration of bias patterns 161a and 161b for supplying bias to the RTD oscillating units 160a and 160b is different from that of the first embodiment.

[0059] As shown in FIG. 7(a), two bias patterns 161a and 161b are independently arranged to supply bias to the RTD oscillator 160a and the RTD oscillator 160b, respectively. A data signal line 162 is electrically connected to the bias pattern 161a via an AC coupling unit 163. As shown in FIG. 7(b), the AC coupling unit 163 may have a metal-insulator-metal (MIM) structure. Specifically, a portion of the dielectric unit 151 is disposed between the conductor pattern of the data signal line 162 and the bias pattern 161a, thereby connecting the data signal line 162 and the bias pattern 161a via capacitive coupling. The data signal line 162 is connected to the signal line of the data signal 109a at a position on the substrate (not shown), and a data signal for modulation is supplied to the bias pattern 161a. Specifically, the bias supplied to the RTD oscillation unit 160a is supplied as a direct current (DC) component from the bias unit through a bias pattern 161a, and the data signal 109a is supplied as an alternating current (AC) component through a data signal line 162.

[0060] Here, the capacitance of the AC coupling unit 163 is approximately several tens to several hundreds of fF and is configured to pass data signals of several GHz or higher. This capacitance can be set to a value that can sufficiently ensure the bandwidth of the data signal relative to the low impedance of the bias pattern 161a while sufficiently eliminating crosstalk due to unnecessary low-frequency components in the bias unit or data signal 109a. In the configuration shown in FIG. 7(b), the data signal line 162 is arranged in the dielectric unit 151 on a layer above the bias pattern 161a using a microstrip line or a coplanar line. However, this is not limited to this. The data signal line 162 may be arranged on the same metal layer as the resonant conductor 111 and connected to a conductor pattern arranged in the AC coupling unit 163 using a via. Furthermore, the AC coupling unit 163 does not have to have the MIM structure shown in FIG. 7(b). For example, in the layer where the bias pattern 161a is arranged, the AC coupling unit 163 can also form capacitive coupling by reducing the distance between the bias pattern 161a and the data signal line 162. Therefore, superimposing the AC component of the data signal line 162 on the bias voltage of the bias pattern 161a can be achieved using various methods and is not limited to the configuration of the present disclosure.

[0061] Changes in the AC component of the voltage (hereinafter sometimes referred to as bias voltage) in the bias pattern 161a change the amplitude of the signal oscillated by the RTD oscillator 160a, which is injection-locked to the reference signal of the reference oscillator 101. This causes the RTD oscillator 160a to function as a modulation section (modulation / demodulation section) that performs amplitude modulation. The RTD sections 110a and 110b in the RTD oscillator 160a have the characteristic that the amplitude of the signal they output changes depending on the bias voltage applied. The RTD oscillator 160b oscillates in synchronization with a reference signal that is delayed by a phase of π / 2 from the reference signal injected into the RTD oscillator 160a by the phase adjuster 106, and generates an oscillation signal that is orthogonal to the oscillation signal of the RTD oscillator 160a. The RTD oscillator 160b has the same bias pattern and data signal line configuration as the RTD oscillator 160a, and outputs a modulated signal that is amplitude-modulated using the data signal 109b. The modulation section of this embodiment shows a form including the characteristics of the AC coupling section 163 provided on the bias patterns 161a and 161b, and the RTD oscillation section whose signal output, such as amplitude, changes depending on the bias voltage.

[0062] The RTD oscillation units 160a and 160b synchronize with the reference signal supplied from the reference oscillator 101 and modulate in response to changes in the bias voltage of the bias patterns 161a and 161b. Therefore, the injection power of the reference signal supplied from the reference oscillator 101 must be strong enough to prevent the lock from being lost due to modulation. Specifically, in FIG. 2(a), the bias voltage in the locking range was 0.69 [V] to 0.71 [V]. Therefore, the amplitude of the data signal is set so that the amplitude of the bias voltage on which the data signal is superimposed is 0.7 ± 0.01 [V]. This locking range can be controlled by the injection power, so it is necessary to appropriately set the locking range of the reference signal and the amplitude of the data signal.

[0063] The modulation of the oscillation signals of the RTD oscillators 160a and 160b using the data signals 109a and 109b is not limited to this embodiment. A typical bias-T circuit can also be formed and connected to the bias unit and bias patterns 161a and 161b. In the bias-T circuit, the direct current (DC) side is connected to a bias power supply, the alternating current (AC) side is connected to the data signal, and the AC / DC output side is connected to the RTD oscillator. The data signal line 162 may also be electrically connected directly to the resonant conductor 111, rather than to the bias pattern 161a. When connecting the data signal line 162 to the resonant conductor 111, the modulation signal can be transmitted by providing an AC coupling unit between the conductor pattern of the resonant conductor 111 and the conductor pattern of the data signal line 162. In this case, the AC coupling unit may be connected in a position that creates different impedances or in a connection configuration that has a capacitance component in two frequency bands: the LO frequency band in which the RTD oscillator 102a oscillates and the IF frequency band of the data signal. Furthermore, modulation is also possible using an impedance adjustment unit, such as by using a connected data signal line 162, to change the impedance of the resonant conductor 111. The impedance adjustment unit uses an impedance variable element, such as a varactor diode, to control the impedance from an external data signal 109a, thereby changing the impedance of the resonant conductor 111.

[0064] FIG. 11 shows a block diagram of a modulation unit 1100 (which may also be a demodulation unit or modulation / demodulation unit) when an impedance adjustment unit using a variable impedance element is used. The modulation unit 1100 includes impedances Z20 and Z21, which are λ / 4 lines corresponding to the terahertz-band signal wavelength oscillated by the RTD oscillation unit 160a, connected in series. The terminal of impedance Z20 opposite the terminal connected to impedance Z21 is open and functions as an open stub. The terminal of impedance Z21 opposite the terminal connected to impedance Z20 is connected to the resonant conductor 111 via AC coupling (capacitive coupling). A variable impedance circuit 1107 is connected to the connection between Z20 and Z21. In the variable impedance circuit 1107, a capacitor C20, an impedance Z22, and a varactor diode Cv, which is a variable capacitance, are connected between GND and a node connecting impedance Z20 and impedance Z21. In order to adjust the capacitance of the varactor diode Cv, the signal line that supplies the data signal 109a passes through an inductor L20 and is connected to the node that connects the impedance Z22 and the varactor diode Cv.

[0065] The modulation unit 1100 is connected to the RTD oscillator 160a and can adjust the impedance of the resonant conductor 111. Changing the impedance of the resonant conductor 111 changes the frequency, phase, and intensity of the oscillation of the RTD oscillator 160a, allowing the modulation unit 1100 to perform modulation. A variable impedance element such as a variable resistor can be used as the variable capacitance of the varactor diode Cv, and a variable impedance circuit using a transistor can also be used as the modulation / demodulation unit of the RTD oscillator. In either configuration, the oscillation signal of the RTD oscillator 160a can be modulated from the data signal 109a. Furthermore, modulation of the RTD oscillators 160a and 160b can be achieved not only by amplitude, but also by frequency or phase modulation. In either method, the oscillation signals generated by the RTD oscillators 160a and 160b can be modulated by inputting and modulating the data signals 109a and 109b to the RTD oscillators 160a and 160b. Other circuit configurations may be the same as those in the first embodiment described above, and therefore will not be described here.

[0066] The conceptual diagram of FIG. 7(c) schematically illustrates functional blocks of this embodiment. In the semiconductor device 700, modulation units 701a and 701b are arranged to include two injection-locked oscillators. The modulation units 701a and 701b are connected to signal lines that input different data signals 109a and 109b, respectively, and are configured to modulate and output signals oscillated by the RTD units 110a and 110b, which are injection-locked oscillators. The RF signals modulated by the multiple modulation units 701a and 701b are combined by a combiner 104 that combines multiple RF signals, and the RF signal combined by the combiner 104 is radiated into space by an antenna 105 that radiates electromagnetic waves into space. In this embodiment, the semiconductor device 700 can be configured using such circuit blocks.

[0067] An example of the circuit configuration around the RTD oscillator in this embodiment will be described using the equivalent circuit in FIG. 14. The connection port 120 is a three-terminal connection terminal. The central pad 170 is a signal line connection terminal, and GND line pads 711 and 712 that determine the reference voltage are arranged above and below it. A hybrid coupler 121 is connected to the pad 170 via a capacitor C30. The capacitor C30 is arranged to remove DC components from the reference signal supplied from the reference oscillator 101. An RTD oscillator 713a is connected to port 124 of the hybrid coupler 121, and an RTD oscillator 713b is connected to port 125. A patch antenna 716 is connected to the terminals of the two RTD oscillators 713a and 713b opposite the hybrid coupler 121. The patch antenna 716 also serves as a combiner that combines signals input from the two RTD oscillators 713a and 713b on the patch antenna. The combined RF signal is radiated into space by the patch antenna 716 .

[0068] The two RTD oscillators 713a and 713b each include two RTD sections 717a and 718b. In the RTD oscillator 713a, the RTD section 717a, stripline 714a, resistor R30, stripline 715a, and RTD section 718a are connected to the connection terminal of the hybrid coupler 121. One terminal of each of the two RTD sections 717a and 718a and resistor R30 is connected to GND. The striplines 714a and 715a function as resonant conductors 111, and the overall frequency of the RTD oscillator 713a is determined by resonating the oscillation signals generated by the RTD sections 717a and 718a, which act as oscillators. The resistor R30 has a DC shunt structure and functions to suppress unwanted parasitic oscillations in the RTD oscillator 713a caused by the RTD sections 717a and 718a. The RTD oscillator 713b has a similar structure.

[0069] In the RTD oscillator 713a, a signal line 719a is connected to the node connecting the resistor R30 and the striplines 714a and 715a. The signal line 719a is a signal line in which a bias DC component is superimposed on a data signal. The signal line 719a is connected to a connection port 721a via a stripline 720a. The connection port 721a is connected to another substrate by a three-terminal pad, similar to the connection port 120.

[0070] Similarly, in the RTD oscillator 713b, a signal line 719b is connected to the node connecting resistor R31 and striplines 714b and 715b. Unlike the RTD oscillator 713a, no DC bias component is superimposed on the signal line 719b; only the AC component of the data signal is supplied. The DC component of the signal input from connection port 721b is filtered using capacitor C31, which is arranged between stripline 720b and connection port 721b. In the RTD oscillator 713a, the signal input from signal line 719a to the RTD oscillator 713a is modulated by the nonlinearity of the RTD units 717a and 718a, converted into an RF signal, and output. At the same time, bias for the RTD units 717a and 718a is also supplied from signal line 719a. In addition, in the RTD oscillator 713b, a signal input from a signal line 719b to the RTD oscillator 713b is modulated by the nonlinearity of the RTD units 717b and 718b, converted into an RF signal, and output. The bias for the RTD units 717b and 718b is supplied from a signal line 719a.

[0071] While FIG. 14 illustrates an example of a transmitter circuit envisioned by this embodiment, the transmitter configuration to which the present invention is applied is not limited to this. The hybrid coupler 121 may be a stripline or rat-race circuit that provides different phase differences between the RTD oscillator 713a and the RTD oscillator 713b. The number of RTDs may be one or three or more. The more RTDs there are, the stronger the oscillation intensity becomes, but it is necessary to address issues such as power consumption, inter-element variation, and suppression of parasitic oscillation. The patch antenna 716 is not limited to a rectangular patch antenna; slot antennas, dipole antennas, loop antennas, and various other planar antennas can also be used. Resistors R30 and R31 form a DC shunt, but an AC shunt structure in which a resistor and a capacitor are connected in series may also be used. While signal line 719a simultaneously supplies a data signal and a bias to the RTD oscillator 713a, the data signal and bias may be separately supplied to the RTD oscillator 713a. Furthermore, although mixing is performed using the RTD oscillator 713a as a mixer, other modulation methods may be used in which a data signal is superimposed on the LO signal of the RTD oscillator 713a. In either case, a terahertz wave transmitter (receiver or transceiver) can be fabricated using this embodiment.

[0072] The semiconductor device 700 of this embodiment can integrate the RTD section (RTD oscillation section), which is an injection-locked oscillator, and the modulation section. This allows the semiconductor device 700 to reduce the scale of the terahertz wave generation and modulation circuits. Although this embodiment requires multiple RTD oscillation sections and modulation sections, the reduced circuit scale allows the semiconductor device 700 to be miniaturized.

[0073] Next, a semiconductor device 800 according to a third embodiment of the present disclosure will be described with reference to FIGS. 8(a) and 8(b). In this embodiment, the semiconductor device 800 includes a plurality of reference oscillators in its reference signal oscillator section. FIG. 8(a) is a schematic diagram of the circuit blocks of the semiconductor device 800, and FIG. 8(b) is a conceptual diagram showing the wiring of the functional blocks of the semiconductor device 800. The reference signal oscillator section of the semiconductor device 800 includes a plurality of reference oscillators 801 and 802. The reference oscillator 801 is an oscillator for supplying a reference signal to the RTD oscillator section 102a. The reference oscillator 802 is an oscillator for supplying a reference signal to the RTD oscillator section 102b. The reference oscillators 801 and 802 are controlled to oscillate at the same frequency. The two reference oscillators 801 and 802 are connected using a synchronization line 803, thereby synchronizing the phases of their oscillation signals. Reference oscillator 802 generates a reference signal to be supplied to RTD oscillating unit 102b based on the signal output by reference oscillator 801 (reference signal supplied to RTD oscillating unit 102a). The reference signal output from reference oscillator 802 is delayed in phase by π / 2 (90°) by phase adjuster 106. As a result, the reference signals input to RTD oscillating unit 102a and RTD oscillating unit 102b are shifted by π / 2. Therefore, RTD oscillating unit 102a and RTD oscillating unit 102b can oscillate at frequencies in the terahertz band in quadrature.

[0074] The synchronization line 803 directly connects the reference oscillator 801 and the reference oscillator 802 as shown in FIGS. 8(a) and 8(b). However, this is not a limitation, and the synchronization line 803 may be a signal line commonly routed from a control circuit that controls the two reference oscillators 801 and 802. As long as the reference oscillators 801 and 802 can oscillate reference signals at the same frequency and phase, the configurations shown in FIGS. 8(a) and 8(b) are not required. Alternatively, instead of the phase adjuster 106, the synchronization line 803 may be used to shift the phases of the reference oscillators 801 and 802, thereby adjusting the phase difference between the reference signals. In this case, the phase adjuster 106 is disposed on the synchronization line 803. These configurations are for generating reference signals that allow the RTD oscillators 102a and 102b to oscillate in quadrature, and are not limited to those described in this embodiment.

[0075] The conceptual diagram of FIG. 8(b) schematically illustrates the above-described functional blocks. Unlike the above-described embodiments, the semiconductor device 800 includes a reference oscillator 801 and a reference oscillator 802, which are connected and synchronized by a synchronization line 803. The semiconductor device 800 includes an RTD oscillating unit 102a (RTD units 110a and 110b) that is an injection-locked oscillator connected to the reference oscillator 801, and an RTD oscillating unit 102b (RTD units 110a and 110b) that is an injection-locked oscillator connected to the reference oscillator 802. Furthermore, a phase adjuster 106 that adjusts the phases of the two injection-locked oscillators is connected between the reference oscillator 802 and the RTD oscillating unit 102b. In this manner, two injection-locked oscillators are configured to generate two systems of oscillation signals with different phases, and the respective oscillation signals are input to mixers 103a and 103b to generate RF signals. In this embodiment, a semiconductor device 800 can be configured using such circuit blocks.

[0076] In the semiconductor device 800 of this embodiment, the reference oscillator 801 and the reference oscillator 802 can be arranged separately, which facilitates optimization of the circuit layout. Furthermore, since there is no need to branch the output of the reference oscillator, it is possible to transmit large power with less loss than in the above-described embodiments to the RTD oscillation units 102a and 102b.

[0077] Next, a semiconductor device 900 according to a fourth embodiment of the present disclosure will be described with reference to Figures 9(a) and 9(b). In the above-described embodiments, phase adjustment is performed on signals supplied to the RTD oscillators 102a and 102b, but in this embodiment, phase adjustment is performed on a signal output from the RTD oscillator 102b. Figure 9(a) is a schematic diagram of the circuit blocks of the semiconductor device 900, and Figure 9(b) is a conceptual diagram showing the wiring of the functional blocks of the semiconductor device 900.

[0078] The reference signal output from reference oscillator 101 is branched by branching unit 108 and supplied to RTD oscillators 102a and 102b (RTD units 110a and 110b), which are injection-locked oscillators. RTD oscillators 102a and 102b oscillate in synchronization with the reference signal, and therefore oscillate at the same frequency and phase in the terahertz band. The signal oscillated by RTD oscillator 102a is input as is to mixer 103a as an LO signal. The signal oscillated by RTD oscillator 102b has its phase adjusted by phase adjuster 106 and is then input to mixer 103b as an LO signal. The LO signal input to mixer 103a and mixer 103b is shifted by π / 2 by adjustment of phase adjuster 106, and therefore the two mixers 103a and 103b operate as mixers that respectively modulate the orthogonal LO signals.

[0079] The conceptual diagram of FIG. 9(b) schematically illustrates the functional blocks described above. Unlike the above-described embodiments, the semiconductor device 900 supplies the output of the reference oscillator 101 in phase to two injection-locked oscillators, namely, RTD oscillators 102a and 102b (RTD units 110a and 110b). The injection-locked oscillators oscillate terahertz-band oscillation signals in phase with the reference signal. The output of the RTD oscillator 102a is supplied directly to the mixer 103a, and the output of the RTD oscillator 102b is phase-adjusted by the phase adjuster 106 and then supplied to the mixer 103b. LO signals with different phases are modulated by the mixers 103a and 103b, and then the same operations as those described in the above-described embodiments are performed. In this embodiment, the semiconductor device 900 can be configured using such circuit blocks.

[0080] In this embodiment, the RTD oscillating unit 102a and the RTD oscillating unit 102b can be made to oscillate in the same phase. Furthermore, the transmission path of the reference signal for injection locking from the reference oscillator 101 to the RTD oscillating units 102a and 102b can be simplified. As a result, the signal quality of the signals generated by the RTD oscillating units 102a and 102b is improved, and the quality of the reference signal is improved by reducing loss in the line from the reference oscillator 101 to the RTD oscillating units 102a and 102b.

[0081] Next, a semiconductor device 1000 according to a fifth embodiment of the present disclosure will be described with reference to FIGS. 10(a) and 10(b). The semiconductor device 1000 of this embodiment is configured to combine modulated signals not on a circuit but in the space where the signals are emitted. FIG. 10(a) is a schematic diagram of the circuit blocks of the semiconductor device 1000, and FIG. 10(b) is a conceptual diagram showing the wiring of the functional blocks of the semiconductor device 1000. In this embodiment, the signal transmission unit of the semiconductor device 1000 includes an antenna 1001a connected to a mixer 103a, which is a modulation unit, and which emits a signal output from the mixer 103a, and an antenna 1001b connected to a mixer 103b, which is a modulation unit, and which emits a signal output from the mixer 103b. The two mixers 103a and 103b generate modulated signals by modulating LO signals with data signals 109a and 109b. The modulated signal from mixer 103a is then supplied to antenna 1001a, which radiates the terahertz wave TWa into space, and the modulated signal from mixer 103b is supplied to antenna 1001b, which radiates the terahertz wave TWb into space.

[0082] Antennas 1001a and 1001b are arranged close to each other on the same substrate. The distance between the antennas may be equal to or less than the wavelength of the signals emitted by RTD units 110a and 110b, which are injection-locked oscillators, or even equal to or less than half the wavelength. Terahertz waves TWa and TWb emitted from antennas 1001a and 1001b, which are close to each other, are combined as they propagate through the same space to become a combined terahertz wave TWm. At this time, terahertz waves TWa and TWb are combined in the near-field (Fresnel region) of antennas 1001a and 1001b, and propagate as a combined terahertz wave TWm in the far-field region (Fraunhofer region). A receiving unit (not shown) can receive and demodulate this combined terahertz wave TWm. Furthermore, the phase difference between multiple antennas can be adjusted more flexibly by using a phase adjuster, in which case beamforming can be achieved by changing the beam direction of the radiated signal. For RF signals with two orthogonal phases, increasing the phase delay of one of the RTD oscillators makes it possible to tilt the plane on which the orthogonal phases appear. This makes it possible to support beamforming by tilting the beam in the direction of the antenna radiating the phase-delayed signal.

[0083] The conceptual diagram of FIG. 10(b) schematically illustrates the above-described functional blocks. Unlike the above-described embodiments, the semiconductor device 1000 has mixers 103a and 103b, which are modulation units, connected to separate antennas 1001a and 1001b, respectively. The combining unit 104 is positioned not between the mixers and the antennas, but in the space of the terahertz waves emitted by the two antennas 1001a and 1001b. This generates a combined RF signal (combined terahertz wave TWm) modulated by two injection-locked oscillators with different phases. In this embodiment, the semiconductor device 1000 can be configured using such circuit blocks.

[0084] In this embodiment, the combiner 104 shown in FIG. 1(a) is not required on the circuit, improving the layout flexibility of the circuit from the mixers 103a and 103b to the antennas 1001a and 1001b. Furthermore, by directly connecting the outputs of the mixers 103a and 103b to the antennas 1001a and 1001b and radiating them, RF signal loss in the circuit can be reduced. Because the RF signals output from the mixers 103a and 103b are broadband, the number of components used from the mixers 103a and 103b to the antennas 1001a and 1001b can be minimized, allowing for efficient placement of only broadband components. As the frequency of the terahertz-band antennas 1001a and 1001b increases, the antenna size becomes smaller, and the modulation section may become larger. Therefore, this embodiment can be selected based on the circuit size and wiring flexibility.

[0085] Next, a semiconductor device 1200 according to a sixth embodiment of the present disclosure will be described with reference to FIG. 12. The semiconductor device 1200 according to this embodiment has a configuration including more than two RTD oscillators. FIG. 12 is a block diagram illustrating an outline of the semiconductor device 1200 according to this embodiment. A reference signal output from a reference oscillator 101 is branched into four signals by a branching unit 1201. The four branched reference signals are supplied to the four RTD oscillators 102a to 102d, respectively. The reference signal supplied to the RTD oscillator 102a is input directly to the RTD oscillator 102a without phase adjustment. The reference signal supplied to the RTD oscillator 102b is input to the RTD oscillator 102b via a phase adjuster 1202a, which applies a delay of π / 2 to the reference signal. The reference signal supplied to the RTD oscillator 102c is input to the RTD oscillator 102c via a phase adjuster 1202b, which applies a delay of π to the reference signal. The reference signal supplied to the RTD oscillator 102d is input to the RTD oscillator 102d via a phase adjuster 1202c, which delays the reference signal by 3π / 2. The frequencies of the RTD oscillators 102a, 102b, 102c, and 102d are equal to each other because they are injection-locked to the reference signal supplied from the reference oscillator 101, and the phases are shifted by π / 2 to oscillate. The oscillation signals output from the RTD oscillators 102a to 102d are input to the mixers 103a to 103d, respectively, as LO signals. Furthermore, the mixers 103a to 103d are supplied with four types of data signals 109a to 109d, and output RF signals obtained by modulating the oscillation signal and the data signal. The outputs of mixers 103a and 103b are combined in a combiner 104a, and the outputs of mixers 103c and 103d are combined in a combiner 104b. Furthermore, the outputs of combiners 104a and 104b are combined in combiner 104c and radiated from antenna 105 as a terahertz transmission signal.

[0086] In this embodiment, the phase adjustments performed by the phase adjusters 1202a to 1202c are all performed with different shift amounts, but the same value can also be used for multiple phase adjusters. Since the signal strength of the terahertz waves can be increased by arranging multiple RTD oscillators 102a to 102d, it is possible to input the terahertz waves to multiple RTD oscillators with the same phase, for example, by performing no phase adjustment on two RTD oscillators and performing phase adjustment with a π / 2 delay on the other two RTD oscillators. Furthermore, as described in the third embodiment, multiple reference oscillators 101 may be provided, and reference signals may be supplied from the respective reference oscillators 101 to the RTD oscillators. Furthermore, in this embodiment, four different data signals 109a to 109d are respectively supplied to the mixers 103a to 103d. However, a redundant configuration is also possible, for example, by inputting the same data or inverted data as the data signal 109a and the data signal 109c and modulating them. Because the RTD oscillator 102a with no phase delay and the RTD oscillator 102c with a phase delay of π are out of phase by 180 degrees, it is possible to invert and modulate one of the data signals, and then invert and combine one of the modulated RF signals. To minimize the impact of changes in the data signal, modulation is performed using both positive (+) and negative (-) data transitions, which makes it possible to stabilize the circuit current and reduce crosstalk.

[0087] Furthermore, instead of modulation using the mixers 103a to 103d, modulation can also be performed using a switch that switches the signal output on and off. By switching the signal output on and off using a data signal, phase modulation, which switches between four phases and outputs, is possible. Furthermore, modulation can be performed using an LO signal from a common RTD oscillator in multiple modulators. Furthermore, as in the fifth embodiment, multiple antennas can be provided for the antenna 105 and combiners 104a to 104c, and some or all of the RF signal combinations can be performed in space. This embodiment demonstrates that the configuration of the RTD oscillator and modulator is not limited to two configurations, and that multiple redundant and efficient distribution and combinations are possible using the circuit of the present disclosure. In either case, complex modulation schemes can be realized by generating oscillation signals with different phases using multiple injection-locked oscillators, modulating them with different data signals, and combining them.

[0088] In this embodiment, it is possible to reduce the variation in values ​​that can be taken by a modulation unit such as a mixer. This simplifies the configuration of the modulation unit, and improvement in signal quality can be expected. In addition, since it is possible to arrange multiple RTD oscillators for one antenna, an improvement in the signal-noise ratio (SNR) can also be expected as the signal strength increases.

[0089] Next, a semiconductor device 1500 according to a seventh embodiment of the present disclosure will be described with reference to FIGS. 15(a), 15(b), and 16(a) to 16(d). The semiconductor device 1500 of this embodiment has a configuration in which a plurality of active antennas, including the transmitter circuits and patch antennas of the above-described embodiments, are arranged in a two-dimensional array. FIG. 15(a) is a schematic diagram of a circuit block of the semiconductor device 1500 of this embodiment, and FIG. 15(b) is a top view of the semiconductor device 1500. FIGS. 16(a), 16(b), 16(c), and 16(d) are cross-sectional views of A-A', B-B', C-C', and D-D', respectively, shown in FIG. 15(b).

[0090] FIG. 15(a) shows a block diagram of two of the four active antennas mounted on a semiconductor device 1500. Two active antennas, AA1 and AA2, are shown in FIG. 15(a). Active antennas AA3 and AA4 are omitted from FIG. 15(a). Active antenna AA1 includes modulation units 701a and 701b including an injection-locked oscillator, a synthesis unit 104, and an antenna 105, and its configuration is similar to that shown in FIG. 7(c). The modulation units 701a and 701b including an injection-locked oscillator include RTD oscillation units 160a and 160b (RTD units 110a and 110b) as injection-locked oscillators, respectively. Active antennas AA2 to AA4 have the same configuration as active antenna AA1, and a common signal line is connected to these active antennas AA1 to AA4.

[0091] The common signal lines are for two systems of data signals 109a and 109b and a reference signal from reference oscillator 101. Data signal 109a is supplied to modulation units 701a and 701c, and data signal 109b is supplied to modulation units 701b and 701d. The reference signal from reference oscillator 101 is supplied to modulation units 701a and 701c, which include injection-locked oscillators, without passing through phase adjuster 106, and is supplied to modulation units 701b and 701d, which also include injection-locked oscillators, via phase adjuster 106. Active antenna AA1 and active antenna AA2 are configured to receive the same signal, and therefore can radiate the same RF signal into space as terahertz waves TW.

[0092] As shown in FIG. 16(a), inside the active antenna AA1, a semiconductor substrate 156, a GND layer 150 which is a conductor pattern, a lower electrode 152a, an RTD layer 153a, an upper electrode 154a, and a conductor pattern 1512 are laminated in this order in the RTD section 110a. The upper electrode 154a is formed inside the dielectric sections 151a to 151d, and the conductor pattern 1512 and the RTD layer 153a are electrically connected via the upper electrode 154a. Here, if the width of the upper electrode 154a is too large, the resonance characteristics of the patch antenna will deteriorate and the radiation efficiency will decrease due to an increase in parasitic capacitance. For this reason, the width of the upper electrode 154a should be set to a dimension that does not interfere with the resonant electric field, typically a dimension that does not exceed the oscillation frequency f that exists in the active antenna AA1. THz The width of the upper electrode 154a may be configured to be 1 / 10 or less of the effective wavelength λ of the terahertz wave. The width of the upper electrode 154a may be small enough not to increase the series resistance, for example, it may be reduced to about twice the skin depth. In order to reduce the series resistance to a level not exceeding 1 Ω, the width of the upper electrode 154a may typically be in the range of 0.1 μm or more and 20 μm or less.

[0093] As shown in FIG. 16(b), the conductor pattern 1512 is electrically connected to the bias pattern 1510, which is a common wiring, via the via 1511, and is electrically connected to the bias unit. The bias unit may also be called a power supply circuit. The bias pattern 1510 is disposed between the dielectric portion 151b and the dielectric portion 151a. The bias pattern 1510 is drawn out from each of the conductor patterns 1512. The common bias pattern 1510 ensures a sufficient wiring width, thereby reducing variations in operating voltage between antennas due to variations in wiring resistance. This stabilizes synchronization even when the number of active antennas is increased. Another advantage is that the structure around the antenna can be made symmetrical, preventing the radiation pattern from collapsing.

[0094] The vias 1511 connecting the conductor pattern 1512 and the bias pattern 1510 are connection parts for electrically and mechanically connecting the bias pattern 1510 to the conductor pattern 1512. In this way, a structure that electrically connects upper and lower layers is called a via. The GND layer 150 and the conductor pattern 1512 constitute a patch antenna. In addition to serving as components, by being connected to these vias, they also serve as electrodes for injecting current into the RTD layer 153a. In this embodiment, the vias are made of a material having a resistivity of 1×10 -6 A conductive material with a resistance of Ω·m or less can be used. Specifically, metals and metal compounds such as Ag, Au, Cu, W, Ni, Cr, Ti, Al, AuIn alloy, and TiN can be used as the via material.

[0095] The width of the via 1511 connecting the conductor pattern 1512 and the bias pattern 1510 is smaller than the width of the conductor pattern 1512. Here, the width of the conductor pattern 1512 is the width in the electromagnetic wave resonance direction (i.e., the A-A' direction) within the active antenna AA1. Also, the width of the portion (connection portion) of the bias pattern 1510 connected to the via 1511 is smaller (thinner) than the width of the conductor pattern 1512 (active antenna AA1). Also, these widths are set to be smaller (thinner) than the width of the oscillating frequency f of the oscillating signal standing in the active antenna. THz The effective wavelength λ of the terahertz wave can be set to 1 / 10 or less (λ / 10 or less) of the effective wavelength λ of the terahertz wave. This is because the radiation efficiency can be improved by arranging the via 1511 and the bias pattern 1510 in a size and position that does not interfere with the resonant electric field in the active antenna.

[0096] The position of the via 1511 is determined based on the oscillation frequency f THz In this case, the via 1511 and the bias pattern 1510 can be arranged at a node of the electric field in the standing wave of the oscillation frequency f THz In other words, the via 1511 and the bias pattern 1510 are configured such that the impedance is sufficiently higher than the absolute value of the negative differential resistance of the RTD layer 153a, which is an RTD, in a frequency band around the oscillation frequency f THzThe active antenna AA is connected to the RTD so that it has high impedance at frequency f THz In this case, the path via the bias pattern 1510 for bias supply is isolated. As a result, the oscillation frequency f induced in each active antenna via the bias pattern 1510 and the bias section THz The current of the active antenna does not affect the adjacent antenna. THz Interference between the electric field of the power supply member and these power supply members is suppressed.

[0097] The bias pattern 1510 is a bias supply wiring common to multiple active antennas. The bias unit may be arranged outside the chip to supply a bias to the RTD layer 153a of each active antenna. The bias unit includes a stabilization circuit for suppressing low-frequency parasitic oscillation. The stabilization circuit is set to have an impedance lower than the absolute value of the negative resistance corresponding to the gain of the RTD layer 153a in the frequency band from DC to 10 GHz. For stabilization of relatively high frequencies from 0.1 to 10 GHz, an AC short circuit having a resistor and a capacitor connected in series may be arranged for each active antenna. In this case, the capacitance is large within the above-mentioned frequency range, for example, a capacitance of about several pF.

[0098] In this embodiment, for example, focusing on the active antenna AA1, as shown in FIG. 15(b), a reference signal is injected from the reference oscillator 101 to port 1 via a control line IL1. Furthermore, a reference signal obtained by shifting the phase of the output of the reference oscillator 101 by a phase adjuster 106 is injected to port 2 via a control line IL2. In the example of FIG. 15(b), a stub 1513 is configured to extend from port 1 to the opposite side of the RTD layer 153a. Here, the stub 1513 is set to a length of one-fourth the wavelength of the terahertz wave and is grounded to GND via an MIM capacitor 1514. Furthermore, the reference signal injected to port 1 is injected into the RTD layer 153a via a control line RL extending to the opposite side of the stub 1513. The length of the control line RL is designed to be one-fourth the wavelength of the terahertz wave so as to function as an RTD resonator.

[0099] The data signal 109a is supplied to the RTD unit 110b via a control line DL1. The control line DL1 transmits only the AC component of the data signal, so it may be connected to the RTD unit 110b using a frequency filter according to the data signal or capacitive coupling for DC cutting. The data signal 109b is supplied to the RTD unit 110a via a control line DL2. In this embodiment, the same control line DL2 parallel to the B-B' direction is connected to the active antennas AA1 and AA3. This is because the frequency f of the data signal D This is because the difference in the line length of the control line DL2 is sufficiently short compared to the wavelength of the data signal, since the wavelength of the data signal is in the order of tens of GHz. On the other hand, by designing the wiring route with equal-length wiring, it is possible to suppress the variation in the signal input to each active antenna, and it is also possible to wire each active antenna independently.

[0100] As shown in FIG. 16(d), a conductor pattern 1515 is disposed between the dielectric portion 151d and the dielectric portion 151c. The conductor pattern 1515 is used as a wiring pattern for the control line IL1, the control line IL2, the stub 1513, the control line RL, and the like, and is provided so that a reference signal can be injected into each active antenna. For the sake of explanation, FIG. 15(b) illustrates the control lines IL1, IL2, the control line RL, and the stub 1513 separately. However, these are a single conductor and are not necessarily separated as shown. The stub 1513 is the conductor pattern 1515 shown in FIG. 16(d), but is connected to the conductor pattern 1516 and grounded to the GND layer 150, which is a conductor pattern, via the MIM capacitor 1517. The control line RL extending to the RTD portion of the conductor pattern 1515 is connected to the upper electrode 154d, as shown in FIG. 16(d), and is configured to inject a reference signal into the RTD layer 153d.

[0101] As shown in FIG. 16(c), a conductor pattern 1518 is disposed between the dielectric portion 151c and the dielectric portion 151b. The conductor pattern 1518 is used as a wiring pattern for the control lines DL1 and DL2, and is provided so that data signals for modulation can be input to each active antenna. The control lines DL1 and DL2 may be provided with DC blocking capacitive coupling sections at various locations, or filters that pass signals in a relatively low frequency band, such as data signals. In this case, as shown in FIG. 16(d), an MIM capacitor grounded to the GND layer 150 may also be disposed. The conductor pattern 1518 is connected to the upper electrode 154c of the RTD section 110c and configured to perform modulation using the RTD layer 153c.

[0102] FIG. 15(b) is a top view of the semiconductor device 1500. The semiconductor device 1500 forms an antenna array, with four active antennas AA1 to AA4 arranged in a 2×2 matrix. Each of these active antennas AA1 to AA4 radiates an RF signal represented by a terahertz wave TW into space. The number of active antennas is not limited to four, and the active antennas may be arranged in a 3×3 matrix or a 2×4 matrix. A transmitter to which the present disclosure is applied can be configured as long as there are M×N active antennas (M and N are natural numbers). The RF signals radiated from these active antennas AA1 to AA4 are signals oscillated by RTD oscillators synchronized by the reference oscillator 101. Therefore, the RF signals from each active antenna can be synchronized with high precision, and as a result, terahertz waves with higher intensity can be radiated into space than with a single active antenna. This improves the gain of the antenna array.

[0103] Although this embodiment uses a common bias pattern 1510, a bias may be supplied to the RTD section via a separate bias pattern for each active antenna. In this case, overcrowding of the wiring pattern can be avoided by forming through-vias in a semiconductor substrate or the like and stacking the RTD section on a CMOS integrated circuit. As shown in FIG. 2(b), the phase of the RTD oscillator can be varied within the locking range by changing the bias voltage. In other words, the phase of each active antenna can be varied according to the frequency difference between the actual oscillation frequency locked by the reference signal of the reference oscillator and the self-oscillation frequency determined by the bias voltage of the RTD oscillator. For example, by applying different bias voltages to the RTD sections of the RTD oscillators 160a and 160b arranged in active antenna AA1 and the RTD oscillators 160c and 160d arranged in active antenna AA2, the RTD sections can be operated so that their self-oscillation frequencies are different. This enables beamforming in active antenna arrays arranged in one-dimensional and two-dimensional planes.

[0104] In this embodiment, the functions of the injection-locked oscillator and the modulator can be integrated into the active antenna, enabling a compact transmitter to be realized. An IQ mixer can be used with a single active antenna, and the active antennas can be arranged in an array. This has the added benefit of providing high-intensity power that cannot be output by individual active antennas, and sharp directivity. This allows the overall transmitter configuration to be made more compact.

[0105] Other embodiments Although the embodiments of the present invention have been described above, the present invention is not limited to these embodiments, and various modifications and changes are possible within the scope of the gist of the present invention.

[0106] For example, while the above-described embodiments have been described using a transmitter in a wireless communication device as an example, the application is not limited thereto and may also be, for example, a receiver or radar device. A radar device requires constant sweeping using a chirp signal, but the circuit of the present invention, such as that shown in FIG. 1, can be used to sweep the changes in the modulated data signal. This makes it possible to determine the delay of the received signal, and by performing absolute value and phase conversion using an IQ signal, it is possible to use the circuit to accurately detect distance and azimuth. In other words, a wave emitted from a transmitter equipped with the semiconductor device of each of the above-described embodiments and reflected by an object can be detected by a receiver, and the distance and azimuth to the object can be measured from the emitted wave and reflected wave.

[0107] In addition, while the above-described embodiments use an RTD oscillator having a resonant structure with an RTD as an injection-locked oscillator, other elements or circuits with gain other than an RTD can also be used as long as the oscillator oscillates in the terahertz band. Furthermore, even in the resonant structure, a resonant circuit may be configured by appropriately arranging lumped constant circuits such as inductance and capacitance to control oscillation. Furthermore, while the above-described embodiments use a square patch antenna as the terahertz wave antenna, the shape of the antenna is not limited to this. For example, a rectangular, triangular, polygonal, circular, or elliptical patch conductor, a planar antenna such as a loop antenna, a log-periodic antenna, or a Vivaldi antenna, or a horn antenna may also be used as the antenna.

[0108] For example, while the above-described embodiments describe an example in which two RTDs are connected to one RTD oscillator, it is also possible to connect only one or more RTDs and oscillate them within the resonator. Using only one RTD allows for control of oscillation without considering RTD variations, while using multiple RTDs allows for the construction of a high-power oscillator. Furthermore, while the above description illustrates a single mixer connected to one RTD oscillator, connections such as 1:N, N:1, or N:N (N is a natural number) may also be used. Components such as combiners and branchers may also be appropriately arranged in various locations in the circuit in a redundant configuration. Furthermore, the inductance and capacitance constants set in the filter circuits used in the above-described embodiments may be selected appropriately depending on the frequency of the signals to be passed or blocked.

[0109] Furthermore, although the structure of the RTD oscillator using a laminated structure has been described, the present invention is not limited to this, and the above discussion can also be applied to oscillator devices that do not use a laminated structure.

[0110] Furthermore, the following combinations may be used as materials for the RTD. GaAs / AlGaAs and GaAs / AlAs, In GaAs / GaAs / AlAs InGaAs / InAlAs, InGaAs / AlAs, In formed on InP substrate GaAs / AlGaAsSb InAs / AlAsSb and InAs / AlSb grown on InAs substrates ·SiGe / SiGe formed on a Si substrate The above-mentioned structure and materials can be selected appropriately depending on the desired frequency and the like.

[0111] Furthermore, circuits with frequencies relatively lower than the terahertz band, such as reference oscillators and data signals, can be connected to semiconductor substrates using general Si-CMOS circuits for transmission. Antennas and other devices can also be patterned on different semiconductor substrates or connected via connectors to mechanical structures fabricated using machining techniques, which can be used as a method to reduce material and manufacturing costs.

[0112] Next, a case where the semiconductor device of each of the above-described embodiments is applied to a terahertz camera system (imaging system) will be described. The following description will be made with reference to FIG. 13( a). The terahertz camera system 1300 includes a transmitter 1301 that emits terahertz waves TW and a detector 1302 that detects the terahertz waves TW. The terahertz camera system 1300 further includes a controller 1303 that controls the operation of the transmitter 1301 and the detector 1302 based on an external signal, and processes or externally outputs an image based on the detected terahertz waves. The semiconductor device of each of the embodiments may be used as the transmitter 1301 or the detector 1302. The terahertz waves emitted from the transmitter 1301 are reflected by a subject 1305 and detected by the detector 1302. A camera system including such a transmitter 1301 and detector 1302 may also be called an active reflection imaging camera system. The camera system can also be applied to a transmission imaging system in which the transmitter 1301 and the detector 1302 are arranged opposite each other and a subject is placed between them, thereby observing terahertz waves that have passed through the subject. Furthermore, in a passive camera system in which the transmitter 1301 is not provided, the semiconductor device of each of the above-described embodiments can be used as the detector 1302. By using the semiconductor device of each of the embodiments that can modulate and demodulate IQ signals, the detection sensitivity of the camera system can be improved, and high-quality images can be obtained.

[0113] Furthermore, a case where any of the devices according to the above-described embodiments is applied to a terahertz communication system (communication device) will be described below. The following description will be made with reference to FIG. 13(b). The semiconductor device can be used as a component of the communication system. Possible communication systems include a simple ASK system, a superheterodyne system, and a direct conversion system. A superheterodyne communication system includes, for example, an antenna 1400, an amplifier 1401, a mixer 1402, a filter 1403, a mixer 1404, a converter 1405, a digital baseband modulator / demodulator 1406, and local oscillators 1407 and 1408. The RTD oscillator 102a of the above-described embodiment can be used as the local oscillator 1407, the antenna 105 as the antenna 1400, and the mixer 103a of FIG. 1 as the mixer 1402.

[0114] In the receiver, terahertz waves received via antenna 1400 are converted into an intermediate frequency signal by mixer 1402. Then, mixer 1404 converts the signal into a baseband signal, and converter 1405 converts the analog waveform into a digital waveform. The digital waveform is then demodulated at baseband to obtain a communication signal. In the transmitter, the communication signal is modulated, then converted from a digital waveform into an analog waveform by converter 1405, and then frequency-converted via mixers 1404 and 1402, and output from antenna 1400 as a terahertz wave.

[0115] The direct conversion communication system includes an antenna 1400, an amplifier 1411, a mixer 1412, a modulator / demodulator 1413, and a local oscillator 1414. In the direct conversion system, during reception, the mixer 1412 directly converts the received terahertz wave into a baseband signal. During transmission, the mixer 1412 converts the baseband signal to be transmitted into a terahertz signal. The other configurations are similar to those of the superheterodyne system. The devices according to the above-described embodiments can simplify the local oscillator and mixer configurations of the semiconductor device and can support QAM modulation, such as generation of IQ signals. Therefore, by using the semiconductor device according to the above-described embodiments capable of IQ modulation, wireless quality such as the signal-to-noise ratio can be improved in the communication system, and large-capacity information transmission can be performed over a wide coverage area at low cost. Furthermore, if frequency sweeping can be performed using the same local oscillator for transmission and reception, it will be possible to measure distance from the delay and phase information of the transmitted signal and the reflected wave as a radar device. By applying this invention, wireless quality such as the signal-to-noise ratio can be improved, and distance measurement accuracy can be improved, all at low cost.

[0116] The disclosure of this specification includes the following semiconductor devices, communication devices, imaging systems, and radar devices.

[0117] (Item 1) a reference signal oscillator for generating a reference signal; a first injection-locked oscillator and a second injection-locked oscillator that oscillate at a frequency in the terahertz band in synchronization with the reference signal; a first modulation unit connected to the first injection-locked oscillator; a second modulation unit connected to the second injection-locked oscillator; a signal transmission unit for transmitting the signal output from the first modulation unit and the signal output from the second modulation unit as electromagnetic waves, A semiconductor device, characterized in that the phase of the signal supplied from the first injection-locked oscillator to the first modulation section is shifted from the phase of the signal supplied from the second injection-locked oscillator to the second modulation section.

[0118] (Item 2) 2. The semiconductor device of claim 1, wherein each of the first injection-locked oscillator and the second injection-locked oscillator includes a resonant tunneling diode.

[0119] (Item 3) 3. The semiconductor device according to item 1 or 2, wherein a phase adjuster that adjusts the phase of the reference signal is disposed between the reference signal oscillating unit and the second injection-locked oscillator.

[0120] (Item 4) 4. The semiconductor device according to item 3, wherein a phase adjuster other than the phase adjuster that adjusts the phase of the reference signal is disposed between the reference signal oscillating unit and the first injection-locked oscillator.

[0121] (Item 5) 3. The semiconductor device according to item 1 or 2, characterized in that a phase adjuster that adjusts the phase of the reference signal is disposed between the second injection-locked oscillator and the second modulation section.

[0122] (Item 6) 6. The semiconductor device according to any one of items 3 to 5, wherein the phase adjuster includes a hybrid coupler, a rat-race circuit, a line delay type phase shifter, or a polyphase filter.

[0123] (Item 7) 7. The semiconductor device according to claim 1, wherein each of the first modulation section and the second modulation section includes a mixer.

[0124] (Item 8) 8. The semiconductor device according to item 7, wherein the mixer includes a single-ended mixer, a single-balanced mixer, a bidirectional diode mixer, or a double-balanced mixer.

[0125] (Item 9) Item 9. The semiconductor device according to item 7 or 8, characterized in that a first signal line for supplying a data signal from a baseband circuit is electrically connected between the first injection-locked oscillator and the first modulation unit, and a second signal line for supplying a data signal from the baseband circuit is electrically connected between the second injection-locked oscillator and the second modulation unit.

[0126] (Item 10) 10. The semiconductor device according to item 9, characterized in that a matching circuit is arranged between the node to which the first signal line is connected and the first modulation unit, and between the node to which the second signal line is connected and the second modulation unit.

[0127] (Item 11) The semiconductor device according to any one of items 7 to 10, characterized in that a filter circuit is arranged between the first injection-locked oscillator and the first modulation section and between the second injection-locked oscillator and the second modulation section.

[0128] (Item 12) 12. The semiconductor device according to any one of items 1 to 11, wherein each of the first injection-locked oscillator and the second injection-locked oscillator comprises a resonant conductor for oscillating at a frequency in the terahertz band.

[0129] (Item 13) Item 13. The semiconductor device of item 12, further comprising a bias pattern connected to the resonant conductor of the first injection-locked oscillator and the resonant conductor of the second injection-locked oscillator to supply power to the first injection-locked oscillator and the second injection-locked oscillator.

[0130] (Item 14) each of the resonant structures including the first injection-locked oscillator and the second injection-locked oscillator includes the resonant conductor and a bottom conductor; Item 14. The semiconductor device according to item 13, wherein a dielectric portion in which the bias pattern is disposed is disposed between the resonant conductor and the lower conductor.

[0131] (Item 15) Item 15. The semiconductor device according to item 14, wherein an AC shunt is connected to the bias pattern.

[0132] (Item 16) each of the first injection-locked oscillator and the second injection-locked oscillator includes a resonant conductor for oscillating at a frequency in the terahertz band; the first modulation unit includes: a first bias pattern connected to the resonant conductor of the first injection-locked oscillator and for supplying power to the first injection-locked oscillator; and a first data signal line capacitively coupled to the first bias pattern and supplied with a data signal from a baseband circuit; The semiconductor device described in any one of items 1 to 6, characterized in that the second modulation unit includes a second bias pattern connected to the resonant conductor of the second injection-locked oscillator and for supplying power to the second injection-locked oscillator, and a second data signal line capacitively coupled to the second bias pattern and through which a data signal is supplied from the baseband circuit.

[0133] (Item 17) each of the resonant structures including the first injection-locked oscillator and the second injection-locked oscillator includes the resonant conductor and a bottom conductor; Item 17. The semiconductor device according to item 16, wherein a dielectric portion in which the first bias pattern and the second bias pattern are arranged is arranged between the resonant conductor and the lower conductor.

[0134] (Item 18) each of the first injection-locked oscillator and the second injection-locked oscillator includes a resonant conductor for oscillating at a frequency in the terahertz band; the first modulation unit includes a first impedance adjustment unit that is capacitively coupled to the resonant conductor of the first injection locked oscillator and that changes the impedance of the resonant conductor of the first injection locked oscillator in response to a data signal supplied from a baseband circuit; The semiconductor device according to any one of items 1 to 6, characterized in that the second modulation unit includes a second impedance adjustment unit that is capacitively coupled to the resonant conductor of the second injection-locked oscillator and changes the impedance of the resonant conductor of the second injection-locked oscillator in response to a data signal supplied from the baseband circuit.

[0135] (Item 19) 19. The semiconductor device according to any one of items 12 to 18, wherein each of the resonant structures including the first injection-locked oscillator and the second injection-locked oscillator is a microstrip-type resonator.

[0136] (Item 20) 20. The semiconductor device according to any one of items 12 to 19, wherein the resonant conductor comprises an open stub of λ / 4.

[0137] (Item 21) 21. The semiconductor device according to any one of items 1 to 20, wherein the reference signal is a signal having a subharmonic frequency of the frequency at which the first injection-locked oscillator and the second injection-locked oscillator oscillate.

[0138] (Item 22) The semiconductor device described in any one of items 1 to 21, characterized in that the first injection-locked oscillator and the second injection-locked oscillator up-convert at least one component of the frequency components of the reference signal to a higher frequency component and output the up-converted component.

[0139] (Item 23) 23. The semiconductor device according to any one of items 1 to 22, wherein the signal transmission unit includes a combining unit that combines the signal output from the first modulation unit and the signal output from the second modulation unit, and an antenna that radiates the signal combined by the combining unit.

[0140] (Item 24) 24. The semiconductor device according to item 23, wherein the combining section includes a Wilkinson coupler.

[0141] (Item 25) The semiconductor device described in any one of items 1 to 22, characterized in that the signal transmission unit includes an antenna connected to the first modulation unit and radiating the signal output from the first modulation unit, and an antenna connected to the second modulation unit and radiating the signal output from the second modulation unit.

[0142] (Item 26) 26. The semiconductor device according to item 25, wherein the distance between the antenna connected to the first modulation unit and the antenna connected to the second modulation unit is equal to or less than the wavelength of the signals oscillated by the first injection-locked oscillator and the second injection-locked oscillator.

[0143] (Item 27) 27. The semiconductor device according to item 25 or 26, characterized in that the signal radiated from the antenna connected to the first modulation unit and the signal radiated from the antenna connected to the second modulation unit are combined in a Fresnel region.

[0144] (Item 28) 28. The semiconductor device of any one of items 23 to 27, wherein the antenna includes a patch antenna.

[0145] (Item 29) further comprising a first semiconductor substrate on which the reference signal oscillator is disposed, and a second semiconductor substrate on which the first injection-locked oscillator and the second injection-locked oscillator are disposed, 29. The semiconductor device according to any one of items 1 to 28, wherein the first semiconductor substrate and the second semiconductor substrate are at least partially stacked.

[0146] (Item 30) 30. The semiconductor device according to any one of items 1 to 29, wherein no active element is arranged in a path between the first injection-locked oscillator and the first modulation unit and in a path between the second injection-locked oscillator and the second modulation unit.

[0147] (Item 31) a third injection-locked oscillator and a fourth injection-locked oscillator that oscillate at a frequency in the terahertz band in synchronization with the reference signal; a third modulation unit connected to the third injection-locked oscillator; a fourth modulation unit connected to the fourth injection-locked oscillator; a signal transmission unit separate from the signal transmission unit for transmitting the signal output from the third modulation unit and the signal output from the fourth modulation unit as electromagnetic waves, A semiconductor device described in any one of items 1 to 30, characterized in that the frequencies of self-oscillation of the third injection-locked oscillator and the fourth injection-locked oscillator are different from the frequencies of self-oscillation of the first injection-locked oscillator and the second injection-locked oscillator.

[0148] (Item 32) The semiconductor device according to any one of items 1 to 31, characterized in that the reference signal oscillation unit includes a reference oscillator common to the first injection locked oscillator and the second injection locked oscillator for supplying the reference signal to the first injection locked oscillator and the second injection locked oscillator.

[0149] (Item 33) The semiconductor device described in any one of items 1 to 31, characterized in that the reference signal oscillation unit includes a first reference oscillator that supplies the reference signal to the first injection-locked oscillator, and a second reference oscillator that supplies the reference signal to the second injection-locked oscillator based on a signal output by the first reference oscillator.

[0150] (Item 34) a reference signal oscillator for generating a reference signal; a first injection-locked oscillator and a second injection-locked oscillator, each of which includes an oscillator that oscillates at a frequency in the terahertz band in synchronization with the reference signal, the first injection-locked oscillator outputs a first signal obtained by modulating a first data signal supplied from a baseband circuit; the second injection-locked oscillator outputs a second signal obtained by modulating a second data signal supplied from the baseband circuit; A semiconductor device, wherein the first signal and the second signal are out of phase with each other.

[0151] (Item 35) Item 35. The semiconductor device of item 34, wherein each of the first injection-locked oscillator and the second injection-locked oscillator includes a resonant tunneling diode.

[0152] (Item 36) a patch antenna for transmitting the first signal and the second signal; the reference signal oscillator is disposed on a first semiconductor substrate; Item 36. The semiconductor device according to item 34 or 35, characterized in that a plurality of active antennas each including the first injection-locked oscillator, the second injection-locked oscillator, and the patch antenna are arranged in a two-dimensional array on a second semiconductor substrate.

[0153] (Item 37) A transmitter including the semiconductor device according to any one of items 1 to 36; a receiving section for detecting a signal emitted from the transmitting section.

[0154] (Item 38) a transmitter that includes the semiconductor device according to any one of items 1 to 36 and emits a signal toward a subject; a detection unit that detects the signal reflected from or transmitted through the subject.

[0155] (Item 39) A transmitter including the semiconductor device according to any one of items 1 to 36; a receiving unit that detects a wave that is emitted from the transmitting unit and reflected by an object, A radar device that measures distance from the radiation wave emitted from the transmitting unit and the reflected wave.

[0156] The invention is not limited to the above-described embodiments, and various changes and modifications can be made without departing from the spirit and scope of the invention. Accordingly, the following claims are appended to apprise the public of the scope of the invention. [Explanation of symbols]

[0157] 100, 700, 800, 900, 1000, 1200, 1500: Semiconductor devices

Claims

1. a reference signal oscillator for generating a reference signal; a first injection-locked oscillator and a second injection-locked oscillator that oscillate at a frequency in the terahertz band in synchronization with the reference signal; a first modulation unit connected to the first injection-locked oscillator; a second modulation unit connected to the second injection-locked oscillator; a signal transmission unit for transmitting the signal output from the first modulation unit and the signal output from the second modulation unit as electromagnetic waves, A semiconductor device, characterized in that the phase of the signal supplied from the first injection-locked oscillator to the first modulation section is shifted from the phase of the signal supplied from the second injection-locked oscillator to the second modulation section.

2. 2. The semiconductor device of claim 1, wherein the first injection-locked oscillator and the second injection-locked oscillator each include a resonant tunneling diode.

3. 2. The semiconductor device according to claim 1, further comprising a phase adjuster arranged between the reference signal oscillator and the second injection-locked oscillator, the phase adjuster adjusting the phase of the reference signal.

4. 4. The semiconductor device according to claim 3, further comprising a phase adjuster, separate from the phase adjuster for adjusting a phase of the reference signal, disposed between the reference signal oscillator and the first injection-locked oscillator.

5. 2. The semiconductor device according to claim 1, further comprising a phase adjuster arranged between the second injection-locked oscillator and the second modulation section, the phase adjuster adjusting the phase of the reference signal.

6. 4. The semiconductor device according to claim 3, wherein the phase adjuster includes a hybrid coupler, a rat-race circuit, a line delay type phase shifter, or a polyphase filter.

7. 2. The semiconductor device according to claim 1, wherein each of the first modulation section and the second modulation section includes a mixer.

8. 8. The semiconductor device of claim 7, wherein the mixer comprises a single-ended mixer, a single-balanced mixer, a bidirectional diode mixer, or a double-balanced mixer.

9. 8. The semiconductor device according to claim 7, wherein a first signal line for supplying a data signal from a baseband circuit is electrically connected between the first injection-locked oscillator and the first modulation unit, and a second signal line for supplying a data signal from the baseband circuit is electrically connected between the second injection-locked oscillator and the second modulation unit.

10. 10. The semiconductor device according to claim 9, wherein a matching circuit is arranged between the node to which the first signal line is connected and the first modulation unit, and between the node to which the second signal line is connected and the second modulation unit.

11. 8. The semiconductor device according to claim 7, wherein a filter circuit is disposed between the first injection-locked oscillator and the first modulation section and between the second injection-locked oscillator and the second modulation section.

12. 2. The semiconductor device according to claim 1, wherein each of the first injection-locked oscillator and the second injection-locked oscillator comprises a resonant conductor for oscillating at a frequency in the terahertz band.

13. 13. The semiconductor device of claim 12, further comprising a bias pattern connected to the resonant conductor of the first injection-locked oscillator and the resonant conductor of the second injection-locked oscillator to supply power to the first injection-locked oscillator and the second injection-locked oscillator.

14. each of the resonant structures including the first injection-locked oscillator and the second injection-locked oscillator includes the resonant conductor and a bottom conductor; 14. The semiconductor device according to claim 13, wherein a dielectric portion in which the bias pattern is disposed is disposed between the resonant conductor and the lower conductor.

15. 15. The semiconductor device according to claim 14, wherein an AC shunt is connected to the bias pattern.

16. each of the first injection-locked oscillator and the second injection-locked oscillator includes a resonant conductor for oscillating at a frequency in the terahertz band; the first modulation unit includes: a first bias pattern connected to the resonant conductor of the first injection-locked oscillator and for supplying power to the first injection-locked oscillator; and a first data signal line capacitively coupled to the first bias pattern and supplied with a data signal from a baseband circuit; 2. The semiconductor device according to claim 1, wherein the second modulation section includes: a second bias pattern connected to the resonant conductor of the second injection-locked oscillator and for supplying power to the second injection-locked oscillator; and a second data signal line capacitively coupled to the second bias pattern and through which a data signal is supplied from the baseband circuit.

17. each of the resonant structures including the first injection-locked oscillator and the second injection-locked oscillator includes the resonant conductor and a bottom conductor; 17. The semiconductor device according to claim 16, wherein a dielectric portion in which the first bias pattern and the second bias pattern are arranged is disposed between the resonant conductor and the lower conductor.

18. each of the first injection-locked oscillator and the second injection-locked oscillator includes a resonant conductor for oscillating at a frequency in the terahertz band; the first modulation unit includes a first impedance adjustment unit that is capacitively coupled to the resonant conductor of the first injection locked oscillator and that changes the impedance of the resonant conductor of the first injection locked oscillator in response to a data signal supplied from a baseband circuit; 2. The semiconductor device according to claim 1, wherein the second modulation unit includes a second impedance adjustment unit that is capacitively coupled to the resonant conductor of the second injection-locked oscillator and that changes the impedance of the resonant conductor of the second injection-locked oscillator in accordance with a data signal supplied from the baseband circuit.

19. 13. The semiconductor device of claim 12, wherein each of the resonant structures comprising the first injection-locked oscillator and the second injection-locked oscillator is a microstrip-type resonator.

20. 13. The semiconductor device of claim 12, wherein the resonant conductor comprises a λ / 4 open stub.

21. 2. The semiconductor device according to claim 1, wherein the reference signal is a signal having a frequency that is a subharmonic of the frequency at which the first injection-locked oscillator and the second injection-locked oscillator oscillate.

22. 2. The semiconductor device according to claim 1, wherein the first injection-locked oscillator and the second injection-locked oscillator up-convert at least one of the frequency components of the reference signal to a higher frequency component and output the up-converted component.

23. 2. The semiconductor device according to claim 1, wherein the signal transmission unit includes a combining unit that combines the signal output from the first modulation unit and the signal output from the second modulation unit, and an antenna that radiates the signal combined by the combining unit.

24. 24. The semiconductor device of claim 23, wherein the combiner comprises a Wilkinson coupler.

25. 2. The semiconductor device according to claim 1, wherein the signal transmission unit includes an antenna connected to the first modulation unit and radiating the signal output from the first modulation unit, and an antenna connected to the second modulation unit and radiating the signal output from the second modulation unit.

26. 26. The semiconductor device of claim 25, wherein the distance between the antenna connected to the first modulation unit and the antenna connected to the second modulation unit is equal to or less than the wavelength of the signals oscillated by the first injection-locked oscillator and the second injection-locked oscillator.

27. 26. The semiconductor device according to claim 25, wherein a signal radiated from the antenna connected to the first modulation section and a signal radiated from the antenna connected to the second modulation section are combined in a Fresnel region.

28. 24. The semiconductor device of claim 23, wherein the antenna comprises a patch antenna.

29. further comprising a first semiconductor substrate on which the reference signal oscillator is disposed, and a second semiconductor substrate on which the first injection-locked oscillator and the second injection-locked oscillator are disposed, The semiconductor device according to claim 1 , wherein the first semiconductor substrate and the second semiconductor substrate are at least partially stacked.

30. 2. The semiconductor device according to claim 1, wherein no active elements are arranged in a path between the first injection-locked oscillator and the first modulation section and in a path between the second injection-locked oscillator and the second modulation section.

31. a third injection-locked oscillator and a fourth injection-locked oscillator that oscillate at a frequency in the terahertz band in synchronization with the reference signal; a third modulation unit connected to the third injection-locked oscillator; a fourth modulation unit connected to the fourth injection-locked oscillator; a signal transmission unit separate from the signal transmission unit for transmitting the signal output from the third modulation unit and the signal output from the fourth modulation unit as electromagnetic waves, 2. The semiconductor device of claim 1, wherein the third injection-locked oscillator and the fourth injection-locked oscillator operate such that the frequency of self-oscillation is different from the frequency of self-oscillation of the first injection-locked oscillator and the second injection-locked oscillator.

32. 2. The semiconductor device according to claim 1, wherein the reference signal oscillator includes a reference oscillator common to the first injection-locked oscillator and the second injection-locked oscillator for supplying the reference signal to the first injection-locked oscillator and the second injection-locked oscillator.

33. 2. The semiconductor device according to claim 1, wherein the reference signal oscillation unit includes: a first reference oscillator that supplies the reference signal to the first injection-locked oscillator; and a second reference oscillator that supplies the reference signal to the second injection-locked oscillator based on a signal output by the first reference oscillator.

34. a reference signal oscillator for generating a reference signal; a first injection-locked oscillator and a second injection-locked oscillator, each of which includes an oscillator that oscillates at a frequency in the terahertz band in synchronization with the reference signal, the first injection-locked oscillator outputs a first signal obtained by modulating a first data signal supplied from a baseband circuit; the second injection-locked oscillator outputs a second signal obtained by modulating a second data signal supplied from the baseband circuit; A semiconductor device, wherein the first signal and the second signal are out of phase with each other.

35. 35. The semiconductor device of claim 34, wherein the first injection-locked oscillator and the second injection-locked oscillator each include a resonant tunneling diode.

36. a patch antenna for transmitting the first signal and the second signal; the reference signal oscillator is disposed on a first semiconductor substrate; 35. The semiconductor device of claim 34, wherein a plurality of active antennas, each including the first injection-locked oscillator, the second injection-locked oscillator, and the patch antenna, are arranged in a two-dimensional array on a second semiconductor substrate.

37. a transmitter comprising a semiconductor device according to any one of claims 1 to 36; a receiving section for detecting a signal emitted from the transmitting section.

38. a transmitter comprising the semiconductor device according to any one of claims 1 to 36, which emits a signal toward a subject; a detection unit that detects the signal reflected from or transmitted through the subject.

39. a transmitter comprising a semiconductor device according to any one of claims 1 to 36; a receiving unit that detects a wave that is emitted from the transmitting unit and reflected by an object, A radar device that measures distance from the radiation wave emitted from the transmitting unit and the reflected wave.

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