Power conversion circuit, semiconductor device, and electronic apparatus

Abstract

This power conversion circuit comprises: a first capacitor having one end connected to an input terminal and the other end connected to a reference potential; a second capacitor having one end connected to an output terminal and the other end connected to the reference potential; an inductor which has one end connected to the one end of the first capacitor, and the other end connected to the one end or the other end of the second capacitor, accumulates magnetic field energy by allowing at least a part of an input current and the current output from the first capacitor to flow as a first current, and induces, with the magnetic field energy, a second current that causes the second capacitor to store electric charges; and a switching element that turns on and off in a substantially constant cycle, wherein the turn-on period of time during one cycle is almost constant, the first current flows through the inductor when turned on, and the switching element turns off when the second current flows through the inductor.

Description

Technical Field

[0001] The present invention relates to a power conversion circuit, a semiconductor device, and an electronic device.

Background Art

[0002] For conversion of electric power generated by an ambient power generation device such as a vibration power generation device, a power conversion circuit such as an AC (Alternating Current)-DC (Direct Current) converter or a DC-DC converter is used. It is known to match the impedance when converting the electric power generated by a vibration power generation device using a piezoelectric element into AC-DC (for example, Non-Patent Document 1).

Prior Art Documents

Non-Patent Documents

[0003]

Non-Patent Document 1

Summary of the Invention

Problems to be Solved by the Invention

[0004] In an ambient power generation device, since the amount of power generation changes moment by moment, the input current input to the power conversion circuit changes moment by moment. Along with the change in the input current, the input impedance of the power conversion circuit changes, and it becomes difficult to match the output impedance of the ambient power generation device and the input impedance of the power conversion circuit. In Non-Patent Document 1, the input impedance is autonomously matched to the output impedance of the ambient power generation device in a step-up or step-down DC-DC converter in which the current flowing through the inductor is discontinuous. For this reason, the switching period is made constant, and the inductance of the inductor and the like are set to predetermined values.

[0005] A description will be given of the first problem. In Non-Patent Document 1, since power generation by an inorganic piezoelectric element with a large capacitance component is assumed, a primary capacitor is not provided on the power conversion circuit side. However, depending on the environmental power generation device, a primary capacitor is provided at the input terminal of the power conversion circuit to stabilize the input voltage. For example, when using an element with a small capacitance component such as an organic piezoelectric element or an MEMS element, a primary capacitor may be provided. When a primary capacitor is provided, the preferable ranges of the inductance of the inductor and the capacitance of the primary capacitor for facilitating impedance matching are not known. If the impedance cannot be matched, the power conversion efficiency decreases.

[0006] A description will be given of the second problem. Also, as in Non-Patent Document 1, even if an attempt is made to set the inductance of the inductor, etc. within a predetermined range, it may be difficult. For example, when the time variation of the output of the power generation device is fast, in the method of keeping the switching period constant, it is necessary to set the switching frequency high, and thus the power consumption increases. If an attempt is made to reduce the power consumption, since the inductance of the inductor, etc. deviates from the predetermined range, impedance matching may not be achievable in some cases. When the impedance is not matched, the power conversion efficiency decreases.

[0007] The present invention has been made in view of the above problems, and an object thereof is to improve the power conversion efficiency.

Means for Solving the Problems

[0008] The present invention relates to a power conversion circuit comprising: an input terminal to which an input current is input; an output terminal from which an output voltage is output; a first capacitor having one end connected to the input terminal and the other end connected to a reference potential; a second capacitor having one end connected to the output terminal and the other end connected to the reference potential; an inductor having one end connected to one end of the first capacitor and the other end connected to one end or the other end of the second capacitor, wherein magnetic field energy is accumulated when at least a part of the input current and the current output from the first capacitor flows as a first current, and a second current for charging the second capacitor is induced by the magnetic field energy; a switching element that turns on and off at a substantially constant period, the on-period during one period being substantially constant, the first current flowing through the inductor when it is turned on, and being turned off when the second current flows through the inductor. When the capacitance of the first capacitor is C1, the inductance of the inductor is L, the period is T1, the length of the on-period of the switching element during one period is T2, and the angular frequency of the input current is ω, the power conversion circuit satisfies C1 < (T2^2 / (2×L)) / (ω×T1). Yes, the input impedance is matched to the output impedance of the vibration power generation device. It is a power conversion circuit.

[0009] In the above configuration, C1 > T2 2 / (2×L) can be set as the configuration.

[0010] In the above configuration, C1 < (T2 2 / (2×L)) / (5×ω×T1) can be set as the configuration.

[0011] In the above configuration, C1 > 5×T2 2 / (2×L) can be set as the configuration.

[0013] In the above configuration, when the output of the vibration power generation device is DC, the output capacitance of the vibration power generation device is smaller than C1, and when the output of the vibration power generation device is AC, the output capacitance of the vibration power generation device via the rectifier circuit is smaller than C1 can be set as the configuration.

[0014] In the above configuration, the When the input voltage applied to the input terminal reaches the threshold voltage, the input voltage is converted The and a voltage conversion operation for outputting the voltage as an output voltage to the output terminal is started And input to the input terminal The The threshold voltage when the input current is large is higher than the threshold voltage when the input current is small It can be configured as .

[0015] In the above configuration, input to the input terminal The Based on the input current, a setting circuit for setting the threshold voltage such that the threshold voltage when the input current is large is higher than the threshold voltage when the input current is small, and a determination circuit for determining whether or not the input voltage has reached the threshold voltage are provided , the Based on the output of the determination circuit, the voltage conversion operation can be started.

[0018] In the above configuration, when the output impedance of the power source connected to the input terminal is Zout and the threshold voltage when the input current input to the input terminal is Iin is Vth, it can be configured such that 0.3×Zout≦Vth / Iin≦3×Zout.

[0019] The present invention is a semiconductor device including the above power conversion circuit and a functional circuit related to the power conversion circuit.

[0020] The present invention is an electronic device including the above power conversion circuit and a functional unit supplied with power from the power conversion circuit.

Advantages of the Invention

[0021] According to the present invention, the power conversion efficiency can be improved.

Brief Description of the Drawings

[0022]

Figure 1

Figure 2

Figure 3

Figure 4

Figure 5

Figure 6

Figure 7

Figure 8

Figure 9

Figure 10

BEST MODE FOR CARRYING OUT THE INVENTION

[0023] Hereinafter, embodiments will be described with reference to the drawings.

Embodiment

[0024] Example 1 for solving the first problem will be described. The power conversion circuit according to Example 1 has a primary capacitor. In a power conversion circuit that keeps the switching period constant and autonomously matches the input impedance to the output impedance of the power generation device, the preferred ranges of the inductance of the inductor and the capacitance of the primary capacitor will be described.

[0025] Figure 1 is a block diagram of a vibration power generation system using the power conversion circuits according to Example 1, Example 2, and their modified examples. The power conversion circuit includes a voltage conversion circuit 10 and a control circuit 20 and the like, which will be described later in FIGS. 3(a) to 3(c) and the like. Capacitors C01 and C02 are included in the voltage conversion circuit 10, but are shown outside the voltage conversion circuit 10 in FIG. 1. As shown in FIG. 1, the vibration power generation system includes a vibration power generation device 14, a rectifier circuit 12, and a voltage conversion circuit 10. The vibration power generation device 14 is provided, for example, on a road or a bridge, and generates electricity by the vibration when a pedestrian or a vehicle passes by. The rectifier circuit 12 rectifies the generated power. The voltage conversion circuit 10 converts the voltage of the rectified power.

[0026] A primary capacitor C01 is connected to the input terminal Tin of the voltage conversion circuit 10, and a secondary capacitor C02 is connected to the output terminal Tout. An input current Iin enters the input terminal Tin from the rectifier circuit 12. The voltage conversion circuit 10 converts the voltage V1 at the input terminal Tin to a voltage V2. An output voltage V2 is output from the output terminal Tout. The input impedance of the voltage conversion circuit 10 is Zin, and the output impedance of the vibration power generation device 14 via the rectifier circuit 12 is Zout.

[0027] Figure 2 is an example schematically showing the vibration applied to the vibration power generation device, and is an example showing the acceleration with respect to time. FIG. 2 shows the vibration when a pedestrian walks. As shown in FIG. 2, accelerations with various periods and frequencies are applied to the vibration power generation device 14. As a result, the power output by the vibration power generation device 14 changes every moment.

[0028] In FIG. 1, the vibration power generation device 14 includes, for example, a vibrator that vibrates due to environmental vibration and a power generation element that converts the vibration of the vibrator into electric power. The power generation element is, for example, an inorganic piezoelectric element using an inorganic piezoelectric material such as PZT (lead zirconate titanate), an organic piezoelectric element using an organic piezoelectric material such as PVDF (polyvinylidene fluoride), or a MEMS (Micro Electro Mechanical Systems) element using MEMS. Compared with the MEMS element, the impedance Zo of the inorganic piezoelectric element and the organic piezoelectric element has a large capacitance component Co, and Zo≒1 / iωCo. The Co of the inorganic piezoelectric element and the organic piezoelectric element is about 100 nF and about 1 nF, respectively. ω / 2π is the frequency of the output current of the piezoelectric element. The typical ω / 2π of the inorganic piezoelectric element is about 100 Hz, for example, 10 Hz to 1 kHz. The typical ω / 2π of the organic piezoelectric element is about 10 Hz, for example, 1 Hz to 100 Hz. Even if the vibration frequency in FIG. 2 is the same, ω is different between the organic piezoelectric element and the inorganic piezoelectric element due to differences such as the resonance frequency of the vibrator. On the other hand, since the capacitance component of the output impedance of the MEMS element is as small as 10 pF to 100 pF, the resistance component Ro is dominant, and Zo≒Ro. The typical value of Ro of the MEMS element is about 1 MΩ. The typical ω / 2π of the MEMS element is about 100 Hz.

[0029] In a vibration power generation device with a large Co like an inorganic piezoelectric element, the voltage V1 is relatively easy to stabilize without providing the primary capacitor C01. However, in a vibration power generation device with a small Co like an organic piezoelectric element and a MEMS element, or a MEMS element with a negligibly small Co, the voltage V1 is stabilized by providing the primary capacitor C01. The input impedance Zin of the voltage conversion circuit 10 is expressed by Equation 1. Zin=(Iin / V1+i×ω×C1) -1 (Equation 1) ω is the angular frequency of the variation of the input current Iin. ω corresponds to, for example, the angular frequency component of the vibration of the vibrator that vibrates under the acceleration shown in FIG. 2. i is the imaginary unit. C1 is the capacitance of the capacitor C01. When matching the output impedance Zout through the rectifier circuit 12 of the vibration power generation device 14 with the input impedance Zin of the voltage conversion circuit 10 having the primary capacitor C01, it is important to consider the capacitance C1 of the capacitor C01.

[0030] FIGS. 3(a) to 3(c) are circuit diagrams showing examples of the power conversion circuit used in the first embodiment. FIG. 3(a) is a circuit diagram of a buck DC-DC converter, FIG. 3(b) is a circuit diagram of a boost DC-DC converter, and FIG. 3(c) is a circuit diagram of a buck-boost DC-DC converter.

[0031] As shown in FIGS. 3(a) to 3(c), the power conversion circuit includes a voltage conversion circuit 10 and a control circuit 20. In any of the DC-DC converters, in the voltage conversion circuit 10, nodes N1 to N3 are provided between the input terminal Tin and the output terminal Tout. One end of the capacitor C01 is connected to the node N1, and the other end is connected to the ground (reference potential). One end of the capacitor C02 is connected to the node N2, and the other end is connected to the ground (reference potential). The control circuit 20 outputs control signals S1 and S2 to the switching elements SW1 and SW2, respectively. For example, when the control signals S1 and S2 are at a high level, the switching elements SW1 and SW2 are turned on, and when the control signals S1 and S2 are at a low level, the switching elements SW1 and SW2 are turned off. The control circuit 20 may be a dedicated logic circuit or a processor operating with software.

[0032] As shown in FIG. 3(a), in a buck DC-DC converter, one end of inductor L0 is connected to node N3, and the other end is connected to node N2. One end of switching element SW1 is connected to node N1, and the other end is connected to node N3. One end of switching element SW2 is connected to node N3, and the other end is connected to ground. Voltage V2 is lower than voltage V1. When switching elements SW1 and SW2 are turned on and off respectively, part of the charge stored in capacitor C01 flows through inductor L0 as current I1 and reaches node N2. As a result, magnetic field energy is stored in inductor L0. Subsequently, switching elements SW1 and SW2 are turned off and on respectively. Thereby, due to the magnetic field energy stored in inductor L0, current I2 flows from ground through inductor L0 and reaches node N2. Through these operations, voltage V2 is maintained at a desired value.

[0033] As shown in FIG. 3(b), in a boost DC-DC converter, one end of inductor L0 is connected to node N1, and the other end is connected to node N3. One end of switching element SW1 is connected to node N3, and the other end is connected to ground. One end of switching element SW2 is connected to node N3, and the other end is connected to node N2. Voltage V2 is higher than voltage V1. When switching elements SW1 and SW2 are turned on and off respectively, part of the charge stored in capacitor C01 flows through inductor L0 as current I1 and reaches ground. When switching elements SW1 and SW2 are turned off and on respectively, due to the magnetic field energy stored in inductor L0, current I2 flows from node N1 through inductor L0 and reaches node N2. Through these operations, voltage V2 is maintained at a desired value.

[0034] As shown in Fig. 3(c), in the buck-boost DC-DC converter, one end of the inductor L0 is connected to node N3, and the other end is connected to ground. One end of the switching element SW1 is connected to node N1, and the other end is connected to node N3. One end of the switching element SW2 is connected to node N3, and the other end is connected to node N2. The voltage V2 is a voltage with the opposite sign to the voltage V1. When the switching elements SW1 and SW2 are turned on and off respectively, a part of the charge stored in the capacitor C01 flows through the inductor L0 as the current I1 and reaches the ground. When the switching elements SW1 and SW2 are turned off and on respectively, the magnetic field energy stored in the inductor L0 causes the current I2 to flow from node N2 through the inductor L0 and reach the ground. By these operations, the voltage V2 is maintained at a desired value. In Figs. 3(a) to 3(c), in addition to the current output from the capacitor C01, at least a part of the input current Iin may be added to the current I1.

[0035] Fig. 4 is a diagram showing the on and off of the switching element with respect to time and the current IL flowing through the inductor in Example 1. As the current IL flowing through the inductor L0, the current I1 is shown by a solid line, and the current I2 is shown by a dashed line.

[0036] As shown in Fig. 4, before time t1, the control circuit 20 controls the switching elements SW1 and SW2 to be turned off. At the time t1 when the current IL is 0, the control circuit 20 turns on the switching element SW1 and controls to maintain the off state of the switching element SW2. The current IL increases between the times t1 and t2. The increase rate of the current IL is ΔVL / L. Here, ΔVL is the voltage difference between both ends of the inductor L0. In the buck DC-DC converter of Fig. 3(a), ΔVL = V1 - V2. In the boost DC-DC converter of Fig. 3(b), ΔVL = V1 - 0 = V1. In the buck-boost DC-DC converter of Fig. 3(c), ΔVL = V1 - 0 = V1. Magnetic field energy is stored in the inductor L0.

[0037] At time t2, the control circuit 20 controls to turn off the switching element SW1 and turn on the switching element SW2. Note that the timing at which the switching element SW1 turns off and the timing at which the switching element SW2 turns on may be slightly offset. The magnetic field energy stored in the inductor L0 is converted into the current IL, and the current I2 flows. The current I2 decreases with time. At time t3, when the current I2 becomes 0, the control circuit 20 controls to turn off the switching element SW2 and maintain the off state of the switching element SW1. After time t1´, the same operation as after time t1 is repeated. Let the on and off periods of the switching elements SW1 and SW2 be T1 (switching period), the length of the on period of the switching element SW1 be T2, and the length of the on period of the switching element SW2 be T3.

[0038] [Impedance matching] When looking at the vibration power generation device 14 from the input terminal Tin of the voltage conversion circuit 10 via the rectification circuit 12, the output impedance Zout (that is, the combined impedance of the rectification circuit 12 and the vibration power generation device 14) is adjusted to be substantially a resistive component. For this reason, the resistive component Re(Zin) of the input impedance Zin of the voltage conversion circuit 10 is matched with the output impedance Zout of the vibration power generation device 14, and the reactance component Im(Zin) of the input impedance Zin of the voltage conversion circuit 10 is made smaller than the resistive component Re(Zin). Thereby, the input impedance Zin of the voltage conversion circuit 10 can be matched with the output impedance Zout of the vibration power generation device 14.

[0039] Assuming that C1 in the input impedance Zin of Equation 1 is sufficiently small, Zin = V1 / Iin. In the buck and inverting DC-DC converters of Figs. 3(a) and 3(c), the input current Iin is the average value of current I1. In the boost DC-DC converter of Fig. 3(b), the input current Iin is the average value of currents I1 and I2. Here, assume that the input current Iin is the average value of current I1. At this time, assuming that the slope of current I1 with respect to time is approximately constant, the input current Iin is the value obtained by multiplying the average current of current I1 in period T2 by T2 / T1. Let the current I1 at time t2 be I1(t2), then the average current of current I1 in period T2 is 1 / 2 × I1(t2). Thus, the input current Iin becomes Equation 2. Iin = 1 / 2 × I1(t2) × T2 / T1 (Equation 2) Here, assuming that the inductance of inductor L0 is L, I1(t2) = T2 × ΔVL / L. Therefore, the input current Iin of Equation 2 becomes Equation 3. Iin = (T2 2 × ΔVL) / (2 × L × T1) (Equation 3)

[0040] If ω × C1 in the input impedance Zin of Equation 1 is sufficiently smaller than Iin / V1, the input impedance Zin becomes Equation 4. Zin = V1 / Iin = (2 × L × T1) / T2 2 × V1 / ΔVL (Equation 4)

[0041] In the buck DC-DC converter of Fig. 3(a), assume that voltage V2 is much smaller than voltage V1. In the boost and inverting DC-DC converters of Figs. 3(b) and 3(c), ΔVL = V1. In these cases, the input impedance Zin of Equation 4 becomes Equation 5. Zin = V1 / Iin = (2 × L × T1) / T2 2 (Equation 5)

[0042] In Equations 4 and 5, the input impedance Zin does not depend on the input current Iin. The period T1, the period T2, and the inductance L of the inductor L0 are appropriately set so that the input impedance Zin shown in Equation 4 or Equation 5 approximately matches the output impedance Zout of the vibration power generation device 14.

[0043] In other words, when the input impedance Zin is the case of Equation 5, the values of the inductance L of the inductor L0, the period T1, and the period T2 are set so that the output impedance Zout is expressed as shown in Equation 6. Zout=(2×L×T1) / T2 2 (Equation 6) The value of the inductance L of the inductor L0 is set during design, and the values of the period T1 and the period T2 are reflected in the parameters that control the on / off timing of the switching elements SW1 and SW2 by the control circuit 20. Thereby, regardless of the magnitude of the input current Iin, the input impedance Zin of the voltage conversion circuit 10 can be matched to the output impedance Zout of the vibration power generation device 14.

[0044] In order to match the input impedance Zin to the output impedance Zout, the inductance L, the period T1, and the period T2 are 0.3×((2×L×T1) / T2 2 )≦Zout≦3×((2×L×T1) / T2 2 ) is preferably set in the range of 0.5×((2×L×T1) / T2 2 )≦Zout≦1.5×((2×L×T1) / T2 2 ) is more preferably set in the range of. Thereby, the input impedance Zin matches the output impedance Zout, and the power conversion efficiency is improved.

[0045] In the boost-type DC-DC converter of Fig. 3(b), when the input current Iin is taken as the average value of currents I1 and I2, and in the buck-type DC-DC converter of Fig. 3(a), even when ΔVL = V1 - V2, the input impedance Zin can be set to be independent of the input current Iin. Therefore, by appropriately setting each value, the input impedance Zin of the voltage conversion circuit 10 can be autonomously matched to the output impedance Zout of the vibration power generation device 14. In the case of the boost-type DC-DC converter of Fig. 3(b), in addition to the period T2, the period T3 is made substantially constant.

[0046] [Upper limit of C1] In Equation 1, when ω×C1 is larger than Iin / V1, even if the input impedance Zin in Equations 4 and 5 is adjusted to match the output impedance Zout, ω×C1 becomes an obstacle to impedance matching. Therefore, it is preferable that Iin / V1 is sufficiently larger than ω×C1. That is, it is preferable that it is Equation 7. 1 / (ω×C1) ≫ V1 / Iin (Equation 7) From Equations 5 and 7, it is preferable that it is Equation 8. C1 ≪ (T2 2 / (2×L)) / (ω×T1) (Equation 8)

[0047] From Equation 8, C1 < (T2 2 / (2×L)) / (ω×T1) is preferable, C1 < (T2 2 / (2×L)) / (ω×T1) / 5 is more preferable, C1 < (T2 2 / (2×L)) / (ω×T1) / 10 is even more preferable. By these, the impedance matching between the output impedance Zout of the vibration power generation device 14 and the input impedance Zin of the voltage conversion circuit 10 becomes easy.

[0048] [Lower limit of C1] When the capacitance C1 of the capacitor C01 is small, the voltage V1 is likely to fluctuate. If the amount of charge released from the capacitor C01 is Q1, the fluctuation amount ΔV1 of the voltage V1 is ΔV1 = Q1 / C1. The amount of charge Q1 is the charge corresponding to the current I1 in the step-down DC-DC converter of Fig. 3(a) and the inverting DC-DC converter of Fig. 3(c), and is the charge corresponding to the currents I1 and I2 in the boost DC-DC converter of Fig. 3(b). If the amount of charge Q1 is the charge corresponding to the current I1, the fluctuation amount ΔV1 of the voltage V1 is obtained by Equation 9. ΔV1 = Q1 / C1 = 1 / 2 × I1(t2) × T2 / C1 (Equation 9) The fluctuation amount ΔV1 of the voltage V1 is preferably sufficiently smaller than the voltage V1. Therefore, assuming ΔV1 ≪ V1 and substituting I1(t2) = T2 × ΔVL / L into Equation 9, we get Equation 10. T2 2 ×ΔVL / (2×L×C1) ≪ V1 (Equation 10)

[0049] Assuming ΔVL = V1, the preferable range of the capacitance C1 is Equation 11. C1 ≫ T2 2 / (2×L) (Equation 11)

[0050] From Equation 11, C1 > T2 2 / (2×L) is preferable, and C1 > 5×T2 2 / (2×L) is more preferable, and C1 > 10×T2 2 / (2×L) is even more preferable. This can reduce the fluctuation of the voltage V1.

[0051] Examples of setting the period T1, the period T2, the inductance L, and the capacitance C1 will be described. From Equations 8 and 11, T1 ≪ 1 / ω. Assuming ω = 2π×50Hz, T1 is preferably 300μs or less. T2 is appropriately one fraction of T1, for example, 100μs. From Equations 8 and 11, L and C1 only need to satisfy 2 / 2 ≪ L×C1 ≪ T2 2 / (2×T1×ω). For example, assuming L = 100mH, C1 = 150nF is preferable.

[0052] [Modification Example 1 of Embodiment 1] Modification Example 1 of Embodiment 1 is an example of a buck-boost DC-DC converter. FIG. 5 is a circuit diagram of the power conversion circuit according to Modification Example 1 of Embodiment 1. As shown in FIG. 5, the power conversion circuit of Modification Example 1 of Embodiment 1 includes a voltage conversion circuit 10a and a control circuit 20a. One end of the switching element SW1 is connected to the node N1, and the other end is connected to the node N3a. One end of the inductor L0 is connected to the node N3a, and the other end is connected to the node N3b. One end of the switching element SW4 is connected to the node N3b, and the other end is connected to the node N2. One end of the switching element SW2 is connected to the node N3a, and the other end is grounded. One end of the switching element SW3 is connected to the node N3b, and the other end is grounded. The control circuit 20a outputs control signals S1 to S4 for turning on and off the switching elements SW1 to SW4, respectively. The other circuit configurations are the same as those in FIGS. 3(a) to 3(c) of Embodiment 1 and the description thereof is omitted.

[0053] FIG. 6 is a diagram showing the on and off of the switching elements, each current, and each voltage with respect to time in Modification Example 1 of Embodiment 1. In the power generation system of FIG. 1, the input current Iin changes moment by moment, but in FIG. 6, an example in which the input current Iin increases uniformly will be described. As shown in FIG. 6, until the time t1, the control circuit 20a controls the switching elements SW1 to SW4 to be off. The currents I1 and I2 are 0. Since the input current Iin charges the capacitor C01, the voltage V1 gradually increases. At the time t1, the control circuit 20a turns on the switching elements SW1 and SW3 and controls to maintain the off state of the switching elements SW2 and SW4. Between the times t1 and t2, the current I1 gradually increases. Since the capacitor C01 discharges, the voltage V1 gradually decreases.

[0054] At time t2, the control circuit 20a controls to turn off the switching elements SW1 and SW3 and turn on the switching elements SW2 and SW4. The magnitudes of the current I1 immediately before time t2 and the current I2 immediately after time t2 are substantially the same. Between time t2 and t3, the current I1 does not flow and the current I2 gradually decreases. Since the input current Iin charges the capacitor C01, the voltage V1 gradually increases. At time t3, the control circuit 20a controls to turn off the switching elements SW2 and SW4 and maintain the off state of the switching elements SW1 and SW3. Between time t3 and t1´, the currents I1 and I2 do not flow and the voltage V1 gradually increases. Thereafter, at time t1´, the switching elements SW1 and SW3 turn on.

[0055] The switching period T1, the period T2 during which the switching elements SW1 and SW3 are on, and the period T3 during which the switching elements SW2 and SW4 are on are each substantially constant. When the input current Iin increases, the peaks of the currents I1 and I2 increase. The voltage V1 gradually increases while repeating up and down. The average of the voltage V1 in each period T1 is substantially the same as the broken line of Zout×Iin in each period T1. That is, Zin = V1 / Vin becomes substantially the same as Zout. In this way, even when the input current Iin fluctuates, the input impedance Zin can be matched to the output impedance Zout of the vibration power generation device 14. Therefore, the power conversion efficiency is improved.

[0056] According to Example 1 and its modified examples, as shown in FIGS. 3(a) to 3(c), one end of the inductor L0 is connected to one end of the capacitor C01 (first capacitor), and the other end is connected to one end on the node N2 side or the other end on the ground side of the capacitor C02 (second capacitor). In the inductor L0, magnetic field energy is accumulated when at least a part of the input current Iin and the current output from the capacitor C01 flows as the current I1 (first current). The capacitor C02 is charged by the current I2 (second current) induced by the magnetic field energy. The switching elements SW1 and SW2 turn on and off at a substantially constant period T1, and the period T2 during which they are on within one period is substantially constant. When the switching element SW1 (the switching elements SW1 and SW3 in Modified Example 1 of Example 1) is on, the current I1 flows through the inductor L0, and when the current I2 flows through the inductor L0, the switching element SW1 (the switching elements SW1 and SW3 in Modified Example 1 of Example 1) is off. Thereby, even when the input current Iin fluctuates, the input impedance Zin can be made constant. Therefore, the power conversion efficiency can be improved. Note that the fact that the period T1, the period T2, and the period T3 are substantially constant allows fluctuations of about ±20% or about ±10%.

[0057] If the capacitance C1 of the capacitor C01 is large, Zin in Equation 1 deviates from V1 / Iin, and it becomes difficult to match the input impedance Zin with the output impedance Zout of the vibration power generation device 14. Therefore, from Equation 8, C1 < (T2 2 / (2×L)) / (ω×T1). Thereby, autonomous impedance matching becomes possible. In the vibration power generation device 14, as shown in FIG. 2, in order to extract vibration energy from environmental vibrations of various periods by the vibrator and generate power, the angular frequency of the input current Iin matches the angular frequency of the vibrator. If the vibrator is a harmonic oscillator, its resonance frequency can be set as ω. If it is a non-harmonic oscillator, it is preferable to set as ω the angular frequency corresponding to the largest peak in the spectrum obtained by Fourier series expansion of the input current Iin. Note that C1 > 0.

[0058] Thus, ω is the angular frequency corresponding to the vibration frequency of the vibration power generation device 14. For example, when the vibration power generation device 14 has a resonance frequency, it is preferable that ω be the angular frequency corresponding to the resonance frequency of the vibration power generation device 14. When there are a plurality of resonance frequencies, it is preferable that ω be the angular frequency corresponding to the resonance frequency at which the input current Iin is the largest. For a non-harmonic oscillator, ω is the angular frequency corresponding to the largest peak in the spectrum obtained by performing Fourier series expansion on the input current Iin. For a non- And When using a non-harmonic oscillator, for this purpose, the non-harmonic oscillator is vibrated in advance, the input current Iin is measured, and ω is set to the angular frequency corresponding to the largest peak from the result of Fourier series expansion.

[0059] Also, when the fluctuation of the voltage V1 becomes large, it becomes difficult to keep the input impedance Zin constant. Therefore, from Equation 11, C1 > T2 2 / (2×L). Thereby, the fluctuation of the voltage V1 can be reduced.

[0060] An example in which the current generated by the vibration power generation device 14 is input as the input current Iin to the input terminal Tin has been described. However, the voltage conversion circuit 10 or 10a may be used for power conversion of an environmental power generation device such as a solar cell or a wind power generator. When the power generation is direct current like a solar cell, the rectifier circuit 12 can be omitted. The power generation amount of the environmental power generation device is likely to fluctuate, and the input impedance of the voltage conversion circuit 10 or 10a is likely to fluctuate. By using the voltage conversion circuit 10 or 10a of Example 1 and its modified example for environmental power generation, the input impedance of the voltage conversion circuit 10 or 10a can be matched with the output impedance of the environmental power generation device. The voltage conversion circuit 10 or 10a of Example 1 and its modified example may be used for other applications.

[0061] When the capacitance of the vibration power generation device 14 is small, such as an organic piezoelectric element or a MEMS element, a primary capacitor C01 is provided to stabilize the voltage V1. Therefore, it is preferable that the capacitance C1 of the capacitor C01 is within the ranges of Formula 7 and Formula 10. In this case, when the output of the vibration power generation device 14 is DC, the output capacitance of the vibration power generation device 14 is smaller than the capacitance C1. When the output of the vibration power generation device 14 is AC, the output capacitance of the vibration power generation device 14 via the rectifier circuit is smaller than C1.

Embodiment

[0062] Example 2 for solving the second problem will be described. In the power conversion circuits according to Example 1 and its modified examples, the period T1 and the period T2 are made constant, and the inductance L and the capacitance C1 are set to satisfy Formula 8 and Formula 11. However, when the inductance L and the capacitance C1 are set to satisfy Formula 8 and Formula 11, the switching frequency may become too high and the power consumption may increase. In the following Example 2, the period T1 and the period T2 do not have to be constant. Also, the inductance L and the capacitance C1 do not have to be set to satisfy Formula 8 and Formula 11.

[0063] Example 2 is an example of a buck DC-DC converter. FIG. 7 is a circuit diagram of the power conversion circuit according to Example 2. As shown in FIG. 7, the comparator 22 outputs a high level as the voltage Vdc when the voltage V1 is equal to or higher than the threshold voltage Vth, and outputs a low level as the voltage Vdc when the voltage V1 is smaller than the threshold voltage Vth. The Vth setting circuit 24 sets the threshold voltage Vth based on the value of the input current Iin detected by a current detection circuit described later with reference to FIG. 8. When the voltage Vdc becomes high level, the control circuit 20b starts generating the control signals S1 and S2. As a result, the voltage conversion circuit 10b starts the voltage conversion operation. The other circuit configurations are the same as those in FIG. 3(a) of Example 1 and the description thereof is omitted.

[0064] FIG. 8 is a circuit diagram showing the current detection circuit in Embodiment 2. As shown in FIG. 8, an NFET M1 is connected between a capacitor C01 and ground. The source of the NFET M1 is connected to ground, the drain is connected to node N4, and a power supply voltage VDD, for example 1V, is supplied to the gate. The NFET M1 functions as a shunt resistor. The current detection circuit 26 includes a plurality of diodes D1 to D8 and comparators 25a to 25c. The diodes D1 to D8 are connected in series in the forward direction between a power supply terminal to which the power supply voltage VDD is supplied and ground. Nodes N5a to N5c are provided at predetermined positions between the power supply terminal and ground. For example, the node N5a is located between the diode D1 and the power supply terminal, the node N5b is located between the diodes D4 and D5, and the node N5c is located between the diodes D7 and D8. The input terminals of the comparators 25a to 25c are connected to the node N4 and the nodes N5a to N5c, respectively. Signals Va to Vc output by the comparators 25a to 25c are output to the Vth setting circuit 24.

[0065] When the input current Iin flowing from the input terminal Tin to the node N1 flows into the capacitor C01, a current Im1 flows from the capacitor C01 to ground via the NFET M1 in order to balance the charge amount between the electrodes of the capacitor C01. If the current flowing from the capacitor C01 to the voltage conversion circuit is not considered, the current Im1 is substantially equal to the input current Iin. When the input current Iin increases, the potential V4 of the node N4 increases. The potential V5a of the node N5a is VDD, the potential V5b of the node N5b is 5 / 8 × VDD, and the potential V5c of the node N5c is 1 / 8 × VDD. Note that the potentials V5a to V5c can be arbitrarily set according to the power supply voltage, the number of the diodes D1 to D8, and the positions of the nodes N5a to N5c. The comparators 25a to 25c compare the potentials of the node N4 and the nodes N5a to N5c, respectively. The comparators 25a to 25c output high levels as the signals Va to Vc, respectively, when the potential of the node N4 is higher than the potentials of the nodes N5a to N5c. The comparators 25a to 25c output low levels as the signals Va to Vc, respectively, when the potential of the node N4 is lower than the potentials of the nodes N5a to N5c.

[0066] When the input current Iin is small and the potential V4 is lower than the potential V5c, all of the signals Va to Vc are at a low level. At this time, the Vth setting circuit 24 sets the threshold voltage Vth to the smallest Vth1. When the input current Iin increases and the potential V4 is between the potentials V5c and V5b, the signals Va and Vb are at a low level and the signal Vc is at a high level. At this time, the Vth setting circuit 24 sets the threshold voltage Vth to Vth2 which is higher than Vth1. When the input current Iin increases and the potential V4 is between the potentials V5b and V5a, the signal Va is at a low level and the signals Vb and Vc are at a high level. At this time, the Vth setting circuit 24 sets the threshold voltage Vth to Vth3 which is higher than Vth2. When the input current Iin increases and the potential V4 is higher than the potential V5a, all of the signals Va to Vc are at a high level. At this time, the Vth setting circuit 24 sets the threshold voltage Vth to Vth4 which is higher than Vth3.

[0067] By using the current detection circuit 26 of FIG. 8, the threshold voltage Vth can be set based on the input current Iin. The number of nodes N5a to N5c and the number of comparators 25a to 25c can be arbitrarily set.

[0068] FIG. 9 is a diagram showing the on and off of the switching element, each current and voltage with respect to time in the second embodiment. As shown in FIG. 9, an example in which the input current Iin increases uniformly will be described. The Vth setting circuit 24 generates Vth such that the threshold voltage Vth increases as the input current Iin increases. The Vth setting circuit 24 sets the threshold voltage Vth to Vth1 when the input current Iin is in the range of Iin1, sets the threshold voltage Vth to Vth2 when the input current Iin is in the range of Iin2, and sets the threshold voltage Vth to Vth3 when the input current Iin is in the range of Iin3. The threshold voltage Vth is set to be approximately Zout×Iin.

[0069] For example, when the input current Iin is within any of the ranges Iin1, Vth1 ≒ Zout × Iin. Similarly, when the input current Iin is within any of the ranges Iin2, Vth2 ≒ Zout × Iin, and when the input current Iin is within any of the ranges Iin3, Vth3 ≒ Zout × Iin.

[0070] The Vth setting circuit 24 (setting circuit) preferably sets the threshold voltage Vth such that 0.3 × Zout ≦ Vth / Iin ≦ 3 × Zout, and more preferably sets it such that 0.5 × Zout ≦ Vth / Iin ≦ 1.5 × Zout. Thereby, the input impedance Zin matches the output impedance Zout, and the power conversion efficiency is improved.

[0071] Until time t1, the control circuit 20b controls the switching elements SW1 and SW2 to be off. The currents I1 and I2 are 0. Since the input current Iin charges the capacitor C01, the voltage V1 gradually increases. At time t1, when the voltage V1 reaches the threshold voltage Vth, the voltage Vdc becomes high level. The control circuit 20b controls the switching element SW1 to be on and maintains the switching element SW2 off during the period T2 after the voltage Vdc becomes high level. Between time t1 and t2, the current I1 gradually increases. The voltage V1 gradually decreases.

[0072] At time t2, the control circuit 20b controls the switching element SW2 to be on and the switching element SW1 to be off. The current I1 becomes 0. Between time t2 and t3, the current I2 gradually decreases and the voltage V1 gradually increases. At time t3, the control circuit 20b controls the switching element SW2 to be off and maintains the switching element SW1 off. Between time t3 and t1´, the currents I1 and I2 are 0. The voltage V1 gradually increases. At time t1´, when the voltage V1 becomes the threshold voltage Vth, the voltage Vdc becomes high level.

[0073] When the input current Iin increases, the threshold voltage Vth becomes Vth2 which is greater than Vth1. When the input current Iin further increases, the threshold voltage Vth becomes Vth3 which is greater than Vth2. As a result, the voltage V1 increases when the input current Iin increases. By setting the threshold voltage Vth to be approximately the same as Zout×Iin, the voltage V1 becomes approximately Zout×Iin. Thus, even if the input current Iin fluctuates, the input impedance Zin = V1 / Iin can be matched to the output impedance Zout of the vibration power generation device 14. Therefore, the power conversion efficiency can be improved.

[0074] According to the second embodiment, when the input voltage V1 applied to the input terminal Tin reaches the threshold voltage Vth, the voltage conversion circuit 10b starts a voltage conversion operation of converting the input voltage V1 and outputting it as the output voltage V2 to the output terminal Tout. The threshold voltage Vth when the input current Iin input to the input terminal Tin is large is higher than the threshold voltage Vth when the input current Iin is small. Thus, since the threshold voltage Vth changes when the input current Iin fluctuates, the change in the input impedance Zin can be reduced even when the input current Iin fluctuates. Therefore, the input impedance Zin can be matched to the output impedance Zout of the vibration power generation device 14. Therefore, the power conversion efficiency can be improved.

[0075] By setting the threshold voltage Vth such that 0.3×Zout ≦ Vth / Iin ≦ 3×Zout, the input impedance Zin can be matched to the output impedance Zout of the vibration power generation device 14. It is more preferable that 0.5×Zout ≦ Vth / Iin ≦ 1.5×Zout.

[0076] As shown in FIG. 9 of Embodiment 2, the Vth setting circuit 24 sets the threshold voltage Vth such that the threshold voltage Vth2 in the range of Iin2 where the input current Iin is large is higher than the threshold voltage Vth1 in the range of Iin1 where the input current Iin is small, based on the input current Iin. The comparator 22 (determination circuit) determines whether or not the input voltage V1 has reached the threshold voltage Vth. The voltage conversion circuit 10b starts a voltage conversion operation based on the output of the comparator 22. Thereby, the input impedance Zin can be matched with the output impedance Zout of the vibration power generation device 14. Therefore, the power conversion efficiency can be improved.

[0077] As Embodiment 2, a buck DC-DC converter has been described as an example, but a boost DC-DC converter or a buck-boost DC-DC converter as shown in FIGS. 3(b) and 3(c) may also be used. That is, in the voltage conversion circuit 10b, one end of the inductor L0 is connected to the input terminal Tin, and the other end is connected to either one of the ends of the capacitor C02 (second capacitor). At least a part of the input current Iin flows as the current I1 (first current) through the inductor L0, and the magnetic field energy is accumulated. The capacitor C02 is charged by the current I2 (second current) induced by the magnetic field energy. When the switching element SW1 is turned on, the current I1 flows through the inductor L0, and when the current I2 flows through the inductor L0, the switching element SW1 is turned off.

[0078] When the input voltage V1 reaches the threshold voltage Vth, the control circuit 20b turns on the switching element SW1. Thereby, the voltage conversion operation of the voltage conversion circuit 10b is started.

[0079] In the second embodiment, an example in which the Vth setting circuit 24 sets the threshold voltage Vth based on the input current Iin has been described. However, the Vth setting circuit 24 may set the threshold voltage Vth based on information related to the input current Iin. For example, the amplitude information of the MEMS element correlates with the output current amount of the MEMS element. Therefore, the Vth setting circuit 24 may set the threshold voltage Vth based on the amplitude information of the MEMS element. For example, the MEMS element includes a main MEMS oscillator that outputs power as the input current of the power conversion circuit, and a sensing MEMS oscillator having an amplitude comparable to that of the main MEMS oscillator. The output current of the sensing MEMS oscillator is changed into a voltage signal by a shunt resistor or the like, which may be used as the amplitude information.

[0080] In the first embodiment and its modified example 1, the capacitance C1 of the capacitor C01 is limited. In the second embodiment, it can be applied to the capacitance C1 without limitation. Further, in the second embodiment, in order to actively match the impedance, it is easier to match the impedance than in the first embodiment and its modified example 1, and the power conversion efficiency is likely to be improved. On the other hand, in the second embodiment, since the comparator 22 is used, the power consumption is large, and when the power generated by the vibration power generation device 14 is small (for example, when it is 100 nW or less), it may be difficult to apply. In the first embodiment and its modified example 1, since the comparator 22 does not have to be used, the power consumption is small, and it is easy to apply when the power generated by the vibration power generation device 14 is small (for example, when it is 100 nW or less).

[0081] In the first and second embodiments and their modified examples, the vibration power generation device 14 has been described as an example of the power source. However, the power source may be other power generation devices or the like. The power conversion circuits of the first and second embodiments and their modified examples may be a SEPIC (Single-Ended Primary Inductor Converter ) , a ZETA circuit, a Cuk circuit, a boost chopper circuit, or a buck chopper circuit.

[0082] In Embodiments 1 and 2 and their modified examples, the switching elements SW1 to SW4 are transistors such as, for example, FET (Field Effect Transistor), IGBT (Insulated Gate Bipolar Transistor), or bipolar transistors. The control signals S1 to S4 are input to the gate or base. By the control signals S1 to S4, the source and drain or the emitter and collector of the switching elements SW1 to SW4 are turned on and off. The switching elements SW2 and SW4 may be diodes.

Example

[0083] Embodiment 3 is an example of a semiconductor device equipped with the power conversion circuits of Embodiments 1 and 2. Fig. 10(a) is a block diagram of the semiconductor device according to Embodiment 3. The semiconductor device 30 includes a power conversion circuit 32, an input section 34, a functional circuit 36, and an output section 38. The semiconductor device 30 is, for example, a semiconductor chip or a package on which a semiconductor chip is mounted. The power conversion circuit 32 is the power conversion circuit according to Embodiment 1 or 2. The input section 34 is, for example, a pad provided on a semiconductor chip or a terminal provided on a package, and power is supplied from an external device such as a vibration power generation device. The output section 38 is, for example, a pad provided on a semiconductor chip or a terminal provided on a package, and supplies the power converted by the power conversion circuit 32 to an external device. The functional circuit 36 is a circuit related to the power conversion circuit, and includes at least a part of, for example, control circuits 20, 20a, 20b for controlling the power conversion circuit, a Vth setting circuit 24, and a current detection circuit 26.

[0084] When the semiconductor device 30 is a semiconductor chip, transistors, capacitors, and inductors that constitute the switches of the power conversion circuit 32 are formed on a semiconductor substrate such as a silicon substrate. Also, at least a part of the transistors, diodes, and passive elements that constitute the functional circuit 36 are formed on the semiconductor substrate.

[0085] When the semiconductor device 30 is a package on which a semiconductor chip is mounted, at least part of transistors, diodes, and passive elements that constitute switches of the power conversion circuit 32, transistors that constitute the control circuits 20, 20a, 20b, the Vth setting circuit 24, and the current detection circuit 26 as the functional circuit 36 are formed on the semiconductor chip. At least part of elements (for example, inductors and capacitors) that are not formed on the semiconductor chip among those of the power conversion circuit 32 may be mounted on the package.

[0086] [Modification Example 1 of Embodiment 3] Modification Example 1 of Embodiment 3 is an example of an electronic device equipped with the power conversion circuits of Embodiments 1 and 2. FIG. 10(b) is a block diagram of the electronic device according to Modification Example 1 of Embodiment 3. The electronic device 40 is an electronic device that functions by an ambient power generation device such as a vibration power generation device, and is an ambient measurement device such as a street lamp, a display board, or a rain gauge.

[0087] The electronic device 40 includes the power conversion circuits 42 of Embodiments 1 and 2, an input unit 44, a functional unit 46, and a capacitor 48. The power conversion circuit 42 is the voltage conversion circuit 10, 10a, or 10b according to Embodiment 1 or 2. The input unit 44 is, for example, a terminal, and power is supplied from an external device such as a vibration power generation device. The capacitor 48 is, for example, a secondary battery or a capacitor, and stores the power converted by the power conversion circuit 42. The power converted by the power conversion circuit 42 or the power stored in the capacitor 48 is supplied to the functional unit 46, which exhibits its function.

[0088] As in Embodiment 3, the power conversion circuit 32 may be mounted on the semiconductor device 30. As in Modification Example 1 of Embodiment 3, the electronic device 40 may include the power conversion circuit 42. Thereby, the power efficiency of the electronic device 40 is improved.

[0089] As described above, the preferred embodiments of the present invention have been described in detail. However, the present invention is not limited to such specific embodiments, and various modifications and changes are possible within the scope of the gist of the present invention described in the claims.

Description of Reference Numerals

[0090] 10, 10a, 10b Voltage conversion circuits 12 Rectifier circuit 14 Vibration power generator 20, 20a, 20b Control circuits 22 Comparator 24 Vth setting circuit 30 Semiconductor device 36 Functional circuit 40 Electronic device 46 Functional unit

Claims

[

1. ] An input terminal to which an input current generated by a vibration power generation device is input, An output terminal that outputs an output voltage, A first capacitor having one end connected to the input terminal and the other end connected to a reference potential, A second capacitor having one end connected to the output terminal and the other end connected to a reference potential, One end is connected to one end of the first capacitor, the other end is connected to one end or the other end of the second capacitor, and at least a part of the input current and the current output from the first capacitor flows as a first current, so that magnetic field energy is accumulated, and an inductor that induces a second current for charging the second capacitor by the magnetic field energy, A switching element that turns on and off at a substantially constant cycle, the period of turning on during one cycle is substantially constant, the first current flows through the inductor when it turns on, and it turns off when the second current flows through the inductor, comprising When the capacitance of the first capacitor is C1, the inductance of the inductor is L, the period is T1, the length of the period during which the switching element is on during one cycle is T2, and the angular frequency of the input current is ω, C1 < (T2^2 / (2×L)) / (ω×T1), and a power conversion circuit in which the input impedance is matched to the output impedance of the vibration power generation device. [

2. ] The power conversion circuit according to claim 1, wherein C1 > T2^2 / (2×L). [

3. ] The power conversion circuit according to claim 1 or 2, wherein C1 < (T2^2 / (2×L)) / (5×ω×T1). [

4. ] The power conversion circuit according to any one of claims 1 to 3, wherein C1 > 5×T2^2 / (2×L). [

5. ] When the output of the vibration power generation device is DC, the output capacitance of the vibration power generation device is smaller than C1, and when the output of the vibration power generation device is AC, the output capacitance of the vibration power generation device via a rectifier circuit is smaller than C1. The power conversion circuit according to any one of claims 1 to 4. [

6. ] When the input voltage applied to the input terminal reaches a threshold voltage, a voltage conversion operation for converting the input voltage and outputting it as an output voltage to the output terminal is started, The threshold voltage when the input current input to the input terminal is large is higher than the threshold voltage when the input current is small, The power conversion circuit according to claim 1. [

7. ] A setting circuit that sets the threshold voltage such that the threshold voltage when the input current is large is higher than the threshold voltage when the input current is small, based on the input current input to the input terminal; A determination circuit that determines whether or not the input voltage has reached the threshold voltage; Comprising: The power conversion circuit according to claim 6, wherein the voltage conversion operation is started based on the output of the determination circuit.

8. The power conversion circuit according to claim 6 or 7, wherein when the output impedance is Zout and the threshold voltage when the input current input to the input terminal is Iin is Vth, 0.3×Zout ≦ Vth / Iin ≦ 3×Zout.

9. A power conversion circuit according to any one of claims 1 to 8; A functional circuit related to the power conversion circuit; A semiconductor device comprising:

10. A power conversion circuit according to any one of claims 1 to 8; A functional unit to which power is supplied from the power conversion circuit; An electronic device comprising:

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