Switching power supply device, its control device, and control method

Abstract

To provide a switching power supply device, and a control device and a control method thereof, which can adjust an optimum circuit operation mode to improve power conversion efficiency.SOLUTION: A DAB type DC / DC converter includes: an inverter 10 having switches 11.. that perform on / off operation by drive pulses S11..; a transformer 20; an inverter 30 having switches 31.. that perform on / off operation by drive pulses S31..; and a control device 40 that generates the drive pulses S11.., S31... The control device 40 generates a variable pulse pattern having duty ratios D1 and D2 for turning on all of the switches 11.., 31.. on an H side and / or all of the switches 12.., 32.. on an L side at the same time, and adds the pulse pattern having a phase difference φ1 to the drive pulses S11.., and / or the pulse pattern having a phase difference φ2 to the drive pulses S31...SELECTED DRAWING: Figure 1

Description

Technical Field

[0001] The present invention relates to a switching power supply device such as a dual active bridge (hereinafter referred to as "DAB") type DC / DC converter, and its control device and control method.

Background Art

[0002] Conventionally, a DAB type DC / DC converter, which is one of switching power supply devices, is a DC / DC converter capable of bidirectional power transmission, for example, by phase shift control of full-bridge inverters on the primary side and secondary side of a transformer as described in Patent Document 1.

[0003] FIG. 5 is a configuration diagram of a conventional three-phase DAB type DC / DC converter (hereinafter referred to as "conventional DAB") described in Patent Document 1, for example. In this conventional DAB, a primary-side inverter 10 is connected in parallel to a primary-side smoothing capacitor 1 that smooths a DC primary-side voltage E1 and a primary-side current I1. The primary-side inverter 10 is a circuit that switches the smoothed primary-side voltage E1 and primary-side current I1 to convert them into a three-phase AC voltage and a three-phase AC current, and is constituted by a full-bridge circuit of a high-level (hereinafter referred to as "H") side switch 11 of the U phase, a low-level (hereinafter referred to as "L") side switch 12 of the U phase, an H side switch 13 of the V phase, an L side switch 14 of the V phase, an H side switch 15 of the W phase, and an L side switch 16 of the W phase. The primary winding of a three-phase transformer 20 is connected to the connection points between switches 11 and 12, the connection points between switches 13 and 14, and the connection points between switches 15 and 16 via three-phase reactors 17, 18, and 19.

[0004] A secondary inverter 30 is connected to the secondary winding of the three-phase transformer 20. Note that the black circles attached near the upper ends of the primary winding and the secondary winding of the three-phase transformer 20 indicate the start of winding. The secondary inverter 30 is a circuit that rectifies the three-phase AC voltage and three-phase AC current output from the secondary winding of the three-phase transformer 20, and is composed of a full-bridge circuit of an H-side switch 31 of the U phase, an L-side switch 32 of the U phase, an H-side switch 33 of the V phase, an L-side switch 34 of the V phase, an H-side switch 35 of the W phase, and an L-side switch 36 of the W phase. The DC voltage and DC current rectified by the secondary inverter 30 are smoothed by the secondary smoothing capacitor 37, and the smoothed DC secondary voltage E2 and secondary current I2 are output.

[0005] The switches 11 to 16 and 31 to 36 that constitute the primary inverter 10 and the secondary inverter 30 are elements that are turned on / off by drive pulses S11 to S16 and S31 to S36 supplied from a control device (not shown), and are composed of power semiconductor elements such as metal-oxide-semiconductor field-effect transistors (hereinafter referred to as "MOSFETs") and insulated gate bipolar transistors (hereinafter referred to as "IGBTs"). A regeneration diode is connected in anti-parallel to each of the switches 11 to 16 and 31 to 36.

[0006] The drive pulses S11 to S16 and S31 to S36 that drive the power conversion unit of the conventional DAB in FIG. 5 have a constant frequency ω, a fixed duty ratio D of 0.5, and a phase difference β of 120° for each of the U, V, and W phases.

[0007] The primary-side inverter 10 converts the DC primary-side voltage E1 and primary-side current I1 into a three-phase AC voltage vp (hereinafter referred to as "output voltage vp") and a three-phase AC current by turning on and off switches 11 to 16 according to primary-side drive pulses S11 to S16 supplied from a control device (not shown). The secondary-side inverter 30 converts the three-phase AC voltage vs (hereinafter referred to as "input voltage vs") and three-phase AC current induced in the secondary winding of the three-phase transformer 20 into a DC secondary-side voltage E2 and secondary-side current I2 by turning on and off switches 31 to 36 according to secondary-side drive pulses S31 to S36 supplied from a control device (not shown).

[0008] The main phase difference φ between the output voltage vp (or output current) of the primary-side inverter 10 and the input voltage vs (or input current) of the secondary-side inverter 30 can control the input voltage (or input current), output voltage (or output current), and power flow. When the voltage vl (= vp - vs) between the primary winding and the secondary winding of the three-phase transformer 20 passes through reactors 17 to 19, a transformer current (which is the same as the reactor current IL, i.e., the U-phase reactor current IL_U, V-phase reactor current IL_V, and W-phase reactor current IL_W) flows. At this time, the U-phase reactor voltage VL_U, V-phase reactor voltage VL_V, and W-phase reactor voltage VL_W, which are the reactor voltages VL, are generated. The output power Po can be calculated from the effective value ILT of the reactor current IL as shown in the following equation (1).

[0009] Po = Nps(E1·2 / ωL)dφ[1 - (φ / π)] = Nps(E1·2 / ωL)dφ[φ - (φ·2 / π)] = Nps(E1·ILT)(E2 / E1)φ[φ - (φ·2 / π)] = Nps·ILT·E2·φ[φ - (φ·2 / π)] (1) However, L: the inductance of reactors 17 to 19 ω: frequency Nps: the winding ratio (or transformation ratio) between the primary winding and the secondary winding of the three-phase transformer 20 E1: primary-side voltage E2: secondary-side voltage d: secondary-side / primary-side voltage ratio (= E2 / E1) φ; The main phase difference between the output voltage vp (or output current) of the primary inverter 10 and the input voltage vs (or input current) of the secondary inverter 30 side inverter 30 phase difference ILT; The effective value of the reactor current IL Thus, in the conventional DAB, by changing the main phase difference φ, step-up / step-down operation and bidirectional power conversion can be easily achieved.

Prior Art Documents

Patent Documents

[0010]

Patent Document 1

Patent Document 2

Summary of the Invention

Problems to be Solved by the Invention

[0011] In the conventional DAB shown in FIG. 5 described in Patent Document 1, when the input / output voltage ratio (for example, the ratio of the secondary voltage E2 to the primary voltage E1) is large, the current circulating in the circuit with respect to the output current (for example, the secondary current I2) increases, mainly resulting in an increase in conduction loss and a tendency for the power conversion efficiency to decrease. As a technique for solving this problem, a control method for a single-phase DAB type DC / DC converter described in Patent Document 2 is known.

[0012] FIG. 6 is a configuration diagram of a conventional single-phase DAB type DC / DC converter described in Patent Document 2. This single-phase DAB type DC / DC converter has a primary-side inverter 10A connected in parallel to a primary-side smoothing capacitor 1 that smooths the DC primary-side voltage E1 and primary-side current I1. The primary-side inverter 10A is a circuit that switches the smoothed primary-side voltage E1 and primary-side current I1 to convert them into a single-phase AC voltage and a single-phase AC current, and is composed of a full-bridge circuit of an H-side switch 11 of arm A, an L-side switch 12 of arm A, an H-side switch 13 of arm B, and an L-side switch 14 of arm B. The primary winding of a single-phase transformer 20A is connected to the connection points between switches 11 and 12 and between switches 13 and 14 via single-phase reactors 17 and 18.

[0013] A secondary-side inverter 30A is connected to the secondary winding of the single-phase transformer 20A. The secondary-side inverter 30A is a circuit that rectifies the AC voltage and AC current output from the secondary winding of the single-phase transformer 20A, and is composed of a full-bridge circuit of an H-side switch 31 of arm A, an L-side switch 32 of arm A, an H-side switch 33 of arm B, and an L-side switch 34 of arm B. The DC voltage and DC current rectified by the secondary-side inverter 30A are smoothed by a secondary-side smoothing capacitor 37, and smoothed DC secondary-side voltage E2 and secondary-side current I2 are output.

[0014] The switches 11 to 14, 31 to 34 that constitute the primary-side inverter 10A and the secondary-side inverter 30A are turned on / off respectively by drive pulses S11 to S14, S31 to S34 supplied from a control device (not shown). In the control method of this single-phase DAB type DC / DC converter, the above problems of Patent Document 1 are solved by performing phase shift control to change the phase difference β1 between the arm A side switches 11, 12 and the arm B side switches 13, 14 in the primary-side inverter 10A, or by performing phase shift control to change the phase difference β2 between the arm A side switches 31, 32 and the arm B side switches 33, 34 in the secondary-side inverter 30A.

[0015] However, in the control method of the conventional DAB in Patent Document 1, since the phase difference β is fixed at 120° between the U, V, and W phases, it is difficult to apply the phase shift control for changing the phase difference β1 or β2 as in Patent Document 2. Therefore, the inventor of the present application previously filed a patent application to solve the above problems of Patent Document 1 (filing date: December 21, 2018, application number: Japanese Patent Application No. 2018-239182, unpublished, hereinafter referred to as the "previous patent application"). FIG. 7 is a pattern diagram (φ = 0°, D = D1 = D2) showing an example of drive pulses without dead time (a state in which the H-side switches and the L-side switches are alternately turned on / off) of the previous patent application using the conventional DAB of FIG. 5. Further, FIG. 8 is a pattern diagram (φ ≠ 0°, D = D1 = D2) showing an example of drive pulses without dead time of the previous patent application using the conventional DAB of FIG. 5.

[0016] In FIGS. 7 and 8, switches 11 to 16 are turned on by H of the primary-side drive pulses S11 to S16, and switches 11 to 16 are turned off by L of the primary-side drive pulses S11 to S16. Similarly, switches 31 to 36 are turned on by H of the secondary-side drive pulses S31 to S36, and switches 31 to 36 are turned off by L of the secondary-side drive pulses S31 to S36. The frequency ω of the primary-side drive pulses S11 to S16 and the secondary-side drive pulses S31 to S36 is constant. Each of the U, V, and W phases has a phase difference β of 120°. The main phase difference φ changes according to the output command value as a DAB.

[0017] In the previous patent application, as shown in FIGS. 7 and 8, a pulse pattern of the primary duty ratio D1 that turns on all of the H-side switches 11, 13, 15 and / or all of the L-side switches 12, 14, 16 in the primary inverter 10 simultaneously is added to the primary drive pulses S11 to S16, and / or a pulse pattern of the secondary duty ratio D2 (= D1) that turns on all of the H-side switches 31, 33, 35 and / or all of the L-side switches 32, 34, 36 in the secondary inverter 30 simultaneously is added to the secondary drive pulses S31 to S36. The pulse pattern of the primary duty ratio D1 is arranged, for example, at the center of the half cycle (T / 2) of the primary drive pulses S11 to S16 (the first phase difference φ1 at this time is 0°). Similarly, the pulse pattern of the secondary duty ratio D2 is arranged, for example, at the center of the half cycle (T / 2) of the secondary drive pulses S31 to S36 (the second phase difference φ2 at this time is 0°).

[0018] FIG. 9 is a waveform diagram (φ ≠ 0°, φ1 = φ2 = 0°, D = D1 = D2) showing an example of the operation of the previous patent application corresponding to FIG. 8. The parameters of the waveform diagram in FIG. 9 are, for example, as follows. Primary voltage E1 = secondary voltage E2, primary duty ratio D1 = secondary duty ratio D2 = D = 0.95, winding ratio Nps of the primary winding and the secondary winding of the three-phase transformer 20 = 1:1, main phase difference φ = 28°, and first phase difference φ1 = second phase difference φ2 = 0°. By changing the duty ratio D (= D1 = D2) added to the drive pulses S11 to S16, S31 to S36, an increase in the current circulating in the circuit with respect to the secondary current I2, which is the output current (for example, the effective value of the U-phase reactor current IL_U), is reduced.

[0019] However, even when controlling the output power Po (voltage · current) required for the DAB type DC / DC converter and controlling (suppressing) the current circulating inside the circuit by a total of two adjustment elements, namely, the output adjustment element based on the main phase difference φ described in Patent Document 1 and the adjustment element for adding the pulse pattern of the duty ratio D (D1, D2) described in the previous patent application, there are cases where it is not possible to optimally control (suppress) the suppression of the current circulating inside the circuit depending on the voltage ratio d (= E2 / E1) of the input and output and the situation of the output power Po.

Means for Solving the Problems

[0020] The switching power supply device of the present invention includes a primary side inverter in which a plurality of H-side and L-side switches are fully bridge-connected, switching a DC primary side voltage and a primary side current to convert them into an AC voltage and an AC current and outputting them, a transformer having a primary winding and a secondary winding, inputting the output voltage and output current of the primary side inverter to the primary winding, and outputting the induced AC voltage and AC current from the secondary winding, and a secondary side inverter in which a plurality of H-side and L-side switches are fully bridge-connected, rectifying the output voltage and output current of the secondary winding, and outputting a DC secondary side voltage and a secondary side current, and a control device.

[0021] The control device outputs a plurality of primary side drive pulses to turn on / off the plurality of H-side and L-side switches in the primary side inverter respectively, and outputs a plurality of secondary side drive pulses to turn on / off the plurality of H-side and L-side switches in the secondary side inverter respectively, and controls the output power of the secondary side inverter by changing the main phase difference between the output value of the primary side inverter and the input value of the secondary side inverter.

[0022] In the control device of the present invention, when all of the L-side switches are in the off state, all of the H-side switches corresponding to the low level side switches are turned on, and when all of the H-side switches are in the off state, all of the L-side switches corresponding to the H-side switches are turned on, duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2)Generate a pulse pattern, and among a first phase difference arranged at a position shifted by φ1 from the center C1 of the period (T / 2) of the primary-side drive pulse and a second phase difference arranged at a position shifted by φ2 from the center C2 of the period (T / 2) of the secondary-side drive pulse, add the pulse pattern having the first phase difference to the plurality of primary-side drive pulses and / or add the pulse pattern having the second phase difference to the plurality of secondary-side drive pulses, and the duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2) The pulse pattern is generated based on the command value of the main phase difference and the primary / secondary voltage ratio of the primary-side voltage and the secondary-side voltage, and the first phase difference and the second phase difference in the pulse pattern are set based on the transformer current flowing through the transformer, the secondary-side voltage, and the secondary-side current. In the control device, an error e between a secondary-side current command value and a secondary-side current is obtained, the error e is corrected by proportional-integral control or the like, an effective value ILT of reactor current is calculated from the measured transformer current, and a main phase difference φ between the output voltage vp of the primary-side inverter and the input voltage vp of the secondary-side inverter is calculated so as to make the corrected error e zero, and the duty ratio of the pulse pattern may be calculated 。 The duty ratio of the pulse pattern is preferably a value in the range of 0 to 1. The primary-side inverter, the secondary-side inverter, and the transformer may be configured to perform single-phase, three-phase, or four-phase or more power conversion

[0023] The control device of the switching power supply device of the present invention is a control device of a switching power supply device including a primary-side inverter in which a plurality of H-side and L-side switches are full-bridge connected, switching a DC primary-side voltage and a primary-side current to convert them into an AC voltage and an AC current and outputting them, a transformer having a primary winding and a secondary winding, inputting the output voltage and output current of the primary-side inverter to the primary winding, and outputting the induced AC voltage and AC current from the secondary winding, and a secondary-side inverter in which a plurality of H-side and L-side switches are full-bridge connected, rectifying the output voltage and output current of the secondary winding, and outputting a DC secondary-side voltage and a secondary-side current.

[0024] Then, in the control device of the present invention, when controlling the output power of the secondary-side inverter by changing the main phase difference between the output value of the primary-side inverter and the input value of the secondary-side inverter by outputting a plurality of primary-side drive pulses to turn on / off the plurality of H-side and L-side switches in the primary-side inverter respectively, and outputting a plurality of secondary-side drive pulses to turn on / off the plurality of H-side and L-side switches in the secondary-side inverter respectively, when all of the L-side switches are in the off state, all of the H-side switches corresponding to the L-side switches are turned on, and when all of the H-side switches are in the off state, all of the L-side switches corresponding to the H-side switches are turned on, duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2) of the pulse pattern is generated, and among the first phase difference arranged at a position shifted by φ1 from the center C1 of the period (T / 2) of the primary-side drive pulse and the second phase difference arranged at a position shifted by φ2 from the center C2 of the period (T / 2) of the secondary-side drive pulse, the pulse pattern having the first phase difference is added to the plurality of primary-side drive pulses, and / or the pulse pattern having the second phase difference is added to the plurality of secondary-side drive pulses, the duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2) of the pulse pattern is generated based on the command value of the main phase difference and the primary-side / secondary-side voltage ratio of the primary-side voltage and the secondary-side voltage, and the first phase difference and the second phase difference in the pulse pattern are set based on the transformer current flowing through the transformer, the secondary-side voltage, and the secondary-side current.

[0025] Furthermore, the control method of the switching power supply device of the present invention includes a primary-side inverter in which a plurality of H-side and L-side switches are full-bridge connected, and the DC primary-side voltage and primary-side current are switched to convert them into an AC voltage and an AC current for output; a transformer having a primary winding and a secondary winding, where the output voltage and output current of the primary-side inverter are input to the primary winding, and the induced AC voltage and AC current are output from the secondary winding; and a secondary-side inverter in which a plurality of H-side and L-side switches are full-bridge connected, and the output voltage and output current of the secondary winding are rectified to output a DC secondary-side voltage and secondary-side current. It is a control method of a switching power supply device comprising these components.

[0026] And in the control method of the present invention, when outputting a plurality of primary-side drive pulses to turn on / off the plurality of H-side and L-side switches in the primary-side inverter respectively, and outputting a plurality of secondary-side drive pulses to turn on / off the plurality of H-side and L-side switches in the secondary-side inverter respectively, and controlling the output power of the secondary-side inverter by changing the main phase difference between the output value of the primary-side inverter and the input value of the secondary-side inverter, when all of the L-side switches are in the off state, all of the H-side switches corresponding to the L-side switches are turned on, and when all of the H-level side switches are in the off state, all of the L-side switches corresponding to the H-level side switches are turned on, a pulse pattern of the duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2) is generated, and among the first phase difference arranged at a position shifted by φ1 from the center C1 of the period (T / 2) of the primary-side drive pulse and the second phase difference arranged at a position shifted by φ2 from the center C2 of the period (T / 2) of the secondary-side drive pulse, the pulse pattern having the first phase difference is added to the plurality of primary-side drive pulses, and / or the pulse pattern having the second phase difference is added to the plurality of secondary-side drive pulses, and the duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2)Generate the pulse pattern based on the command value of the main phase difference and the primary / secondary voltage ratio of the primary-side voltage and the secondary-side voltage, and set the first phase difference and the second phase difference in the pulse pattern based on the transformer current flowing through the transformer, the secondary-side voltage, and the secondary-side current 。

Advantages of the Invention

[0027] According to the switching power supply device, its control device, and the control method of the present invention, when all of the switches on the L side are in the off state, all of the switches on the H side corresponding to the switches on the L side are turned on, and when all of the switches on the H side are in the off state, all of the switches on the L side corresponding to the switches on the H side are turned on, the duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2) Generate a pulse pattern of, and add the pulse pattern having a first phase difference arranged at a position shifted by φ1 from the center C1 of the period (T / 2) of the primary-side drive pulse to a plurality of primary-side drive pulses, and / or add the pulse pattern having a second phase difference arranged at a position shifted by φ2 from the center C2 of the period (T / 2) of the secondary-side drive pulse to a plurality of secondary-side drive pulses, the duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2) Generate the pulse pattern based on the command value of the main phase difference and the primary / secondary voltage ratio of the primary-side voltage and the secondary-side voltage, and set the first phase difference and the second phase difference in the pulse pattern based on the transformer current flowing through the transformer, the secondary-side voltage, and the secondary-side current 。 Therefore, similar to the previous patent application, it is only necessary to add a modulation method for changing the first and second phase differences without changing the configuration of the power conversion unit from the conventional DAB. As a result, it is possible to prevent an increase in hardware cost and adjust to a more optimal circuit operation mode than the conventional DAB. As a result, the power conversion efficiency can be improved by reducing the power loss. In particular, since the switching power supply device of the present invention operates bidirectionally, the same effect as described above can be achieved even when the input / output operation is reversed

Brief Description of the Drawings

[0028]

Figure 1

Figure 2

Figure 3

Figure 4

Figure 5

Figure 6

Figure 7

Figure 8

Figure 9

Embodiments for Carrying Out the Invention

[0029] The embodiments for carrying out the present invention will become apparent when reading the following description of preferred embodiments in conjunction with the accompanying drawings. However, the drawings are for illustrative purposes only and do not limit the scope of the present invention.

Examples

[0030] (Configuration of Example 1) FIG. 1 is a configuration diagram of the three-phase DAB type DC / DC converter in Embodiment 1 of the present invention. The three-phase DAB type DC / DC converter of the first embodiment includes a power conversion unit having a primary smoothing capacitor 1, a primary inverter 10, three-phase reactors 17, 18, 19, a three-phase transformer 20, a secondary inverter 30, and a secondary smoothing capacitor 37, which are the same as those of the conventional DAB, and a control device 40 different from the conventional one.

[0031] Similar to the conventional DAB, the switches 11 to 16, 31 to 36 that make up the primary inverter 10 and the secondary inverter 30 are elements that are turned on / off by the primary drive pulses S11 to S16 and the secondary drive pulses S31 to S36 supplied from the control device 40, and are composed of power semiconductor elements such as MOSFETs and IGBTs. A regeneration diode is connected in anti-parallel to each of the switches 11 to 16, 31 to 36. When each of the switches 11 to 16, 31 to 36 is composed of a MOSFET, for example, the parasitic capacitance of the MOSFET may be used.

[0032] In addition, three-phase reactors are connected in series to the primary winding and the secondary winding of the three-phase transformer 20, respectively. These reactors may be replaced by the leakage inductance of the three-phase transformer 20. In FIG. 1, for the sake of simplifying the illustration, reactors 17, 18, 19 are connected in series to the primary winding side of the three-phase transformer 20, respectively.

[0033] The control device 40 outputs six primary-side drive pulses S11 to S16 to turn on / off the six H-side switches 11, 13, 15 and L-side switches 12, 14, 16 in the primary-side inverter 10 respectively, and outputs six secondary-side drive pulses S31 to S36 to turn on / off the six H-side switches 31, 33, 35 and L-side switches 32, 34, 36 in the secondary-side inverter 30 respectively, and controls the output power Po of the secondary-side inverter 30 by changing the main phase difference φ between the output value (for example, output voltage vp or output current) of the primary-side inverter 10 and the input value (for example, input voltage vs or input current) of the secondary-side inverter 30. Each of the switches 11 to 16, 31 to 36 is turned on by H and turned off by L of the respective drive pulses S11 to S16, S31 to S36.

[0034] In particular, in the control device 40 of the first embodiment, a duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2) that turns on all of the H-side switches 11, 13, 15 and 31, 33, 35 and / or all of the L-side switches 12, 14, 16 and 32, 34, 36 simultaneously of generates a pulse pattern, and adds the pulse pattern having the first phase difference φ1 among the first phase difference φ1 and the second phase difference φ2 shifted from a certain phase to the six primary-side drive pulses S11 to S16, and / or adds the pulse pattern having the second phase difference φ2 to the six secondary-side drive pulses S31 to S36. The primary-side duty ratio D1 and the secondary-side duty ratio D2 are values in the range of 0 to 1.

[0035] The control device 40 generates a pulse pattern based on, for example, a command value of the main phase difference φ and a secondary / primary voltage ratio d (= E2 / E1) of the primary-side voltage E1 and the secondary-side voltage E2. Further, based on the transformer current flowing through the three-phase transformer 20 (the same as the U-phase reactor current IL_U, the V-phase reactor current IL_V, and the W-phase reactor current IL_W which are the reactor current IL), the secondary-side voltage E2, and the secondary-side current I2, the first phase difference φ1 and the second phase difference φ2 in the pulse pattern are set. In the three-phase reactors 17, 18, and 19, a U-phase reactor voltage VL_U, a V-phase reactor voltage VL_V, and a W-phase reactor voltage VL_W which are reactor voltages VL are generated respectively. Such a control device 40 is composed of, for example, a central processing unit (CPU) and individual circuits such as semiconductor elements.

[0036] (Control method of Embodiment 1) FIG. 2 is a pattern diagram showing an example of drive pulses S11 to S16, S31 to S36 without dead time in the steady state of FIG. 1 (φ = 0°, φ1 ≠ 0°, φ2 ≠ 0°, D = D1 = D2). Further, FIG. 3 is a pattern diagram showing an example of drive pulses S11 to S16, S31 to S36 without dead time in the steady state of FIG. 1 (φ ≠ 0°, φ1 ≠ 0°, φ2 ≠ 0°, D = D1 = D2).

[0037] As shown in FIGS. 2 and 3, switches 11 to 16 are turned on by the H of the primary side drive pulses S11 to S16, and are turned off by the L of the primary side drive pulses S11 to S16. The frequency ω of the primary side drive pulses S11 to S16 is constant, and the on / off duty ratio of each of the switches 11 to 16 is 0.5. As shown in the circuit modes (1) to (6) of one cycle (T), each of the U, V, and W phases has a phase difference β1 of 120° from each other. In each of the circuit modes (1) to (6), a pulse pattern of the primary side duty ratio D1 in which all the L-side switches 12, 14, and 16 are turned on, and a pulse pattern of the primary side duty ratio D1 in which all the H-side switches 11, 13, and 15 are turned on, are added. The pulse pattern of the primary side duty ratio D1 is arranged at a position shifted by a first phase difference φ1 from a certain phase (for example, the center C1 of the half cycle (T / 2) of the primary side drive pulses S11 to S16), as shown in FIG. 3.

[0038] Similarly, as shown in FIGS. 2 and 3, switches 31 to 36 are turned on by the H of the secondary side drive pulses S31 to S36, and are turned off by the L of the secondary side drive pulses S31 to S36. The frequency ω of the secondary side drive pulses S31 to S36 is constant, and the on / off duty ratio of each of the switches 31 to 36 is 0.5. As shown in the circuit modes (11) to (16) of one cycle (T), each of the U, V, and W phases has a phase difference β2 of 120° from each other. In each of the circuit modes (11) to (16), a pulse pattern of the secondary side duty ratio D2 in which all the L-side switches 32, 34, and 36 are turned on, and a pulse pattern of the secondary side duty ratio D2 in which all the H-side switches 31, 33, and 35 are turned on, are added. The pulse pattern of the secondary side duty ratio D2 is arranged at a position shifted by a second phase difference φ2 from a certain phase (for example, the center C2 of the half cycle (T / 2) of the secondary side drive pulses S31 to S36), as shown in FIG. 3. As shown in FIG. 3, a main phase difference φ as an output command value is provided between the primary side drive pulses S11 to S16 and the secondary side drive pulses S31 to S36.

[0039] Next, the normal constant current control operation (A1) when the primary-side voltage E1 and the secondary-side voltage E2 are close, and the constant current control operation (A2) in the case of a secondary-side short circuit at no load (main phase difference φ = 0°) will be described.

[0040] (A1) Normal constant current control operation when the primary-side voltage E1 and the secondary-side voltage E2 are close For example, in the three-phase DAB type DC / DC converter of FIG. 1, when a DC primary-side current I1 is input and a constant DC secondary-side current I2 is supplied to a load (not shown) connected to the output side of the secondary-side smoothing capacitor 37, the normal constant current control operation when the primary-side voltage E1 and the secondary-side voltage E2 are close will be described. In this normal constant current control operation, ideally, pulse patterns of duty ratios D1 and D2 are not added to the circuit modes (1) to (6), (11) to (16) of FIG. 3.

[0041] In the control device 40 of FIG. 1, for example, when the measured secondary-side current I2 fluctuates with respect to the secondary-side current command value Ith2, an error e between the secondary-side current command value Ith2 and the secondary-side current I2 is obtained, and the error e is corrected by proportional-integral control (PI control) or the like. Further, the effective value ILT of the reactor current is calculated from the measured transformer current (the same as the reactor current IL). The control device 40 calculates the main phase difference φ between the output voltage vp (or output current) of the primary-side inverter 10 and the input voltage vp (or input current) of the secondary-side inverter 30 so as to make the corrected error e zero, and when the primary-side / secondary-side voltage ratio 1 / d (= E1 / E2) is approximately 1, calculates the primary-side duty ratio D1 = the secondary-side duty ratio D2 = 0. Then, primary-side drive pulses S11 to S16 are generated to turn on / off the switches 11 to 16 in the primary-side inverter 10, and secondary-side drive pulses S31 to S36 are generated to turn on / off the switches 31 to 36 in the secondary-side inverter 30.

[0042] When the circuit mode (1) of FIG. 3 without the pulse pattern of the primary duty ratio D1 is added, the H-side switch 11 of the U-phase in the primary inverter 10 of FIG. 1 is off, the L-side switch 12 is on, the H-side switch 13 of the V-phase is on, the L-side switch 14 is off, the H-side switch 15 of the W-phase is off, and the L-side switch 16 is on. Further, when shifted by the main phase difference φ from the circuit mode (1) and the circuit mode (11) of FIG. 3 without the pulse pattern of the secondary duty ratio D2 is added, the H-side switch 31 of the U-phase in the secondary inverter 30 of FIG. 1 is off, the L-side switch 32 is on, the H-side switch 33 of the V-phase is on, the L-side switch 34 is off, the H-side switch 35 of the W-phase is off, and the L-side switch 36 is on.

[0043] Then, in FIG. 1, the primary current I1 flows through the path of the + side of the primary voltage E1 source → the H-side switch 13 → the reactor 18 → the primary winding of the three-phase transformer 20 → the reactor 17 → the L-side switch 12 → the - side of the primary voltage E1 source, and the path of the + side of the primary voltage E1 source → the H-side switch 13 → the reactor 18 → the primary winding of the three-phase transformer 20 → the reactor 19 → the L-side switch 16 → the - side of the primary voltage E1 source. Correspondingly, an induced electromotive force is generated in the secondary winding of the three-phase transformer 20, and the secondary current I2 flows through the path of the secondary winding of the three-phase transformer 20 → the diode of the H-side switch 33 → the load → the diode of the L-side switch 32 → the secondary winding, and the path of the secondary winding of the three-phase transformer 20 → the diode of the H-side switch 33 → the load → the diode of the L-side switch 36 → the secondary winding.

[0044] Furthermore, the other circuit modes (2) to (6), (12) to (16) of FIG. 3 without the pulse pattern of the primary duty ratio D1 are executed, and the switching operation for one period (T) is completed.

[0045] Here, since the voltage vl (= vp - vs) between the primary winding and the secondary winding of the three-phase transformer 20 passes through the reactors 17 to 19, a transformer current (the same as the reactor current IL) flows. The output power Po can be calculated from the effective value ILT of the reactor current IL in the same manner as in the above formula (1). Po = Nps·ILT·E2·φ[φ - (φ2 / (π)) However, L is the inductance of reactors 17 to 19 ω is the frequency Nps is the turns ratio (or voltage ratio) between the primary winding and the secondary winding of the three-phase transformer 20 E1 is the primary-side voltage E2 is the secondary-side voltage d is the secondary-side / primary-side voltage ratio (= E2 / E1) φ is the main phase difference between the output voltage vp (or output current) of the primary-side inverter 10 and the input voltage vs (or input current) of the secondary-side inverter 30 between them ILT is the effective value of the reactor current IL

[0046] Therefore, due to the main phase difference φ between the output voltage vp (or output current) of the primary-side inverter 10 and the input voltage vs (or input current) of the secondary-side inverter 30, the secondary-side current I2 is controlled in a constant current manner so as to match the secondary-side current command value Ith2 In addition, in the above normal constant current control operation, when the switching of the presence or absence of the duty ratio during operation is not performed, a small amount of duty ratios D1 and D2 may be added so that the influence of the duty ratio addition is reduced

[0047] (A2) Constant current control operation in the case of secondary-side short circuit at no load (main phase difference φ = 0°) For example, the constant current control operation when the secondary-side voltage E2 becomes 0 V (short-circuit state) due to load fluctuations will be described. In this operation, a pulse pattern of the primary-side duty ratio D1 is added during the circuit modes (1) to (6) in FIG. 2, and further, a pulse pattern of the secondary-side duty ratio D2 is added during the circuit modes (11) to (16)

[0048] When the secondary voltage E2 becomes 0V, the charge stored in the smoothing capacitor 37 is discharged, and an instantaneous pulsed large current is generated. After that, the secondary current I2 is maintained at a constant value. At this time, the control device 40 calculates the main phase difference φ = 0°, and since the primary / secondary voltage ratio 1 / d (= E1 / E2) is the maximum value E1, the desired primary duty ratio D1 and secondary duty ratio D2 (for example, D1 = D2 = 0.66) are calculated. Then, the control device 40 generates primary drive pulses S11 to S16 with the primary duty ratio D1 added, turns on / off the switches 11 to 16 in the primary inverter 10, and further generates secondary drive pulses S31 to S36 with the secondary duty ratio D2 added, and turns on / off the switches 31 to 36 in the secondary inverter 30.

[0049] Then, due to the pulse pattern of the added primary duty ratio D1, in circuit mode (1) of FIG. 2, the H-side switch 13 of the V phase transitions from on to off, and the L-side switch 14 transitions from off to on, and all the L-side switches 12, 14, 16 are turned on. Further, due to the pulse pattern of the added primary duty ratio D1, in circuit mode (2), all the H-side switches 11, 13, 15 are turned on, in circuit mode (3), all the L-side switches 12, 14, 16 are turned on, in circuit mode (4), all the H-side switches 11, 13, 15 are turned on, in circuit mode (5), all the L-side switches 12, 14, 16 are turned on, and in circuit mode (6), all the H-side switches 11, 13, 15 are turned on.

[0050] Similarly, due to the pulse pattern of the added secondary duty ratio D2, in circuit modes (11) to (16) of FIG. 2, all the L-side switches 32, 34, 36 are turned on, or all the H-side switches 31, 33, 35 are turned on. Therefore, when the primary duty ratio D1 and the secondary duty ratio D2 of the added pulse pattern are, for example, 0.66, the pulse patterns of the control method of the conventional DAB do not occur at all in the circuit modes (1) to (6), (11) to (16), and the current circulating in the circuit theoretically becomes 0 A. Therefore, it is possible to suppress the reactive current circulating in the circuit, which becomes particularly significant when the voltage difference between the input and output is large.

[0051] However, even when controlling the output power Po (voltage·current) required for the DAB type DC / DC converter and controlling (suppressing) the current circulating inside the circuit by a total of two adjustment elements, namely, the output adjustment element based on the main phase difference φ and the adjustment element for adding the pulse pattern of the primary duty ratio D1 and / or the secondary duty ratio D2, there may be cases where the suppression of the current circulating inside the circuit cannot be optimally controlled (suppressed) depending on the voltage ratio d (=E2 / E1) between the input and output and the situation of the output power Po.

[0052] Therefore, in the control device 40 of the first embodiment, a modulation method is added to change the first phase difference φ1 of the pulse pattern of the primary duty ratio D1 added to the primary drive pulses S11 to S16 and / or the second phase difference φ2 of the pulse pattern of the secondary duty ratio D2 added to the secondary drive pulses S31 to S36.

[0053] FIG. 4 is a waveform diagram (φ≠0°, φ1≠0°, φ2≠0°, D = D1 = D2) showing an example of the operation of FIG. 1. The parameters of this waveform diagram in FIG. 4 are, for example, as follows. Primary voltage E1 = secondary voltage E2, primary duty ratio D1 = secondary duty ratio D2 = D = 0.95, turns ratio Nps of the primary winding and the secondary winding of the three-phase transformer 20 = 1:1, main phase difference φ = +16°, and first phase difference φ1 = second phase difference φ2 = +51°. The switching period (T) is the same as in FIG. 9. The dead time is not inserted. The numerical values of these parameters are just an example, and only parameters with the same frequency and input / output power (current) and with characteristics in the waveform and being easy to understand are selected, not optimal values or the like.

[0054] Compare the waveform diagram of FIG. 9 in the previous patent application with the waveform diagram of FIG. 4 in the first embodiment. For the waveforms in both FIG. 9 and FIG. 4, the input-output voltage (i.e., the primary-side voltage E1 / secondary-side voltage E2) is the same (1:1), and the main phase difference φ, the first phase difference φ1, and the second phase difference φ2 are changed so that the output current value (I2) (and thus the input current value (I1) is also the same because it is an ideal circuit) is the same in FIG. 9 and FIG. 4. In FIG. 9 of the previous patent application, the first phase difference φ1 and the second phase difference φ2 do not exist, but in accordance with the representation in FIG. 4, φ1 = φ2 = 0° (fixed value).

[0055] It can be seen that the transformer current flowing through the three-phase transformer 20 (e.g., the same as the U-phase reactor current IL_U) is slightly reduced in FIG. 4 compared to FIG. 9. This means that the reactive current component circulating in the circuit is also reduced by the same value. Therefore, with the newly added controllable control parameters (the first phase difference φ1 and the second phase difference φ2) for the previous patent application, while obtaining the same output power Po (current), the degree of freedom for parameter adjustment is further improved.

[0056] (Mechanism of the first embodiment) The reason why the reactive current component circulating in the circuit can be suppressed more than the conventional DAB by changing (modulating) the first phase difference φ1 and the second phase difference φ2 will be explained in detail below.

[0057] In FIG. 7, the pattern diagrams of the drive pulses S11~S16, S31~S36 of the previous patent application are shown. At this time, the main phase difference φ is 0°, the primary-side duty ratio D1 and the secondary-side duty ratio D2 of the added pulse pattern are the same value, and furthermore, the primary-side voltage E1 and the secondary-side voltage E2 are the same. In the pattern diagram of FIG. 7, since the primary-side drive pulses S11 to S16 and the secondary-side drive pulses S31 to S36 are exactly the same, and the primary-side voltage E1 and the secondary-side voltage E2 are also the same, no reactor voltage VL_U is applied to the reactors 17 to 19 (for example, the reactor 17 of the U phase). Since no reactor voltage VL_U is applied across the reactor 17 of the U phase, no current flows through the reactor 17 of the U phase. Similarly, no current flows through the reactor 18 of the V phase and the reactor 19 of the W phase, which differ only in phase by 120°.

[0058] On the other hand, FIG. 2 shows a pattern diagram when the first phase difference φ1 and the second phase difference φ2 are changed in the first embodiment. In FIG. 2, a waveform example is shown when the phase is set so as to intentionally increase the reactor current flowing through the reactor 17 of the U phase.

[0059] In the waveform of FIG. 2, except for changing the first phase difference φ1 and the second phase difference φ2, the conditions are the same as those of the waveform of FIG. 7. It can be confirmed that a mode in which the reactor voltage VL_U is applied across the reactor 17 of the U phase appears with respect to the waveform of FIG. 7. Therefore, under this condition, it can be seen that, unlike the waveform of FIG. 7, a current component circulating in the circuit is generated in the waveform of FIG. 2.

[0060] In the waveform examples of FIGS. 7 and 2, for the sake of clarity, the main phase difference φ used for the output control of the DAB converter is 0°, that is, the condition under which no output power Po is generated (when the control method of the first embodiment is not used). However, even when the main phase difference φ is actually set to a value other than 0° at which the output power Po is generated, similarly, by changing (modulating) the first phase difference φ1 and the second phase difference φ2, a change occurs in the reactor voltage generated across the reactor. Therefore, a difference occurs in the current component (both the active component and the reactive component) circulating in the circuit.

[0061] In the DAB converter using the control method of the previous patent application when the output power Po is generated by the above mechanism, the reduction ratio of the reactive current component becomes larger than that of the active current component. In particular, when using the control method of this Example 1, since both the active and reactive current components change, by setting the first phase difference φ1 and the second phase difference φ2, it becomes possible to further reduce the reactive current component circulating in the circuit with respect to the control method of the previous patent application.

[0062] (Effect of Example 1) According to the three-phase DAB type DC / DC converter, its control device 40, and the control method of this Example 1, the output power Po can be changed by adjusting any of the parameters of the main phase difference φ, the primary side duty ratio D1, the secondary side duty ratio D2, the first phase difference φ1, and the second phase difference φ2. Therefore, while preventing an increase in hardware cost, it is possible to adjust to a more optimal circuit operation mode than the conventional DAB. As a result, by reducing power loss, the power conversion efficiency can be improved. In particular, since the three-phase DAB type DC / DC converter of this Example 1 is a bidirectional converter, even if the input / output operation is reversed, the same effect as described above can be achieved.

[0063] (Modification) The present invention is not limited to the above Example 1, and various usage forms and modifications are possible. Examples of such usage forms and modification examples include, for example, the following (a) to (c). (a) In FIGS. 2 to 4, a control method in which the pulse patterns of the primary side duty ratio D1 and the secondary side duty ratio D2 are added to both the primary side drive pulses S11 to S16 and the secondary side drive pulses S31 to S36 has been described. However, it may be changed to a control method in which the pulse pattern of the primary side duty ratio D1 is added only to the primary side drive pulses S11 to S16, or a control method in which the pulse pattern of the secondary side duty ratio D2 is added only to the secondary side drive pulses S31 to S36. Even if such a control method is adopted, the same operation as in the above Example 1 can be achieved. (b) The configuration of the power conversion unit in the DAB-type DC / DC converter shown in FIG. 1 may be changed to a configuration other than that shown in the figure. (c) Even if the primary-side inverter 10, the secondary-side inverter 30, and the three-phase transformer 20 are changed to single-phase or four or more phases instead of three phases, the control method of the present invention can be applied.

Explanation of Reference Numerals

[0064] 1 Primary-side smoothing capacitor 10 Primary-side inverter 11 - 16, 31 - 36 Switches 17 - 19 Reactors 20 Three-phase transformer 30 Secondary-side inverter 37 Secondary-side smoothing capacitor 40 Control device

Claims

1. A primary inverter in which a plurality of high-level side and low-level side switches are fully bridge-connected, and a DC primary-side voltage and a primary-side current are switched to convert them into an AC voltage and an AC current for output; A transformer having a primary winding and a secondary winding, inputting the output voltage and output current of the primary inverter to the primary winding, and outputting the induced AC voltage and AC current from the secondary winding; A secondary inverter in which a plurality of high-level side and low-level side switches are fully bridge-connected, rectifying the output voltage and output current of the secondary winding, and outputting a DC secondary-side voltage and a secondary-side current; A control device that outputs a plurality of primary-side drive pulses to turn on / off the plurality of high-level side and low-level side switches in the primary inverter respectively, and outputs a plurality of secondary-side drive pulses to turn on / off the plurality of high-level side and low-level side switches in the secondary inverter respectively, and controls the output power of the secondary inverter by changing the main phase difference between the output value of the primary inverter and the input value of the secondary inverter; Comprising: The control device: When all of the low-level side switches are in the off state, all of the high-level side switches corresponding to the low-level side switches are turned on, and when all of the high-level side switches are in the off state, all of the low-level side switches corresponding to the high-level side switches are turned on, generating a pulse pattern of a duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2), and among the first phase difference arranged at a position shifted by φ1 from the center C1 of the period (T / 2) of the primary-side drive pulse and the second phase difference arranged at a position shifted by φ2 from the center C2 of the period (T / 2) of the secondary-side drive pulse, adding the pulse pattern having the first phase difference to the plurality of primary-side drive pulses and / or adding the pulse pattern having the second phase difference to the plurality of secondary-side drive pulses; The pulse pattern of the duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2) is generated based on the command value of the main phase difference and the primary-side / secondary-side voltage ratio of the primary-side voltage and the secondary-side voltage. A switching power supply device, wherein a first phase difference and a second phase difference in the pulse pattern are set based on a transformer current flowing through the transformer, a secondary-side voltage, and a secondary-side current. **Claim 2** In the control device, an error e between a secondary-side current command value and a secondary-side current is obtained, the error e is corrected by proportional-integral control or the like, a reactor current effective value ILT is calculated from the measured transformer current, and a main phase difference φ between an output voltage vp of the primary-side inverter and an input voltage vp of the secondary-side inverter is calculated so that the corrected error e becomes zero, and a duty ratio of the pulse pattern is calculated. The switching power supply device according to claim 1, characterized in that. **Claim 3** The duty ratio of the pulse pattern is The switching power supply device according to claim 1 or 2, characterized in that it is a value in the range of 0 to 1. **Claim 4** The primary-side inverter, the secondary-side inverter, and the transformer are configured to perform single-phase, three-phase, or four-phase or more power conversion. The switching power supply device according to claim 1 or 2, characterized in that. **Claim 5** A primary-side inverter in which a plurality of high-level side and low-level side switches are full-bridge connected to switch a DC primary-side voltage and a primary-side current into an AC voltage and an AC current and output them, A transformer having a primary winding and a secondary winding, inputting an output voltage and an output current of the primary-side inverter to the primary winding, and outputting an induced AC voltage and an AC current from the secondary winding, A secondary-side inverter in which a plurality of high-level side and low-level side switches are full-bridge connected to rectify an output voltage and an output current of the secondary winding and output a DC secondary-side voltage and a secondary-side current, A control device for a switching power supply device, comprising: When outputting a plurality of primary-side drive pulses to turn on / off the plurality of high-level side and low-level side switches in the primary-side inverter respectively, and outputting a plurality of secondary-side drive pulses to turn on / off the plurality of high-level side and low-level side switches in the secondary-side inverter respectively, and controlling the output power of the secondary-side inverter by changing the main phase difference between the output value of the primary-side inverter and the input value of the secondary-side inverter, When all of the switches on the low level side are in the off state, turn on all of the switches on the high level side corresponding to the switches on the low level side. When all of the switches on the high level side are in the off state, turn on all of the switches on the low level side corresponding to the switches on the high level side. Generate a pulse pattern of duty ratio D (primary side duty ratio D1, secondary side duty ratio D2), and among the first phase difference arranged at a position shifted by φ1 from the center C1 of the period (T / 2) of the primary side drive pulse and the second phase difference arranged at a position shifted by φ2 from the center C2 of the period (T / 2) of the secondary side drive pulse, add the pulse pattern having the first phase difference to the plurality of primary side drive pulses and / or add the pulse pattern having the second phase difference to the plurality of secondary side drive pulses. The pulse pattern with the duty ratio D (primary side duty ratio D1, secondary side duty ratio D2) is generated based on the command value of the main phase difference and the primary side / secondary side voltage ratio of the primary side voltage and the secondary side voltage. Based on the transformer current flowing through the transformer, the secondary side voltage, and the secondary side current, the first phase difference and the second phase difference in the pulse pattern are set. A control device for a switching power supply device, characterized by this.

6. A primary side inverter in which a plurality of high level side and low level side switches are full bridge connected, and which switches a DC primary side voltage and a primary side current to convert them into an AC voltage and an AC current and outputs them. A transformer having a primary winding and a secondary winding, which inputs the output voltage and output current of the primary side inverter to the primary winding and outputs the induced AC voltage and AC current from the secondary winding. A secondary side inverter in which a plurality of high level side and low level side switches are full bridge connected, and which rectifies the output voltage and output current of the secondary winding and outputs a DC secondary side voltage and a secondary side current. A control method for a switching power supply device, comprising: Output a plurality of primary-side drive pulses to turn on / off the plurality of high-level side and low-level side switches in the primary-side inverter respectively, and output a plurality of secondary-side drive pulses to turn on / off the plurality of high-level side and low-level side switches in the secondary-side inverter respectively. When controlling the output power of the secondary-side inverter by changing the main phase difference between the output value of the primary-side inverter and the input value of the secondary-side inverter, When all of the low-level side switches are in the off state, turn on all of the high-level side switches corresponding to the low-level side switches, and when all of the high-level side switches are in the off state, turn on all of the low-level side switches corresponding to the high-level side switches, generate a pulse pattern of the duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2), and among the first phase difference arranged at a position shifted by φ1 from the center C1 of the period (T / 2) of the primary-side drive pulse and the second phase difference arranged at a position shifted by φ2 from the center C2 of the period (T / 2) of the secondary-side drive pulse, add the pulse pattern having the first phase difference to the plurality of primary-side drive pulses, and / or add the pulse pattern having the second phase difference to the plurality of secondary-side drive pulses, The pulse pattern of the duty ratio D (primary-side duty ratio D1, secondary-side duty ratio D2) is generated based on the command value of the main phase difference and the primary-side / secondary-side voltage ratio of the primary-side voltage and the secondary-side voltage, Based on the transformer current flowing through the transformer, the secondary-side voltage, and the secondary-side current, the first phase difference and the second phase difference in the pulse pattern are set. A control method for a switching power supply device, characterized by this.

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