Motor control device

Abstract

Improves the power factor and distortion of the power supply input current caused by suppressing and controlling LC resonance. [Solution] This motor control device includes a rectifier circuit that rectifies and outputs AC output from a three-phase AC power supply, an inverter circuit, a DC link capacitor whose capacitance is set to suppress ripple voltage, and a reactor whose capacitance is set to suppress ripple current, the DC link capacitor and reactor functioning as a filter, and is characterized in that, with the phase Θ of one line voltage of the three-phase AC power supply as a reference, the motor control device controls the magnitude of the output voltage vector to change from positive to negative or from negative to positive, based on the average value of the output voltage vector of the inverter circuit when the phase Θ is in a range that satisfies Equation 1, at a timing when the phase Θ starts at π / 3×n (n is an integer of 0 or greater). (Equation 1) n×π / 3≦Θ<(n+1)×π / 3 (n is an integer of 0 or greater).

Description

[Technical Field]

[0001] The present disclosure relates to a motor control device. [Background technology]

[0002] In a motor control device that functions as an LC filter, consisting of a rectifier circuit connected to an AC power supply, an inverter, a capacitor (C), and a reactor (L) connected between the AC power supply and the capacitor, the inverter is controlled to reduce the harmonic components of the DC voltage caused by the resonance of the LC filter. This reduces the harmonic components of the voltage across the capacitor, thereby reducing distortion in the current input to the motor control device. In a control method in which the AC component of the voltage detected by voltage detection that detects the DC link voltage between capacitors is extracted by a filter and resonance is suppressed by a compensation amount based on the ripple amount, depending on the magnitude of the motor load, the phase angle of the inverter output does not become appropriate, resulting in voltage saturation and insufficient resonance suppression. Therefore, the control method described in Patent Document 1 describes a phase advance means that outputs a leading AC component that is 90 degrees ahead of the AC component extracted by the filter, a weighting unit that outputs the sum of a value obtained by multiplying the AC component by a gain p (p is a variable that increases as the inverter output increases and varies within the range of 0≦p≦1) and a value obtained by multiplying the leading AC component by a gain (1−p), and a gain that outputs a correction signal obtained by multiplying the output of this weighting unit by a predetermined gain, thereby adjusting p according to the inverter output and achieving a resonance suppression effect with a small compensation amount regardless of the magnitude of the motor load. Patent Document 2 describes a control method for a power conversion device in which a reactor voltage (VL) is detected with the potential on the capacitor side as a reference potential, a compensation amount based on the magnitude of VL is subtracted from a voltage control rate command which is the ratio of the amplitude of the AC voltage output by the inverter circuit to the average value of the DC voltage, and a switching signal generated based on the voltage control rate command is given to the inverter circuit. The compensation amount increases as the reactor voltage increases, and the absolute value of the compensation amount decreases as the amplitude of the AC current output from the inverter circuit increases. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Patent No. 5591215 [Patent Document 2] Patent No. 5712987 Summary of the Invention [Problem to be solved by the invention]

[0004] In a motor control device that controls the power supplied by an inverter circuit and an LC filter made up of a rectifier circuit, a capacitor (C) that tolerates DC voltage pulsation, and a reactor (L), the filter extracts the DC voltage pulsation component and compensates for it in the inverter control amount. In this case, compensation control is performed up to pulsation of harmonic components that are 6 x m times the power supply frequency (m is an integer greater than or equal to 0), which can result in a decrease in the power factor of the power supply input current and significant distortion. The present disclosure aims to improve the power factor and distortion of the input current of a power supply caused by suppression control of LC resonance using a rectifier circuit, a capacitor (C) that tolerates DC voltage pulsation, an LC filter made of a reactor (L), and an inverter circuit. [Means for solving the problem]

[0005] A motor control device according to a first aspect includes: a rectifier circuit that rectifies and outputs AC output from a three-phase AC power supply; an inverter circuit having a plurality of switching elements that converts the DC output by the rectifier circuit into AC by switching operation of the plurality of switching elements and supplies the AC to a motor; a capacitor connected between input nodes of the inverter circuit, that tolerates fluctuations in voltage output by the rectifier circuit and has a capacitance value set so as to suppress ripple voltage resulting from switching operation of the inverter circuit; and a reactor between the three-phase AC power supply and the capacitor, that has a capacitance value set so as to suppress ripple current resulting from switching operation, the motor control device having a filtering function with the capacitor and the reactor, characterized in that the motor control device controls the magnitude of the output voltage vector to change from positive to negative, or from negative to positive, based on the average value of the output voltage vector of the inverter circuit when the phase Θ is in a range that satisfies Equation 1, at a timing when the phase Θ starts at π / 3×n (n is an integer equal to or greater than 0) using the phase Θ of one line voltage of the three-phase AC power supply as a reference. Equation 1 satisfies the condition n×π / 3≦Θ<(n+1)×π / 3 (n is an integer equal to or greater than 0). In this case, the power factor and distortion of the power supply input current caused by suppression control of LC resonance can be improved by using a rectifier circuit, a capacitor (C) that tolerates DC voltage pulsation, an LC filter made up of a reactor (L), and an inverter circuit. A motor control device according to a second aspect can be the motor control device according to the first aspect, characterized in that the output voltage of the inverter circuit is controlled so that the output voltage vector pulsates one or more times in synchronization with the period of LC resonance formed by the reactor and capacitor, within a range in which the phase Θ of one line voltage of a three-phase AC power supply is used as a reference, and the phase Θ satisfies Equation 2. Equation 2 satisfies n×π / 3≦Θ≦n×π / 3+π / 6 (n is an integer equal to or greater than 0). A motor control device according to a third aspect is the motor control device according to the first aspect, characterized in that when controlling the magnitude of the output voltage vector of the inverter circuit to change from positive to negative or from negative to positive, the amount of change in the output voltage vector of the inverter circuit from positive to negative or from negative to positive is controlled to be inversely proportional to the effective value of the current flowing through the motor to which AC is supplied from the inverter circuit. A motor control device according to a fourth aspect is the motor control device according to the first aspect, characterized in that the DC voltage between the capacitors contains 6×m-th order (m is an integer not less than 0) harmonic components with the frequency of the line voltage as a fundamental wave, the amplitude of the 6×m-th order (m is an integer not less than 0) harmonic is 6% or less of the direct current component of the DC voltage, and the phase of the 6×m-th order (m is an integer not less than 0) harmonic is controlled to be π±π / 16 based on the fundamental wave phase of the line voltage. A motor control device according to a fifth aspect is the motor control device according to the first aspect, characterized in that a transfer characteristic formed by the DC voltage across the capacitor and the maximum value among the absolute values ​​of the three line voltages of the three-phase AC power supply becomes a second-order lag element, and the output voltage of the inverter circuit is controlled so that the attenuation coefficient of the transfer function of the second-order lag element is greater than 0.7. A motor control device according to a sixth aspect can be the motor control device according to the first or fifth aspect, characterized in that the output voltage of the inverter circuit is controlled so that the value of the DC voltage across the capacitor is equal to or less than the maximum absolute value of the three line voltages of the three-phase AC power supply. [Brief explanation of the drawings]

[0006] [Figure 1] 1 is a diagram showing the overall configuration of a motor control device according to an embodiment of the present invention; [Figure 2] FIG. 2 is a diagram showing values ​​of line voltages output from an AC power supply. [Figure 3] FIG. 2 is a diagram for explaining an output voltage vector. [Figure 4] FIG. 2 is a control block diagram of a control unit. [Figure 5]FIG. 10 is a diagram showing timing of control. [Figure 6] 1 is an example of a model for controlling a motor control device. [Figure 7] This is the original equivalent control model of the motor control device. [Figure 8] FIG. 2 is a circuit diagram showing a simple equivalent circuit of the motor control device of FIG. [Figure 9] FIG. 9 is a diagram showing a comparative example 1 of a control block that controls the circuit shown in FIG. [Figure 10] FIG. 9 is a diagram showing a comparative example 2 of a control block that controls the circuit shown in FIG. [Figure 11] FIG. 2 is a diagram showing a control block of the present embodiment. [Figure 12] FIG. 10 is a diagram showing the frequency characteristics of a transfer function when there is no power supply impedance (hereinafter referred to as power supply Z). [Figure 13] FIG. 10 is a diagram illustrating the frequency characteristics of the transfer function when a power supply Z is present. [Figure 14] FIG. 10 is a diagram illustrating the frequency characteristics of the transfer function of a second-order delay element. [Figure 15] FIG. 10 is a diagram showing the relationship between the magnitude of the power supply Z and the magnitude of the attenuation coefficient. [Figure 16] FIG. 10 is a diagram showing a waveform of a DC voltage. [Figure 17] FIG. 10 is a diagram illustrating harmonic components in the present embodiment. DETAILED DESCRIPTION OF THE INVENTION

[0007] Hereinafter, embodiments of the present disclosure will be described in detail with reference to the accompanying drawings. FIG. 1 is a diagram showing the overall configuration of a motor control device 1 according to this embodiment. The motor control device 1 receives three-phase AC power from an AC power supply 20, converts the AC power, and supplies it to a motor 65. The motor control device 1 includes a rectifier circuit 2, an inverter circuit 3, a capacitor 4, a reactor 5, and a control unit 10.

[0008] The rectifier circuit 2 is connected to an AC power supply 20, and rectifies and outputs the AC output from the three-phase AC power supply. The rectifier circuit 2 is a three-phase full-wave rectifier circuit. The inverter circuit 3 converts the DC output from the rectifier circuit 2 into three-phase AC and supplies it to the motor 65. The three-phase AC output from the inverter circuit 3 are referred to as u-phase, v-phase, and w-phase from the top line in FIG. 1. The voltage between the u-phase and v-phase is referred to as the line voltage V uv The voltage between the v and w phases is the line voltage V vw The voltage between the w and u phases is the line voltage V wu Let's say.

[0009] The capacitor 4 is provided in the DC link section where the rectifier circuit 2 and the inverter circuit 3 are connected. The capacitance value of the capacitor 4 is set to allow fluctuations in the voltage output by the rectifier circuit 2 and to suppress ripple voltage caused by the switching operation of the inverter circuit 3. Let the capacitance of the capacitor be C and the voltage across the capacitor 4 be V. dc , the current flowing through capacitor 4 is i c In addition, the current flowing from the DC link to the inverter circuit 3 is expressed as i inv The capacitor 4 is, for example, a film capacitor.

[0010] The reactor 5 is provided between the three-phase AC power supply and the capacitor 4. The capacity of the reactor 5 is set so as to suppress ripple current caused by the switching operation of the inverter circuit 3. The capacity of the reactor 5 is set to L. In the illustrated example, the reactor 5 is provided between the rectifier circuit and the DC link unit, but the reactor 5 may also be provided between the three-phase power supply and the rectifier circuit. The reactor 5 and the capacitor 4 form an LC filter.

[0011] The AC power supply 20 is a three-phase AC power supply, and in Fig. 1, the lines are designated as r-phase, s-phase, and t-phase from the top. Here, the voltage between the r-phase and the s-phase is referred to as the line voltage V rs The voltage between the s-phase and t-phase is the line voltage V st The voltage between the t phase and the r phase is the line voltage V trLet's say. The impedances included in the AC power supply 20 itself and the line from the AC power supply 20 to the rectifier circuit 2 are shown as source impedances 21r, 21s, and 21t in FIG. 1. The value of the impedance per phase varies depending on the surrounding environment. Here, the value of the source impedance per phase is L a Let's say [H].

[0012] The motor 65 is, for example, an interior permanent magnet motor (IPM motor). The motor 65 drives a compressor provided in the refrigerant circuit.

[0013] The control unit 10 controls the inverter circuit 3 and converts the AC power supplied to the motor 65 to operate the motor 65 efficiently. The control unit 10 includes a device for processing various types of information. The control unit 10 can be configured using a microcomputer, a memory device in which software for operating the microcomputer is stored, and the like. The control unit 10 includes a phase detection unit 11 and a voltage detection unit 12 . The phase detection unit 11 detects at least one line voltage of the three-phase AC power supply, processes it in a phase detection circuit to convert it into a pulse, and transmits it to the control unit 10. In the illustrated example, the phase detection unit 11 detects the line voltage between the r phase and the s phase. The voltage detection unit 12 detects the voltage across the capacitor 4 .

[0014] FIG. 2 is a diagram showing values ​​of the line voltage output from the AC power supply 20. As shown in FIG. In Figure 2, the vertical axis shows the value of the line voltage output from the three-phase AC power supply. rs The value of the line voltage V st The value of the line voltage V tr The dashed line indicates the maximum absolute value of the three line voltages. In the example shown in Figure 2, the line voltage V rs The value of is 0V, and the line voltage V rs The timing when the line voltage V goes from negative to positive is set as 0°. rsand line voltage V st The phase difference between the line voltage and the line voltage is 240 degrees. rs and line voltage V tr The phases of these voltages are shifted by 120 degrees. As described above, capacitor 4 (see FIG. 1) shown in FIG. 1 allows fluctuations in the voltage output by rectifier circuit 2 (see FIG. 1). For this reason, capacitor 4 does not smooth the three-phase full-wave rectified voltage output from rectifier circuit 2, and a potential that is the maximum of the absolute values ​​of the three line voltages is generated in capacitor C, causing it to pulsate at a frequency six times the frequency of AC power supply 20.

[0015] Here, the output voltage vector of the inverter is defined. Figure 3 is a diagram for explaining the output voltage vector. The circuit diagram in Figure 3 shows the state in which the capacitance C of the capacitor 4 in Figure 1 is 2C and it is divided into two capacitors 4a. The voltage vector of the u phase is V un Similarly, with the potential across the capacitor 4a as the reference, the voltage vector of the v phase is V vn The voltage vector of the w-phase is V wn This voltage vector V un And, V vn And, V wn Using the above, the output voltage vector of the inverter is defined by equation (1-1). Also, the magnitude of the output voltage vector V o is defined by equation (1-2).

[0016]

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[0017] FIG. 4 is a control block diagram of the control unit 10 (see FIG. 1). The control unit 10 includes a phase estimation unit 101, a voltage averaging unit 102, a voltage command generation unit 103, a comparison unit 104, a gain correction unit 105, a multiplier 106, a manipulated variable limiter 107, and a modulation factor command generation unit 108.

[0018] [Phase estimation section] The phase estimation unit 101 estimates the phase of the AC power supply 20. The phase estimation unit 101 acquires pulses transmitted from the phase detection unit 11 and calculates the phase. The method for estimating the phase is not particularly limited, and may be, for example, the method for estimating the phase by using the voltage V applied across the capacitor 4. dc Detects the voltage V dc By filtering the voltage V dc The pulsation may be extracted and the frequency and phase may be calculated from the pulsation component.

[0019] [Voltage averaging section] The voltage averaging unit 102 calculates the voltage V dc The voltage averaging unit 102 calculates the average value of the detected voltage V dc is passed through a low-pass filter to obtain the voltage V dc Average voltage V DCAVE Furthermore, the voltage averaging unit 102 may average instantaneous values ​​in a phase range from 60n° to 60(nm)° (n and m are integers) using the phase of the line voltage from the phase estimation unit 101 as a reference.

[0020] [Voltage command generation unit] The voltage command generator 103 receives the power supply voltage estimated phase θ VDC and obtains the average voltage V DCAVE The voltage command generator 103 obtains the obtained estimated power supply voltage phase θ and the average voltage V DCAVE From this, an ideal three-phase full-wave rectification is generated, and this ideal three-phase full-wave rectification is applied to the voltage command V dc * Let's say.

[0021] [Comparison section] The comparison unit 104 compares the voltage command V dc * and the detection value V detected by the voltage detection unit 12. dc Specifically, the comparison unit 104 compares the voltage command V dc * to the detected value V dc Subtract.

[0022] [Gain correction section] The gain correction unit 105 calculates the resonance suppression gain k FB In this embodiment, i inv The resonance suppression gain k FB In this embodiment, i inv Resonance suppression is performed by controlling i inv Since it is difficult to control directly, the inverter output voltage vector is manipulated to suppress resonance. inv The magnitude of the pulsation (resonance suppression gain k FB ) will cause excessive pulsation when the output voltage vector is controlled by the motor current effective value I rms The magnitude of the pulsation of the output voltage vector (resonance suppression gain k FB Specifically, the gain corrector 105 corrects the resonance suppression gain k FB is used as the numerator, and then the motor current effective value I rms Divide using as the denominator.

[0023] [Operation amount limiter] The manipulated variable limiter 107 determines upper and lower limits of the manipulated variable so as not to interfere with the manipulated variable that controls the fundamental wave of the motor current and to prevent abnormal numerical results from being output.

[0024] [Modulation rate command generation unit] The modulation factor command generating unit 108 (subtractor) generates a modulation factor command value K that controls the fundamental wave of the motor current. s ´, the modulation rate correction value K s By subtracting ´´, the modulation rate command value K s * The magnitude of the output voltage vector V o is the modulation rate command value K s * is proportional to.

[0025] [Control Timing] FIG. 5 is a diagram showing the timing of control. The control unit 10 determines the voltage command V dc * and the voltage command V dc *and the detected value V dc The control unit 10 corrects the modulation factor based on the extracted resonance component, thereby correcting the pulsation caused by resonance. inv This presses the DC link against the ground to suppress resonance in the DC link.

[0026] In Figure 5, the vertical axis on the left side shows the magnitude of the DC voltage [V], and the vertical axis on the right side shows the magnitude of the inverter output voltage vector [V]. The horizontal axis shows time [s]. In Figure 5, the detected value of the DC voltage V dc is indicated by a solid line. The command value V dc * is indicated by a dotted circle line. The instantaneous value of the output vector is indicated by a dashed line, and the average value of the output voltage vector is indicated by a broken line. In addition, in Fig. 5, the broken line drawn vertically indicates the phase of the DC voltage command. Note that the phase of the DC voltage command is the phase estimated by the phase estimation unit 101 (see Fig. 4). In addition, in Fig. 5, the two-dot chain line drawn vertically indicates part of the timing when the output voltage vector pulsates.

[0027] The control unit 10 (see FIG. 4) controls the magnitude of the output voltage vector of the inverter to change from positive to negative or from negative to positive, based on the average value of the output voltage vector of the inverter between the phases of n×(π / 3) and (n+1)×(π / 3), with the phase of one line voltage of the three-phase AC power supply as the reference, at a timing when the phase starts at n×(π / 3) (n is an integer).

[0028] Here, the magnitude of the output voltage vector is positive relative to the average value of the inverter output voltage vector when the magnitude of the output voltage vector is greater than the average value of the output voltage vector, and the magnitude of the output voltage vector is negative relative to the average value of the inverter output voltage vector when the magnitude of the average value of the output voltage vector is smaller.

[0029] Furthermore, the control unit 10 may control the magnitude of the inverter output vector to change from positive to negative or from negative to positive at more timings in addition to the timing of (π / 3)×n from the reference phase. In this embodiment, the output magnitude of the output voltage vector changes depending on the difference between the DC voltage command and the detected DC voltage, so the output line voltage vector pulsates in synchronization with the resonance period of the LC filter. In the example shown in the figure, the output voltage vector pulsates one or more times in synchronization with the LC resonance period when the phase θ is in the range of n×(π / 3)≦θ≦n×(π / 3)+(π / 6).

[0030] [Gain correction details] Next, the gain correction unit 105 calculates the resonance suppression gain k FB is used as the numerator, and then the effective motor current value I rms This section explains the division using as the denominator. FIG. 6 is an example of a model for controlling the motor control device 1. inv ´ is the current value determined by motor control. inv '' is the value of the current controlled by the resonance suppression control. inv and i inv ´ and i inv The relationship with ´´ is expressed as equation (1-3).

[0031]

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[0032] FIG. 7 is an original equivalent control model of the motor control device 1. Figure 7 shows the output voltage V0 applied to the motor load i inv is determined. The motor load value also changes depending on the motor's operating point. V0' in Figure 7 is the voltage value determined by the motor's operation. V0'' is the voltage value manipulated by resonance suppression control. The relationship between V0, V0', and V0'' is expressed by equation (1-4).

[0033]

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[0034] Here, we consider the magnitude of the pulsation of V0, taking into account the motor load. The magnitude of the pulsation of V0 is determined by the modulation factor correction value K s Since it is determined by the resonance suppression gain correction value k FBVDC The magnitude of the pulsation of V0 can be adjusted by changing the subtractor 203 in Fig. 6, the resonance suppression gain 272, and the equation (1-3) to derive the equation (1-5).

[0035]

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[0036] Furthermore, equation (1-6) can be derived from Figure 4.

[0037]

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[0038] Furthermore, the subtractor 203 and gain 271 in FIG. 7 and equation (1-4) lead to equation (1-7).

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[0039] Here, assuming that the inverter efficiency is 1 and the motor power factor is 1, the effective motor current value I rms and inverter current i inv and inverter output voltage V0 and voltage V dc The relationship is expressed by the following equation (1-8).

[0040]

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[0041] Transforming equation (1-8), output voltage V0 and voltage V dc , modulation rate command value K s * This leads to the relation (1-9).

[0042]

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[0043] Applying equations (1-5) and (1-6) to equation (1-8), we can change both sides to V dc Dividing by this yields equation (1-11).

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[0044] Here, the second term on the left side and the second term on the right side of equation (1-11) are terms determined by the magnitude of pulsation due to resonance suppression. Also, the first term on the left side and the first term on the right side are terms determined by the operating point of the motor. If we extract the magnitude of pulsation due to resonance suppression from the second term on the left side and the second term on the right side, we get the following equation (1-12).

[0045]

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[0046] In this way, k FBVDC The correction value of I rms is inversely proportional to.

[0047] FIG. 8 is a circuit diagram showing a simple equivalent circuit of the motor control device 1 of FIG. Voltage V in is the maximum value among the absolute values ​​of the three line voltages of the three-phase AC power supply 20. L is the current flowing through the reactor 5. The voltage V s is the DC voltage output from the rectifier circuit 2. dc is the voltage across the capacitor 4. The current i c is the current flowing through capacitor 4. Current i inv is the current flowing through the inverter circuit 3.

[0048] FIG. 9 is a diagram showing a comparative example 1 of a control block that controls the circuit shown in FIG. This comparative example 1 is a control model that does not perform resonance suppression control. Comparative example 1 includes three subtractors 201, 202, and 204, and three transfer functions G1, G2, and G3. The subtractor 201 subtracts the current i L is fed back through the transfer function G1, and V in Subtract from the voltage V s Output. The subtractor 202 subtracts the voltage V dc is fed back to the voltage V s to voltage V dc Subtract the voltage V L Output. The transfer function G2 is L Input the current i L Output. The subtractor 204 subtracts the current i L From the current i inv Subtracting current i c Output. The transfer function G3 is c Enter V dc Output.

[0049] By rearranging this control block, the transfer function G NoDamp (s)=V dc / V in This gives us equation (1-13).

[0050]

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[0051] FIG. 10 is a diagram showing a comparative example 2 of a control block that controls the circuit shown in FIG. In this comparative example 2, the voltage V applied across the reactor 5 is L and resonance suppression gain k FB Multiplying by and gives the current i invThe comparative example 2 differs in that it outputs the following: The comparative example 2 includes three subtractors 201, 202, and 204, three transfer functions G1, G2, and G3, and a resonance suppression gain 272.

[0052] The subtractor 201 subtracts the current i L is fed back through the transfer function G1, and V in Subtract from the voltage V s Output. The subtractor 202 subtracts the voltage V dc is fed back to the voltage V s to voltage V dc Subtract the voltage V L Output. The transfer function G2 is L Input the current i L Output. The gain of 272 is the voltage V L and gain k FB Multiplying by and gives the current i inv Output. The subtractor 204 subtracts the current i L From the current i inv Subtracting current i c Output. The transfer function G3 is c Enter V dc Output.

[0053] The control block of this comparative example 2 is rearranged to obtain a transfer function G VL (s)=V dc / V in This gives us equation (1-14).

[0054]

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[0055] FIG. 11 is a diagram showing a control block of this embodiment. The control block of this embodiment includes four subtractors 201, 202, 203, and 204, three transfer functions G1 to G3, and a gain 273. The subtractor 201 subtracts the current iL is fed back through the transfer function G1, and V in Subtract from the voltage V s Output. The subtractor 202 subtracts the voltage V dc is fed back to the voltage V s to voltage V dc Subtract the voltage V L Output. The subtractor 203 subtracts the command value V dc * to the detected value V dc Subtract. The gain 273 is the command value V dc * to the detected value V dc The gain k is subtracted from the value FB Multiplying by the current i inv Output. The subtractor 204 subtracts the current i L From the current i inv Subtracting current i c Output. The transfer function G3 is c Enter V dc Output.

[0056] The control block of this embodiment is organized as follows: VDC (s)=V dc / V in This gives us equation (1-15).

[0057]

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[0058] Here, the frequency characteristics of the transfer function are shown using equations (1-13), (1-14), and (1-15). FIG. 12 is a diagram showing the frequency characteristics of the transfer function when there is no power supply Z. The horizontal axis represents frequency [Hz], and the vertical axis represents gain [dB] and phase [deg]. Comparative Example 2 is shown by a solid line, the present embodiment is shown by a dashed line, and Comparative Example 1 is shown by a dotted line. In the absence of power supply Z, the gain at the resonant frequency increases in Comparative Example 1, which does not perform resonance suppression. On the other hand, the gain at the resonant frequency is 0 or less in both Comparative Example 2, which performs resonance control, and this embodiment, which performs resonance control, and the increase in gain is suppressed.

[0059] FIG. 13 is a diagram showing the frequency characteristics of the transfer function when a power supply Z is present. 13, similar to FIG. 12, the horizontal axis represents frequency [Hz], and the vertical axis represents gain [dB] and phase [deg]. Comparative Example 2 is represented by a solid line, the present embodiment is represented by a dashed line, and Comparative Example 1 is represented by a dotted line. In the example shown in the figure, the frequency characteristics of the transfer function when the magnitude of the power supply Z is 1.5 [mH] are shown. When the power supply Z is present, the resonance component at the resonance frequency cannot be attenuated and a peak is generated in both Comparative Examples 1 and 2. On the other hand, the resonance control of the present embodiment attenuates the resonance component. In the resonance control of this embodiment, the gain characteristic is 0 [dB] or less, so the maximum value of the DC voltage is equal to or less than the maximum absolute value of the voltages between the power supply lines.

[0060] FIG. 14 is a diagram showing the frequency characteristics of the transfer function of the second-order lag element. In Figure 14, the horizontal axis represents frequency [Hz], and the vertical axis represents gain [dB] and phase [deg]. The solid line represents the frequency characteristics when the attenuation coefficient is 10. The dashed-dotted line represents the frequency characteristics when the attenuation coefficient is 3, and the dashed-two-dotted line represents the frequency characteristics when the attenuation coefficient is 1. The long-dashed line represents the frequency characteristics when the attenuation coefficient is 0.7, the dashed line represents the frequency characteristics when the attenuation coefficient is 0.4, and the dotted line represents the frequency characteristics when the attenuation coefficient is 0.1. As shown in Fig. 14, when the damping coefficient is 0.1 and when the damping coefficient is 0.4, the gain at the resonance frequency increases. On the other hand, when the damping coefficient is 0.7 or more, the gain is 0 or less, and the gain at the resonance frequency does not increase.

[0061] FIG. 15 is a diagram showing the relationship between the magnitude of the power supply Z and the magnitude of the attenuation coefficient ξ. 15, the horizontal axis represents the power supply Z [mH], and the vertical axis represents the magnitude of the damping coefficient ξ. The solid line represents Comparative Example 2, and the dashed line represents this embodiment. Here, the damping coefficient ξ of Comparative Example 2 VL can be expressed as equation (1-16) based on equation (1-14).

[0062]

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[0063] In addition, the damping coefficient ξ VDC can be expressed as equation (1-17) based on equation (1-15).

[0064]

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[0065] Figure 15 shows the resonant gain k FB When is uniquely determined, it is shown that the attenuation coefficient ξ changes as the power supply Z increases in both Comparative Example 2 and this embodiment. In the example shown in the figure, in this embodiment, when the power supply Z is 0, the attenuation coefficient ξ is 0.7 or greater, and as the power supply Z increases, the value of the attenuation coefficient ξ increases. The value of the power supply Z varies depending on the commercial power supply environment and is often unknown. If the attenuation coefficient ξ is designed to be 0.7 when the power supply Z is 0, the attenuation coefficient ξ will be 0.7 or greater regardless of the power supply Z, and the resonant component can be suppressed. Furthermore, if the attenuation coefficient ξ is designed to be greater than 0.7 when the power supply Z is 0, the gain will be excessive when the power supply Z is 0, resulting in unstable control.

[0066] [DC voltage waveform] FIG. 16 is a diagram showing the waveform of a DC voltage. In Fig. 16, the upper table shows the case where there is no power supply Z, and the lower table shows the case where the power supply Z is 0.5 [mH]. In the table, the horizontal axis shows time [s] and the vertical axis shows the magnitude of the DC voltage [V]. The DC voltage waveform of Comparative Example 2 is shown by a solid line, and the DC voltage waveform of this embodiment is shown by a dotted line.

[0067] As shown in the lower waveform of FIG. 16, when the power supply Z is 0.5 mH, the difference between the maximum and minimum values ​​of the DC voltage in Comparative Example 2 is larger than the difference between the maximum and minimum values ​​of the DC voltage in this embodiment. With the resonant control of this embodiment, as shown in FIG. 13, even when the power supply Z is 1.5 mH, the gain characteristic is 0 dB or less, so the maximum value of the DC voltage is less than the maximum absolute value of the voltages between the multiple power supply lines. On the other hand, in Comparative Example 2, when the power supply Z is present, as shown in FIG. 13, the gain increases at the resonant frequency, and the resonant component cannot be sufficiently attenuated. Therefore, in Comparative Example 2, the maximum value of the DC voltage is greater than the maximum absolute value of the voltages between the multiple power supply lines, as shown in the lower waveform of FIG. 16.

[0068] [Harmonic components] FIG. 17 is a diagram showing harmonic components contained in a DC voltage waveform in this embodiment. In Figure 17, the horizontal axis represents the phase [deg], and the vertical axis represents the ratio of the DC component to the harmonic component of the DC voltage. In this embodiment, since capacitor 4 (see Figure 1) does not smooth the DC voltage, the DC voltage contains 6xm-th order (m is an integer equal to or greater than 0) harmonic components with the frequency of the power supply line voltage as the fundamental wave. The DC voltage can be separated into a DC component and a harmonic component and expressed by the following equation:

[0069]

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[0070] The first term on the right side of equation (1-18) is the DC component of the DC voltage. The second term on the right side of equation (1-18) is the 6×m-th harmonic component. Furthermore, the third and subsequent terms on the right side of equation (1-18) contain carrier components and the like. In Figure 17, this DC component V dc0 and harmonic components V dc6m cos(6mω g t+θ 6m ) is shown, and the harmonic components up to m = 3 are shown.

[0071] In Fig. 17, the reference phase is indicated by a dotted line. Note that the reference phase here is the phase of one line voltage of the AC power supply 20. In Fig. 17, the 6th harmonic is indicated by a solid line, the 12th harmonic is indicated by a dashed line, and the 18th harmonic is indicated by a broken line. As shown in Figure 17, the ratio of the 6th harmonic to the DC component of the DC voltage is kept within 6%, the ratio of the 12th harmonic to the DC component of the DC voltage is kept within 2%, and the ratio of the 18th harmonic to the DC component of the DC voltage is kept within 1%.

[0072] It can be controlled within the range of equations (1-19) and (1-20) by using a command value for a DC voltage with amplitude and phase of the 6×mth harmonic component (m is an integer greater than or equal to 0) of an ideal three-phase full-wave rectified waveform.

[0073]

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[0074] [Action and effect] The motor control device 1 of the present disclosure includes a rectifier circuit 2 that rectifies and outputs AC output from a three-phase AC power supply 20, an inverter circuit 3 having a plurality of switching elements that converts the DC output from the rectifier circuit 2 into AC by switching operations of the plurality of switching elements and supplies the AC to a motor, a capacitor 4 that is connected between input nodes of the inverter circuit 3 and has a capacitance value that allows fluctuations in the voltage output by the rectifier circuit 2 and suppresses ripple voltage caused by the switching operations of the inverter circuit 3, and a switch between the three-phase AC power supply 20 and the capacitor 4. and a reactor 5 having a capacitance value set to suppress ripple current caused by rectifying operation, the capacitor 4 and the reactor 5 functioning as a filter, the motor control device controlling the magnitude of the output voltage vector from the inverter circuit 3 when the phase Θ is in a range satisfying n×π / 3≦Θ<(n+1)×π / 3 (n is an integer greater than or equal to 0) at a timing starting from π / 3×n (n is an integer greater than or equal to 0) with the phase Θ of one line voltage of the three-phase AC power supply 20 as a reference, controls so that the magnitude of the output voltage vector changes from positive to negative or from negative to positive based on the average value of the output voltage vector from the inverter circuit 3 when the phase Θ is in a range satisfying n×π / 3≦Θ<(n+1)×π / 3 (n is an integer greater than or equal to 0). The motor control device improves distortion of the power supply input current caused by suppression control of LC resonance by the LC filter formed by the rectifier circuit 2, the capacitor 4 which allows DC voltage pulsation, and the reactor 5, and the inverter circuit 3. Here, the motor control device 1 controls the output voltage of the inverter circuit 3 so that the output voltage vector pulsates one or more times in synchronization with the period of the LC resonance formed by the reactor 5 and the capacitor 4, within a range where the phase Θ of one line voltage of the three-phase AC power supply 20 satisfies n×π / 3≦Θ≦n×π / 3+π / 6 (n is an integer equal to or greater than 0), using the phase Θ as a reference. When the motor control device 1 controls the magnitude of the output voltage vector of the inverter circuit 3 to change from positive to negative or from negative to positive, it controls the magnitude of the change in the output voltage vector of the inverter circuit 3 from positive to negative or from negative to positive so that it is inversely proportional to the effective value of the current flowing through the motor to which AC is supplied from the inverter circuit 3. In this case, deterioration of the control stability or resonance suppression performance is prevented. Here, the motor control device 1 controls the output voltage of the inverter circuit 3 so that the DC voltage across the capacitor 4 contains 6×m-th order (m is an integer greater than or equal to 0) harmonic components with the line voltage frequency as the fundamental wave, the amplitude of the 6×m-th order (m is an integer greater than or equal to 0) harmonic being 6% or less of the direct current component of the DC voltage, and the phase of the 6×m-th order (m is an integer greater than or equal to 0) harmonic being π±π / 16 based on the fundamental wave phase of the line voltage. In this case, the power factor of the input current on the power supply side is improved and power supply harmonics are suppressed. The motor control device 1 is characterized in that the transfer characteristic formed by the DC voltage across the capacitor 4 and the maximum value among the absolute values ​​of the three line voltages of the three-phase AC power supply 20 becomes a second-order lag element, and the output voltage of the inverter circuit 3 is controlled so that the attenuation coefficient of the transfer function of the second-order lag element is greater than 0.7. In this case, the deterioration of the resonance suppression performance is suppressed. The motor control device 1 controls the output voltage of the inverter circuit 3 so that the value of the DC voltage across the capacitor 4 is equal to or less than the maximum value of the absolute values ​​of the three line voltages of the three-phase AC power supply 20. In this case, the deterioration of the resonance suppression performance is suppressed.

[0075] Although the embodiments have been described above, the technical scope of the present disclosure is not limited to the scope of the above-described embodiments. It is clear from the claims that combinations of two or more of the above-described embodiments, and various modifications or improvements to the above-described embodiments, are also included in the technical scope of the present disclosure. [Explanation of symbols]

[0076] 1...motor control device, 2...rectifier circuit, 3...inverter circuit, 4...capacitor, 5...reactor, 10...control unit, 11...phase detection unit, 12...voltage detection unit, 20...AC power supply, 21r, 21s, 21t...power supply impedance, 65...motor, 101...phase estimation unit, 102...voltage averaging unit, 103...voltage command generation unit, 104...comparison unit, 105...gain correction unit, 106...multiplier, 107...operation amount limiter, 108...modulation factor command generation unit

Claims

1. a rectifier circuit that rectifies and outputs AC output from a three-phase AC power supply; an inverter circuit having a plurality of switching elements, which converts the direct current output by the rectifier circuit into alternating current by switching operations of the plurality of switching elements and supplies the alternating current to a motor; a DC link capacitor connected between input nodes of the inverter circuit, which allows fluctuations in the voltage output by the rectifier circuit and has a capacitance set to suppress ripple voltage caused by the switching operation of the inverter circuit; a reactor having a capacitance value set so as to suppress ripple current caused by switching operations, between the three-phase AC power supply and the DC link capacitor; A motor control device having a filter function with the DC link capacitor and the reactor, A motor control device that controls the magnitude of an output voltage vector to change from positive to negative, or from negative to positive, with the average value of the output voltage vector of the inverter circuit when the phase Θ is in a range that satisfies Equation 1, at a timing when the phase Θ starts at π / 3×n (n is an integer greater than or equal to 0) using the phase Θ of one line voltage of a three-phase AC power supply as a reference. (Formula 1) n×π / 3≦Θ<(n+1)×π / 3 (n is an integer equal to or greater than 0)

2. 2. The motor control device according to claim 1, wherein the output voltage of the inverter circuit is controlled so that the output voltage vector pulsates one or more times in synchronization with a period of LC resonance formed by the reactor and the DC link capacitor within a range in which the phase Θ of one of the line voltages of the three-phase AC power supply satisfies Equation 2, using the phase Θ as a reference. (Formula 2) n×π / 3≦Θ≦n×π / 3+π / 6 (n is an integer equal to or greater than 0)

3. When controlling the magnitude of the output voltage vector of the inverter circuit so as to change from positive to negative or from negative to positive, The magnitude of the change in the output voltage vector of the inverter circuit from positive to negative or from negative to positive is inversely proportional to the effective value of the current flowing through the motor to which AC is supplied from the inverter circuit, 2. The motor control device according to claim 1, wherein the motor control device controls the motor.

4. The DC voltage between the DC link capacitors contains 6×m-th order (m is an integer of 0 or more) harmonic components with the frequency of the line voltage as a fundamental wave, the amplitude of the 6×m-th order (m is an integer of 0 or more) harmonic being 6% or less of the DC component of the DC voltage, and the phase of the 6×m-th order (m is an integer of 0 or more) harmonic being π±π / 16 with respect to the fundamental wave phase of the line voltage.

2. The motor control device according to claim 1, wherein the output voltage of the inverter circuit is controlled.

5. A transfer characteristic consisting of the DC voltage between the DC link capacitor and the maximum value among the absolute values ​​of the three line voltages of the three-phase AC power supply becomes a second-order lag element, and an attenuation coefficient of the transfer function of the second-order lag element is set to be greater than 0.

7.

2. The motor control device according to claim 1, wherein the output voltage of the inverter circuit is controlled.

6. The value of the DC voltage across the DC link capacitor is equal to or less than the maximum value of the absolute values ​​of the three line voltages of the three-phase AC power supply.

6. The motor control device according to claim 1, wherein the output voltage of the inverter circuit is controlled.

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