Method for processing a signal received by a receiver, the received signal being derived from two continuous phase modulation signals

Abstract

The invention relates to a method for processing a received signal derived from two continuous phase modulation signals, the method comprising: frequency-shifting (102) the square of the received signal so as to produce a first shifted signal and a second shifted signal having a shifted spectrum of D / 2 and of -D / 2, respectively, relative to a spectrum of the square of the signal, where D is a bit rate of each of the two continuous phase modulation signals: determining (108) a first frequency and a second frequency at which a power spectral density of the first shifted signal and of the second shifted signal takes a first maximum value and a second maximum value, respectively, and determining a third frequency at which a power spectral density of the square of the signal takes a third maximum value; performing a central tendency measure of the frequencies in order to estimate (110) a frequency difference between the received signal and each of the two continuous phase modulation signals.

Description

[0001]DESCRIPTION TITLE: Method for processing a signal received by a receiver, the received signal being derived from two continuous phase-modulated signals. FIELD OF THE INVENTION This disclosure relates to the field of MISO (Multiple Input Single Output) communications, in which several transmitters send signals to a single receiver. This disclosure relates more particularly to a method for processing a signal received by a receiver, the received signal being derived from two continuous phase-modulated signals. PRIOR TECHNOLOGY The document “Space Time Coding for Aeronautical Telemetry: Part 2 Decoder and system performance Rice”, Palmer, Lain, Nelson, IEEE Transactions, describes a method for processing a signal transmitted to a receiver using MISO technology. This method includes detecting a pilot signal in the received signal, then estimating a frequency difference between the received signal and signals transmitted by two transmitters.and from which the received signal originates. This process also includes an estimation of characteristic parameters of the transmission channel between the transmitters and the receiver. Knowledge of this frequency difference allows the received signal to be corrected before it is demodulated using the estimated channel parameters. The estimation of the frequency difference between the received signal and the transmitted signals described in this document is carried out in two phases: first, a rough frequency search is performed using a Fast Fourier Transform (FFT), then a more precise estimation of the difference is performed using an algorithm called "Zoom FFT". Next, the Golden Section Search (GSS) algorithm is used to produce a final estimate, evaluating frequencies close to the one estimated from the rough search. At the end of this first estimation,An initial estimation of the channel parameters is performed. In a second iteration, these updated parameters are fed back into the frequency estimation to improve its accuracy. This process continues iteratively, for example, until a stopping criterion is reached. One drawback of this method is that it requires the incorporation of several pilot signals into the received signal. However,These pilot signals cause harmonics to appear in the spectrum of the received signal, inducing a very high probability of false alarms when the Doppler acquisition range exceeds 3 kHz at a bit rate of 20 Mbps (which is the case, in particular, in aeronautical applications). A second drawback of this method is that it includes an iterative loop comprising the module that performs the frequency deviation estimation and the module that estimates the channel parameters. The module that performs the frequency deviation estimation requires estimates of the transmission channel parameters. Because of this iterative loop,The demodulator loses more data to "lock onto" the received signal (on the order of 80 kbits). A third drawback of this method is that it does not allow reaching the lowest demodulation threshold of the demodulator described in application WO 2020 / 148511 A1. DESCRIPTION OF THE INVENTION One problem to be solved is that of estimating the frequency difference between a received signal and transmitted signals from which the received signal originates, without needing parameters of the transmission channel used or pilot signals. This problem is solved by a method for processing a signal received by a receiver, the received signal being derived from two continuous phase-modulated signals previously transmitted by two respective transmitters, the processing method comprising the following steps implemented by a signal processing device: squaring the received signal so as to produce a squared signal,^ frequency shift of the squared signal so as to produce:^ a first shifted signal having a spectrum shifted by a first frequency step equal to D / 2 with respect to a spectrum of the squared signal, where D is a bit rate of each of the two continuous phase-modulated signals, ^ a second shifted signal having a spectrum shifted by a second frequency step equal to -D / 2 with respect to the spectrum of the squared signal;^ determination of frequencies comprising:^ a first frequency at which a power spectral density of the first shifted signal takes a first maximum value, ^ a second frequency at which a power spectral density of the second shifted signal takes a second maximum value,^a third frequency at which a power spectral density of the square wave signal takes a third maximum value;^ estimation of a frequency difference between the received signal and each of the two continuous phase-modulated signals by determining a measure of the center tendency of the frequencies;^ frequency correction of the received signal using the frequency difference, so as to produce a corrected signal; and^ demodulation of the corrected signal. The inventors were able to determine that in the absence of a frequency difference, the spectrum of the square wave signal would exhibit peaks at specific frequencies, including the frequencies 0, D / 2 and -D / 2, and that in the presence of a frequency difference ^, ^ between the received signal and the previously emitted signals, such peaks are found at shifted frequencies, notably 2^ ^,2^^ + ^ / 2, and 2^^ − ^ / 2. The squaring and frequency shifting steps introduce a non-linearity that allows for the production of shifted signals whose power spectral densities have pure components where the modulation of the emitted signals has been removed. It should be noted that squaring and frequency shifting lead to a signal-to-noise ratio much lower than that obtained with squaring. A signal is the medium that allows information to be transmitted over a distance, from its source to its destination. This signal could, for example, be a phase-modulated radio frequency signal using telemetry information. The process may also include the following optional features, taken alone or in combination whenever possible: - the central tendency measurement is an average; - the two continuously phase-modulated signals have a modulation scheme SOQPSK ;- It comprises the following steps:^ calculation of a power spectral density ratio between the third value and a fourth power spectral density value depending on at least one of the first and second values;^ from the power spectral density ratio, estimation of the absolute value of the difference between the respective propagation delays of two continuous phase-modulated signals between the two transmitters and the receiver;- the fourth value is a measure of the central tendency of the first and second values;- the estimation of the absolute value of the difference between the propagation delays comprises the following substeps:^ in a plurality of predefined ranges respectively associated with candidate values ​​for the absolute value of the difference between the propagation delays, selection of a range including the power spectral density ratio,^selection of the candidate value associated with the selected range as the absolute value of the difference between the propagation delays; - it further includes steps of:^ generation:^ of a first score indicating a similarity between a pilot signal received by the receiver and a first pilot signal time-shifted according to the time difference in a first direction relative to a reference pilot signal, and ^of a second score indicating a similarity between the pilot signal received by the receiver and a second pilot signal time-shifted according to the time difference in a second direction relative to the reference pilot signal, the second direction being opposite to the first direction,^Determination of the sign of the difference between the propagation delays from the first score and the second score; - the sign of the difference is determined by a pilot signal detector having detected the received pilot signal; - it further includes:^ the generation of:^ a first score indicating a similarity between a received pilot signal detected in the received signal and a first pilot signal time-shifted according to the time difference in a first direction relative to a reference pilot signal, ^a second score indicating a similarity between the pilot signal detected in the received signal and a second pilot signal time-shifted according to the time difference in a second direction relative to the reference pilot signal, the second direction being opposite to the first direction, and ^a third score indicating a similarity between the pilot signal detected in the received signal and the reference pilot signal,^zeroing the absolute value of the difference provided that the first score, the second score, and the third score indicate that there is a greater similarity between the detected pilot signal in the received signal and the reference pilot signal than between the detected signal in the received signal and either of the first time-shifted pilot signal or the second time-shifted pilot signal. The problem is also solved by a signal processing device comprising:^ an estimator including:^ an input for obtaining a signal received by a receiver and resulting from two previously transmitted continuous phase-modulated signals from two respective transmitters, ^ a squaring module configured to square the received signal so as to produce a squared signal,^a frequency-shifting module configured to produce:^ a first shifted signal having a spectrum shifted by a first frequency step equal to D / 2 with respect to a spectrum of the squared signal, where D is a bit rate of each of the two continuous phase-modulated signals, ^a second shifted signal having a spectrum shifted by a second frequency step equal to -D / 2 with respect to the spectrum of the squared signal, ^a spectral analysis module configured to determine frequencies comprising: ^a first frequency at which a power spectral density of the first shifted signal takes a first maximum value, ^a second frequency at which a power spectral density of the second shifted signal takes a second maximum value, ^a third frequency at which a power spectral density of the squared signal takes a third maximum value,^a frequency estimation module configured to estimate a frequency deviation between the received signal and each of two phase-modulated signals by determining a measure of the center tendency of the frequencies;^ a frequency corrector configured to correct a frequency of the received signal using the frequency deviation, so as to produce a corrected signal; and^ a demodulator configured to demodulate the corrected signal. The device may also include the following optional features, taken alone or in combination whenever possible: - the estimator further includes:^ a time estimation module configured to:^ calculate a power spectral density ratio between the third value and a fourth power spectral density value depending on at least one of the first and second values,^ from the power spectral density ratio,to estimate the absolute value of a difference between the respective propagation delays of the two continuous phase-modulated signals between the two transmitters and the receiver; - it further comprises:^ a pilot detector configured to detect a pilot signal in the received signal, ^ a channel estimator configured to estimate, from the pilot signal, the parameters of a channel used for the transmission of the two continuous phase-modulated signals, the demodulator being configured to demodulate the corrected signal taking into account the parameters; - it further comprises:^ a second frequency corrector configured to correct a frequency of the pilot signal using the frequency difference, so as to produce a corrected pilot signal,The channel estimation module is configured to estimate the parameters from the corrected pilot signal. The channel estimation module is configured to determine the sign of the difference between propagation delays from the pilot signal. DESCRIPTION OF FIGURES Other features, purposes, and advantages of the invention will become apparent from the following description, which is purely illustrative and not limiting.and which must be read in conjunction with the accompanying drawings, in which: Figure 1 schematically illustrates a receiver according to a first embodiment. Figure 2 is a functional block representation of a signal processing device according to a first embodiment. Figure 3 is a flowchart of steps for a process implemented by a frequency deviation estimator according to one embodiment. Figure 4 is a functional block representation of a signal processing device according to a second embodiment. Figure 5 is a flowchart of steps for a process for estimating the absolute value of a difference between two propagation delays, according to one embodiment. Figure 6 is a flowchart of steps for determining the aforementioned difference from its absolute value. Throughout the figures, similar elements bear identical reference numerals. DETAILED DESCRIPTION OF THE INVENTION With reference to Figure 1,A MISO (Multiple Input Single Output) radio communication network comprises two transmitters and one receiver. 1. The two transmitters are configured to transmit two signals in parallel ^, ^ , ^ ^The two emitted signals have the same bit rate, denoted D, and are respective continuous phase modulation (CPM) signals. In the embodiment shown in Figure 1, the two transmitters are mounted on the same aircraft. Preferably, the emitted signals use the SOQPSK (Spectrally Efficient Offset Quadrature Phase Shift Keying) modulation scheme. This modulation scheme can be supplemented by space-time coding (STC). The channel used for transmitting the two aforementioned signals to the receiver can be characterized, in particular, by the following parameters: Δ, Δ: channel gains seen by the signals respectively; Δ, Δ, Δ, Δ: propagation delay between the transmitting and receiving antennas. For example, the signals could be telemetry signals during flight tests or for launch vehicles (Ariane, etc.).The receiver 1 comprises an antenna 2, a sampler 4 and a processing device 4. The antenna 2 is suitable for receiving a signal ^ resulting from the two signals ^^, ^^ respectively emitted by the two transmitters in parallel according to the MISO principle. The received signal ^ is of the following form:. where ^ is Gaussian white noise, and ^^ is a "common" baseband frequency gap. Sampler 4 is configured to sample the signal into a sampled signal, denoted ^^, which is supplied to the processing device 6. The sampled signal s’écrit : = ^ℎ ^ ^ + ℎ ^ ^^ ^^^^^^^ ^,^^^ ^ ^,^^^ + ^^^ ^^^^ , ^^^^ ^^^^ ^^ ^^é^^^^^^ ^^é^ℎ^^^^^^^^^^, ^^ = Referring to Figure 2, the processing device 6 comprises an input buffer, an estimator 10, a frequency corrector 12, and a demodulator 14. The input buffer 10 is intended to store the received signal ^^ (after sampling). The estimator 10's function is to estimate the frequency difference ^^ between the received signal r and each of the two transmitted signals ^^, ^^. In this text, estimated data is denoted with a hat symbol; the estimate of the frequency difference determined by the estimator is therefore denoted ^^^. The operation of the estimator will be detailed later. The frequency corrector 12 is arranged at the output of the estimator 10. Its function is to eliminate the frequency difference that appeared during the transmission of the signals ^ ^ , ^ ^Based on the frequency deviation provided by the estimator, the frequency corrector 10 is known from the prior art. The demodulator 14 is arranged at the output of the frequency corrector 12. Its function is to demodulate a signal corrected by the frequency corrector. Its operation is known from the prior art. The processing device 6 also includes a pilot detector 16, a pilot buffer 18, a second frequency corrector 20, and a channel estimator 22. The pilot detector 16's function is to detect, in the received signal stored in the input buffer, a pilot signal, which is then stored in the pilot buffer 18. The pilot signal is a predefined bit sequence. The pilot signal is used for synchronizing the demodulator. The pilot detector is known from the prior art.The second frequency corrector 20 is configured to apply a correction to a pilot signal stored in the pilot buffer, based on a frequency deviation provided by the estimator. Its operation is similar to that of the corrector 12, except that the signal it corrects is the pilot signal. ^Channel estimator 22 is arranged at the output of the second corrector 20. The function of channel estimator 22 is to determine the transmission channel parameters of the two continuous phase-modulated signals (from the two transmitters to the receiver). A particular feature of estimator 10 is that it is blind, meaning it operates without requiring any pilot signal as input. Furthermore, it does not take as input the channel parameters estimated by channel estimator 22. Estimator 10 only takes as input the signal received in the input buffer 8. The modules 10, 12, 14, 16, 20, and 22 of the processing device 6 shown in Figure 2 can be distinct physical components, each specialized for its task.Alternatively, the processing unit 6 includes a processor and memory storing a computer program to be executed by the processor, and modules 10, 12, 14, 16, 20, and 22 are software modules, i.e., different parts of this program. The processor can be of any type; in particular, it may have one or more cores (to implement parallel processing). The processor may be an FPGA, an ASIC, or another type of electronic circuit. In one embodiment, the estimator 10 takes the form of an electronic board. The processing unit 4 may then include, in addition to this electronic board, a workstation, an electronic acquisition board, an FEC board (which may be omitted), and an optional time-in-motion (TIM) remote board. The processing unit 4 may include an RF IN input to receive the signal received by the antenna, which is then transmitted to the TIM board.This signal is a baseband RF signal; it is digitized by an analog-to-digital converter, which may be part of the acquisition board. The digital signal output from this converter is transmitted to the electronic board that constitutes or includes the estimator 10. It is then demodulated and transmitted to the FEC board (if present). The FEC board includes a channel decoding function to correct transmission errors induced by the transmission channel. A process for processing the received signal ^. ^ implemented by the processing device 6 includes the following steps. The estimator 10 estimates a frequency difference ^^^ between the received signal r and each of the two emitted signals ^ ^ , ^ ^ using the steps shown in Figure 3. In step 100, an estimator elevation module squares the received signal ^ ^, so as to produce a squared signal. In step 102, a frequency-shifting module of the estimator takes as input the squared signal and the bit rate D of each of the two continuous phase-modulated signals, to produce two shifted signals: a first shifted signal having a spectrum shifted by a first frequency step equal to D / 2 with respect to a spectrum of the squared signal, where D is a bit rate of each of the two continuous phase-modulated signals, and a second shifted signal having a spectrum shifted by a second frequency step equal to -D / 2 with respect to the spectrum of the squared signal. Formally, the first shifted signal can be seen as the result of multiplying the squared signal by the complex term ^ ^^^^^.^ / ^Similarly, the second shifted signal can be seen as the result of multiplying the squared signal by the complex term ^^^^^.^ / ^. In the flowchart of Figure 3, this represents the multiplication of the squared signal by the term ^ ^^^^^.^, but this is only for symmetry purposes since in reality this term is equal to 1. It is therefore not truly a calculation performed by the estimator 10. In a step 104, a spectral analysis module of the estimator 10 uses the squared signal, the first shifted signal, and the second shifted signal, to determine the following frequencies: a first frequency f1 at which a power spectral density of the first shifted signal takes a first maximum value, a second frequency f2 at which a power spectral density of the second shifted signal takes a second maximum value, a third frequency f3 at which a power spectral density of the squared signal takes a third maximum value. Each of these frequency steps can typically be determined in three steps 104, 106, 108 by the aforementioned spectral analysis module.A discrete Fourier transform over N points is applied to the first shifted signal (step 104), producing a first spectrum; then the power spectral density of the first shifted signal is calculated by squaring the magnitude of the first spectrum (step 106), and the first maximum value and the first frequency f1 associated with this first maximum value are then sought within this power density (step 108). Similarly, a discrete Fourier transform over N points is applied to the second shifted signal (step 104), producing a second spectrum; then the power spectral density of the second shifted signal is calculated by squaring the magnitude of the second spectrum (step 106), and the second maximum value and the second frequency f2 associated with this first maximum value are then sought within this power density (step 108).Similarly, a discrete Fourier transform on N points is applied to the squared signal, producing a third spectrum (step 104); then the power spectral density of the squared signal is calculated by calculating the square of the magnitude of the third spectrum (step 106), and the third maximum value and the third frequency f3 associated with this third maximum value are then sought within this power density (step 108). The discrete Fourier transform used in step 104 can be modified. According to one modification, the discrete Fourier transform is governed by the following formula, involving a configurable frequency range and a configurable frequency step: This second variant is called "Modified DFT" in the literature, abbreviated as M-DFT. According to a second variant, the discrete Fourier transform is governed by the following formula, which is classical: In step 110, a frequency estimation module of the estimator estimates the frequency difference ^^^ between the received signal and each of two phase-modulated signals by determining a central tendency measure of the three frequencies determined by the spectral analysis module. This central tendency measure is, for example, a mean. In the embodiment shown in Figure 3, it is an arithmetic mean, represented by the summation operator Σ and the multiplication operator by the factor 1 / 3. Returning to Figure 2, the frequency difference ^^^ is provided to the frequency corrector 12. The frequency corrector 12 corrects the received signal ^ ^ in a corrected signal ^ ^′, using the frequency deviation ^^^ determined by estimator 10. This correction aims to eliminate the term ^^^^^^^ which appears in the signal formula ^(^) provided previously. Furthermore, the pilot detector 16 detects in the received signal ^^ a pilot signal ^^ of predefined shape. The pilot signal ^ ^ is stored in the pilot buffer. The frequency corrector 20 corrects the pilot signal ^ ^ in a corrected pilot signal ^ ^ ′, using the frequency gap ^^^ determined by estimator 10. Channel estimator 22 is based on the corrected pilot signal ^^′ to estimate the channel parameters h^, h^, ^^, ^^. The estimates produced by this channel estimator 22 are thus denoted h^^, h^^, ^^̂, ^^̂. Channel estimator 22 uses a method known from the prior art. Demodulator 14 demodulates the corrected signal ^^′ taking into account the estimates h^^, h^^, ^^ ^ ̂, ^ ^- channel parameters. The demodulator 14 also takes into account a frame start identified by the pilot detector in the received signal. The demodulator uses a method known from the prior art. Figure 4 shows a signal processing device 6' according to a second embodiment, which can replace the signal processing device 6 in the receiver 1 according to the first embodiment. The processing device 6' comprises the frequency corrector 12, the demodulator 14, as well as the pilot detector 16 and the buffers 8, 18 as described previously. The processing device further includes an estimator 11. The estimator 11 includes, on the one hand, the frequency deviation estimator 10 of the first embodiment, and has the other function of estimating the absolute value of a difference between the respective propagation delays of the two continuous phase-modulated signals between the two transmitters and the receiver.Let ^Δ^^^ = |^^̂ − ^^̂| be the estimate produced by this time estimator. We will see later that some of the steps implemented by estimator 10 are used within estimator 11 to produce this time difference. The second frequency compensator 20 of the first embodiment is absent from the second embodiment, and the channel estimator 22 is replaced by a channel estimator 23 taking as input the pilot signal ^^ and the data A process implemented by the processing device according to the second embodiment comprises the following steps. Estimator 11 estimates the frequency gap ^^^ as in the first embodiment, i.e. by carrying out the steps shown in Figure 3. Furthermore, estimator 11 carries out the steps shown in Figure 5 to estimate the absolute value ^Δ^^^ of the difference between the propagation delays ^^̂, ^^̂.In the following, we adopt the following notations: ^ ^^ : first value discussed previously, i.e., the maximum value of the power spectral density of the first shifted signal, associated with the first frequency determined in step 108; ^ ^^ : second value discussed previously, i.e., the maximum value of the power spectral density of the second shifted signal, associated with the second frequency determined in step 108; ^ ^^ : third value discussed previously, i.e., the maximum value of the power spectral density of the squared signal, associated with the third frequency determined in step 108. In a step 200, the estimator 11 calculates a power spectral density ratio between the third value ^^ and a fourth power spectral density value ^^ depending on at least one of the first value ^^ and the second value ^^. Preferably, the fourth value ^. ^is a measure of central tendency of the first value ^^ and the second value ^^, for example, an average of the first and second values. In the embodiment of Figure 5, the fourth value ^^ is an arithmetic mean of the first and second values. In a step 204, the estimator 11 uses the power spectral density ratio calculated in step 202 to estimate ^Δ^^^ = |^^̂ − ^^̂|, in other words, the absolute value of the difference between the respective propagation delays of two continuous phase-modulated signals between the two transmitters and the receiver. In one embodiment of step 204, the estimator 11 can use a plurality of predefined ranges respectively associated with candidate values ​​for the absolute value ^Δ ^These candidate values ​​are stored in memory accessible by the estimator. The estimator selects the predefined range that includes the power spectral density ratio. Then, the estimator elects as the absolute value the candidate value associated with the selected range. Preferably, the higher the lower bound of a range, the higher the associated candidate value. Furthermore, preferably, the higher the lower bound of a range, the wider the range. Such a configuration is particularly suitable when the signals ^ , ^ ^The initially issued values ​​are of the form SOQPSK. In one example embodiment, the plurality of predefined ranges comprises four ranges, in this order: a first range whose lower bound is 0, associated with the candidate value Δ = 0; a second range associated with the candidate value; a third range associated with the candidate value; and a fourth range associated with the candidate value. T is the inverse of the bit rate D: ^ = 1 / ^ In another embodiment of step 204, the estimator 11 uses a neural network trained to estimate the absolute value from the power spectral density ratio. At this stage, the signal processing device 6' knows the absolute value ^Δ^^^, but does not yet know the sign of Δ^^, and therefore does not know the value Δ^^. In other words, the signal processing device 6' does not know which of the two propagation delays ^^ and ^ ^is the highest. The signal processing device 6' implements the steps shown in Figure 6 to determine this sign. These steps are implemented, for example, by the channel estimator 23. The processing device 6' is aware of a reference pilot signal assumed to correspond to the pilot signal ^ ^detected in an ideal situation. In preliminary steps, the following were generated: a first pilot signal, time-shifted by the time difference in one direction relative to the reference pilot signal; a second pilot signal, time-shifted by the time difference in a second direction relative to the reference pilot signal, the second direction being opposite to the first. Thus, the first and second pilot signals are shifted by +Δ and −Δ, respectively. These two shifted pilot signals may have been stored in a memory accessible by the processing device 6'. The signal processing device generates a first score indicating a similarity between the pilot signal and the first pilot signal, time-shifted by +Δ (step 300).Furthermore, the signal processing device generates a second score indicating a similarity between the pilot signal and the second pilot signal, which is time-shifted by −^Δ^^^ (step 302). The first and second similarity scores are typically obtained by correlating the pilot signal with the first time-shifted pilot signal and with the second time-shifted pilot signal. The signal processing device then compares the two similarity scores produced (step 306). If both scores indicate a greater similarity between the pilot signal and the first pilot signal, which is time-shifted by +^Δ^^^, then the processing device considers the sign to be positive.Conversely, if two scores indicate a greater similarity between the pilot signal and the second pilot signal time-shifted by −Δ, then the processing device considers the sign to be negative. Optionally, the signal processing device generates a third score indicating a similarity between the pilot signal detected in the received signal and the reference pilot signal (i.e., without the shift), in step 304. If the first, second, and third scores indicate a greater similarity between the pilot signal detected in the received signal and the reference pilot signal than between the signal detected in the received signal and either of the first time-shifted pilot signal or the second time-shifted pilot signal, then the processing device can set the absolute value of the difference Δ previously provided by the estimator to zero.This reset corrects a situation where the estimator has overestimated this absolute value, when in fact it should be neglected in subsequent processing. It was previously indicated that steps 300, 302, 304, and 306 can be implemented by estimator 23. In another embodiment, all or part of these steps can be implemented by the pilot detector 16, so that the sign of Δ^^ is determined by the pilot detector 16. Furthermore, the pilot detector 16 can not only determine the sign of Δ^^ but also estimate its absolute value based on additional information indicating the position of the sample that maximizes the score for each of the two pilot sequences.For example, if the similarity score between the received signal and the first time-shifted pilot signal is maximal at sample n, and the dissimilarity score between the received signal and the second time-shifted pilot signal is maximal at sample n+2, we can deduce that one of the two channels is 2 samples behind the other channel. Channel estimator 23 calculates the estimates h^^, h^^, Δ^^ of the corresponding channel parameters, from the pilot signal ^. ^ and the difference Δ ^ ^, which is deduced from the absolute value and sign previously determined by the estimator. The operation of the frequency corrector 12 and the demodulator 14 is unchanged compared to the first embodiment.

Claims

CLAIMS 1. A method for processing a signal received by a receiver, the received signal being derived from two continuous phase-modulated signals previously emitted by two respective transmitters, the processing method comprising the following steps implemented by a signal processing device:^ squaring (100) the received signal so as to produce a squared signal,^ frequency shifting (102) of the squared signal so as to produce:^ a first shifted signal having a spectrum shifted by a first frequency step equal to D / 2 with respect to a spectrum of the squared signal, where D is a bit rate of each of the two continuous phase-modulated signals,^ a second offset signal having a spectrum shifted by a second frequency step equal to -D / 2 with respect to the spectrum of the squared signal;^ determination (108) of frequencies comprising:^ a first frequency at which a power spectral density of the first offset signal takes a first maximum value,^ a second frequency at which a power spectral density of the second offset signal takes a second maximum value,^ a third frequency at which a power spectral density of the squared signal takes a third maximum value;^ estimation (110) of a frequency difference between the received signal and each of the two continuous phase-modulated signals by determining a measure of the central tendency of the frequencies;^ frequency correction of the received signal using the frequency difference, so as to produce a corrected signal; and^ demodulation of the corrected signal.

2. Method according to claim 1,in which the central tendency measure is an average.

3. A method according to any one of the preceding claims, wherein the two continuous phase-modulated signals have a SOQPSK modulation scheme.

4. A method according to any one of the preceding claims comprising the following steps:^ calculation (202) of a power spectral density ratio between the third value and a fourth power spectral density value depending on at least one of the first and second values,^ from the power spectral density ratio, estimation (204) of the absolute value of a difference between the respective propagation delays of the two continuous phase-modulated signals between the two transmitters and the receiver.

5. A method according to the preceding claim, wherein the fourth value is a central tendency measure of the first and second values.

6. A method according to any one of claims 4 and 5,wherein the estimation (204) of the absolute value of the difference between the propagation delays comprises the following substeps: ^in a plurality of predefined ranges respectively associated with candidate values ​​for the absolute value of the difference between the propagation delays, selection of a range including the power spectral density ratio,^ selection of the candidate value associated with the selected range as the absolute value of the difference between the propagation delays.

7. A method according to any one of claims 4 to 6, further comprising steps, de :^ generation (300):^ of a first score indicating a similarity between a pilot signal received by the receiver and a first pilot signal time-shifted by the time difference in a first direction relative to a reference pilot signal, and^ of a second score indicating a similarity between the pilot signal received by the receiver and a second pilot signal time-shifted by the time difference in a second direction relative to the reference pilot signal, the second direction being opposite to the first direction, ^ determination (302) of a sign of the difference between the propagation delays from the first score and the second score.

8. Method according to the preceding claim, wherein the sign of the difference is determined by a pilot signal detector having detected the received pilot signal.

9. Method according to any one of claims 4 to 8,including further:^ generation (300):^ of a first score indicating a similarity between a received pilot signal detected in the received signal and a first pilot signal time-shifted by the time difference in a first direction relative to a reference pilot signal, ^ of a second score indicating a similarity between the pilot signal detected in the received signal and a second pilot signal time-shifted by the time difference in a second direction relative to the reference pilot signal, the second direction being opposite to the first direction, and^ of a third score indicating a similarity between the pilot signal detected in the received signal and the reference pilot signal, ^ zeroing (302) of the absolute value of the difference provided that the first score,The second and third scores indicate that there is a greater similarity between the detected pilot signal in the received signal and the reference pilot signal than between the detected signal in the received signal and either of the first time-shifted pilot signal or the second time-shifted pilot signal.

10. Signal processing device (6, 6') comprising:^ an estimator (10, 11) comprising:^ an input for obtaining a signal received by a receiver and resulting from two DC phase-modulated signals previously emitted by two respective transmitters,^ a squaring module configured to square the received signal so as to produce a squared signal,^ a frequency-shifting module configured to produce:^ a first shifted signal having a spectrum shifted by a first frequency step equal to D / 2 with respect to a spectrum of the squared signal, where D is a bit rate of each of the two DC phase-modulated signals,^a second offset signal having a spectrum shifted by a second frequency step equal to -D / 2 with respect to the spectrum of the squared signal, ^a spectral analysis module configured to determine frequencies comprising: ^a first frequency at which a power spectral density of the first offset signal takes a first maximum value, ^a second frequency at which a power spectral density of the second offset signal takes a second maximum value, ^a third frequency at which a power spectral density of the squared signal takes a third maximum value, ^a frequency estimation module configured to estimate a frequency difference between the received signal and each of two phase-modulated signals by determining a measure of the center tendency of the frequencies; ^a frequency corrector (12) configured to correct a frequency of the received signal using the frequency difference,so as to produce a corrected signal; and a demodulator (14) configured to demodulate the corrected signal.

11. Signal processing device (6, 6') according to claim 10, wherein the estimator (11) further comprises: a time estimation module configured to: calculate a power spectral density ratio between the third and fourth power spectral density values ​​depending on at least one of the first and second values; from the power spectral density ratio, estimate the absolute value of a difference between the respective propagation delays of the two continuous phase-modulated signals between the two transmitters and the receiver.

12. Signal processing device (6, 6') according to any one of claims 10 and 11, further comprising: a pilot detector (16) configured to detect a pilot signal in the received signal; a channel estimator (22, 23) configured to estimate,From the pilot signal, the parameters of a channel are used to transmit the two signals using continuous phase modulation, the demodulator (14) being configured to demodulate the corrected signal taking into account the parameters.

13. Signal processing device (6) according to claim 12, further comprising:^ a second frequency corrector (20) configured to correct a frequency of the pilot signal using the frequency deviation, so as to produce a corrected pilot signal, the channel estimation module being configured to estimate the parameters from the corrected pilot signal.

14. Signal processing device (6') according to claim 12, wherein:^ the channel estimation module (23) is configured to determine a sign of the difference between the propagation delays from the pilot signal.

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