Motor control device

Abstract

[Problem] To improve the power factor and distortion of an input current from a power supply which are induced by LC resonance suppression control when using a rectifier circuit, an inverter circuit 3, and an LC filter having a reactor (L) and a capacitor (C) that allows the pulsation of a DC voltage. [Solution] This motor control device comprises: a rectifier circuit that rectifies an alternating current output from a three-phase AC power supply and outputs the rectified current; an inverter circuit that has a plurality of switching elements, converts the direct current output from the rectifier circuit into an alternating current by switching the plurality of switching elements, and supplies the alternating current to a motor; a DC link capacitor that is connected between the input nodes of the inverter circuit, allows fluctuations in the voltage output from the rectifier circuit, and has a capacitance value set so as to suppress a ripple voltage caused by the switching of the inverter circuit; and a reactor that is connected between the three-phase AC power supply and the DC link capacitor and has a capacity value set so as to suppress the ripple current caused by the switching, in which the DC link capacitor and the reactor exhibit a filter function. The motor control device is characterized by performing control so that the magnitude of an output voltage vector changes from positive to negative or from negative to positive with respect to the average value of the output voltage vector of the inverter circuit when the phase Θ of one inter-line voltage of the three-phase AC power supply is within the range satisfying formula 1 at the timing at which the phase Θ is at π / 3 × n (n is an integer of 0 or greater) as a starting point. (Formula 1) n×π / 3≤Θ＜(n＋1)×π / 3 (n is an integer of 0 or more)

Description

Motor control device

[0001] This disclosure relates to a motor control device.

[0002] In a motor control device that functions as an LC filter, comprising a rectifier circuit connected to an AC power supply, an inverter, a capacitor (C), and a reactor (L) connected between the AC power supply and the capacitor, the inverter is controlled to reduce the harmonic components of the DC voltage caused by the resonance of the LC filter. This reduces the harmonic components of the voltage across the capacitor, and consequently reduces the distortion of the current input to the motor control device. Patent Document 1 describes a control method in which a voltage detection device detects the DC link voltage between capacitors, extracts the AC component of the detected voltage using a filter, and suppresses resonance with a compensation amount based on the amount of pulsation. However, depending on the magnitude of the motor load, the phase angle of the inverter output may not be appropriate, causing voltage saturation and thus insufficient resonance suppression. Therefore, the patent document describes a control method that includes a phase-advancing means that outputs a phase-advancing AC component obtained by advancing the AC component extracted by the filter by 90 degrees, a weighting unit that outputs a value obtained by adding a value obtained by multiplying the AC component by a gain p (p is a variable that increases in accordance with the increase in the inverter output and varies in the range of 0 ≤ p ≤ 1) and a value obtained by multiplying the phase-advancing AC component by a gain (1 - p), and a gain that outputs a correction signal obtained by multiplying the output of the weighting unit by a predetermined gain. By adjusting p according to the output of the inverter, a resonance suppression effect can be obtained regardless of the magnitude of the motor load and with a small amount of compensation. Patent Document 2 describes a control method for a power converter in which the voltage (VL) of a reactor is detected with the potential on the capacitor side as the reference potential, and a compensation amount based on the magnitude of VL is subtracted from a voltage control ratio command, which is the ratio of the amplitude of the AC voltage output by the inverter circuit to the average value of the DC voltage, and a switching signal generated based on that voltage control ratio command is supplied to the inverter circuit. The compensation amount is larger as the reactor voltage is larger, and the absolute value of the compensation amount is smaller as the amplitude of the AC current output from the inverter circuit is larger.

[0003] Patent No. 5591215 Patent No. 5712987

[0004] In a motor control device that controls power supplied by an LC filter consisting of a rectifier circuit, a capacitor (C) that tolerates DC voltage pulsation, and a reactor (L), and an inverter circuit, the filter extracts the pulsating component of the DC voltage and compensates for it in the inverter control amount. Compensation control is performed up to the pulsation of harmonic components (m is an integer greater than or equal to 0) that are 6 × m times the power supply frequency, which may result in a decrease in the power factor and increased distortion of the power supply's input current. This disclosure aims to improve the power factor and distortion of the power supply's input current caused by suppressing LC resonance in the LC filter consisting of a rectifier circuit, a capacitor (C) that tolerates DC voltage pulsation, and a reactor (L), and an inverter circuit.

[0005] The motor control device according to the first aspect comprises a rectifier circuit that rectifies and outputs AC power output from a three-phase AC power supply, an inverter circuit having a plurality of switching elements that converts the DC power output from the rectifier circuit into AC power through the switching operation of the plurality of switching elements and supplies it to the motor, a capacitor connected between the input nodes of the inverter circuit and set to a capacitance value that allows fluctuations in the voltage output from the rectifier circuit and suppresses ripple voltage caused by the switching operation of the inverter circuit, and a reactor between the three-phase AC power supply and the capacitor set to a capacitance value that suppresses ripple current caused by the switching operation, wherein the capacitor and reactor function as a filter, and is characterized in that, with respect to the phase Θ of one line voltage of the three-phase AC power supply as a reference, the motor control device controls the magnitude of the output voltage vector from positive to negative or negative to positive, with respect to the average value of the output voltage vector of the inverter circuit when the phase Θ is within the range that satisfies Equation 1, at a timing starting from phase Θ π / 3 × n (where n is an integer of 0 or more), with respect to the phase Θ of one line voltage of the three-phase AC power supply as a reference. Equation 1 is given by n × π / 3 ≤ Θ < (n + 1) × π / 3 (where n is a non-negative integer). In this case, the power factor and distortion of the input current of the power supply can be improved by the suppression control of LC resonance in the LC filter and inverter circuit, which consists of a rectifier circuit, a capacitor (C) that allows DC voltage pulsation, and a reactor (L). The motor control device in the second aspect is the motor control device in the first aspect, characterized in that, with respect to the phase Θ of one line voltage of the three-phase AC power supply, the output voltage of the inverter circuit is controlled so that the output voltage vector pulsates once or more in synchronization with the period of LC resonance by the reactor and capacitor, within the range in which the phase Θ satisfies Equation 2. Equation 2 is given by n × π / 3 ≤ Θ ≤ n × π / 3 + π / 6 (where n is a non-negative integer). A motor control device according to a third perspective is a motor control device according to a first perspective, characterized in that, when controlling the magnitude of the output voltage vector of the inverter circuit from positive to negative or negative to positive, the magnitude of the change in the output voltage vector of the inverter circuit from positive to negative or negative to positive is inversely proportional to the effective value of the current flowing to the motor supplied with AC from the inverter circuit.The motor control device according to the fourth aspect is the motor control device according to the first aspect, characterized in that the DC voltage between capacitors contains 6 × m (m is an integer greater than or equal to 0) harmonic components with the frequency of the line voltage as the fundamental wave, the amplitude of the 6 × m (m is an integer greater than or equal to 0) harmonic is 6% or less of the DC component of the DC voltage, and the output voltage of the inverter circuit is controlled such that the phase of the 6 × m (m is an integer greater than or equal to 0) harmonic is π ± π / 16 with respect to the fundamental wave phase of the line voltage. The motor control device according to the fifth aspect is the motor control device according to the first aspect, characterized in that the transfer characteristic consisting of the DC voltage between capacitors and the maximum value among the absolute values ​​of the three line voltages of the three-phase AC power supply becomes a second-order lag element, and the output voltage of the inverter circuit is controlled such that the damping coefficient of the transfer function of the second-order lag element is greater than 0.7. A motor control device according to the sixth aspect is a motor control device according to the first or fifth aspect, characterized in that it controls the output voltage of an inverter circuit such that the value of the DC voltage between capacitors is less than or equal to the maximum value among the three absolute values ​​of the line voltages of the three-phase AC power supply.

[0006] This figure shows the overall configuration of the motor control device according to this embodiment. This figure shows the line voltage values ​​output from the AC power supply. This figure is for explaining the output voltage vector. This is a control block diagram of the control unit. This figure shows the timing of control. This is an example of a model for controlling the motor control device. This is the original equivalent control model of the motor control device. This is a circuit diagram showing a simplified equivalent circuit of the motor control device in Figure 1. This figure shows comparative example 1 of the control block that controls the circuit shown in Figure 8. This figure shows comparative example 2 of the control block that controls the circuit shown in Figure 8. This figure shows the control block of this embodiment. This figure shows the frequency characteristics of the transfer function when there is no power supply impedance (hereinafter referred to as power supply Z). This figure shows the frequency characteristics of the transfer function when there is a power supply Z. This figure shows the frequency characteristics of the transfer function of a second-order lag element. This figure shows the relationship between the magnitude of power supply Z and the magnitude of the attenuation coefficient. This figure shows the waveform of the DC voltage. This figure shows the harmonic components in this embodiment.

[0007] The embodiments of this disclosure will be described in detail below with reference to the attached drawings. Figure 1 is a diagram showing the overall configuration of the motor control device 1 according to this embodiment. The motor control device 1 receives three-phase AC power from an AC power source 20, converts the supplied AC power, and supplies it to the motor 65. The motor control device 1 comprises a rectifier circuit 2, an inverter circuit 3, a capacitor 4, a reactor 5, and a control unit 10.

[0008] The rectifier circuit 2 is connected to the AC power supply 20 and rectifies the AC output from the three-phase AC power supply. The rectifier circuit 2 is a three-phase full-wave rectifier circuit. The inverter circuit 3 converts the DC output from the rectifier circuit 2 into three-phase AC and supplies it to the motor 65. The three-phase AC output by the inverter circuit 3 is referred to as the u-phase, v-phase, and w-phase from top to bottom in Figure 1. The voltage between the u-phase and v-phase is called the line voltage V. ｕｖ Let the voltage between the v-phase and w-phase be the line voltage V. ｖｗ Let the voltage between the w phase and the u phase be the line voltage V. ｗｕ Let's assume that.

[0009] Capacitor 4 is provided in the DC link section where the rectifier circuit 2 and the inverter circuit 3 are connected. The capacitance value of capacitor 4 is set to allow fluctuations in the voltage output by the rectifier circuit 2 and to suppress ripple voltage caused by the switching operation of the inverter circuit 3. Let the capacitance of the capacitor be C, and the voltage across capacitor 4 be V. ｄｃ The current flowing through capacitor 4 is i ｃ Let i be the current flowing from the DC link section to the inverter circuit 3. ｉｎｖ Capacitor 4 is, for example, a film capacitor.

[0010] The reactor 5 is placed between the three-phase AC power supply and the capacitor 4. The capacitance value of the reactor 5 is set to suppress the ripple current caused by the switching operation of the inverter circuit 3. Let the capacitance value of the reactor 5 be L. In the illustrated example, the reactor 5 is placed between the rectifier circuit and the DC link section, but the reactor 5 may also be placed between the three-phase power supply and the rectifier circuit. This reactor 5 and the capacitor 4 constitute an LC filter.

[0011] The AC power supply 20 is a three-phase AC power supply, which is referred to as the r-phase, s-phase, and t-phase from the upper line in FIG. 1. Here, the voltage between the r-phase and the s-phase is defined as the line voltage V ｒｓ , the voltage between the s-phase and the t-phase is defined as the line voltage V ｓｔ , and the voltage between the t-phase and the r-phase is defined as the line voltage V ｔｒ . Also, the impedances included in the AC power supply 20 itself and the line from the AC power supply 20 to the rectifier circuit 2 are denoted as power supply impedances 21r, 21s, and 21t, as shown in FIG. 1. The value of the impedance per phase varies depending on the surrounding environment. Here, the value of the power supply impedance per phase is denoted as L ａ [H].

[0012] The motor 65 is, for example, an IPM motor (Interior Permanent Magnet Motor). The motor 65 drives the compressor provided in the refrigerant circuit.

[0013] The control unit 10 controls the inverter circuit 3 and efficiently operates the motor 65 by converting the alternating current supplied to the motor 65. The control unit 10 includes a device for processing various information. The control unit 10 can be configured using a microcomputer and a memory device or the like that stores software for operating it. The control unit 10 includes a phase detection unit 11 and a voltage detection unit 12. The phase detection unit 11 detects at least one line voltage of the three-phase AC power supply, converts it into a pulse by processing in a phase detection circuit, and transmits it to the control unit 10. In the illustrated example, the phase detection unit 11 detects the line voltage between the r-phase and the s-phase. The voltage detection unit 12 detects the voltage across the capacitor 4.

[0014] FIG. 2 is a diagram showing the values of the line voltages output from the AC power supply 20. In FIG. 2, the values of the line voltages output from the three-phase AC power supply are shown on the vertical axis. In FIG. 2, the value of the line voltage V ｒｓ is shown by a solid line, the value of the line voltage V ｓｔ is shown by a dashed line, and the value of the line voltage V ｔｒ is shown by a dotted-dashed line. Also, the maximum value of the absolute values of the three line voltages is shown by a dashed line. In the example shown in FIG. 2, the line voltage Vｒｓ The value of is 0V, and the line voltage V ｒｓ The timing of the change from negative to positive is defined as 0°. Line voltage V ｒｓ and line voltage V ｓｔ The phase difference is 240 degrees, and the line voltage V ｒｓ and line voltage V ｔｒ The phase difference is 120 degrees. As described above, capacitor 4 (see Figure 1), shown in Figure 1, allows for fluctuations in the voltage output by the rectifier circuit 2 (see Figure 1). For this reason, capacitor 4 does not smooth the three-phase full-wave rectification output from the rectifier circuit 2, and a potential equal to the maximum of the absolute values ​​of the three line voltages is generated at capacitor C, pulsating at a frequency six times that of the AC power supply 20.

[0015] Here, we define the output voltage vector of the inverter. Figure 3 is a diagram illustrating the output voltage vector. The circuit diagram in Figure 3 shows the state in which the capacitance C of capacitor 4 in Figure 1 is set to 2C and divided into two capacitors 4a. Using the potential between these two divided capacitors 4a as a reference, the voltage vector of the u phase is defined as V ｕｎ Similarly, using the potential across capacitor 4a as a reference, the voltage vector of the v-phase is V ｖｎ Let the voltage vector of the w phase be V ｗｎ Let's assume this voltage vector V ｕｎ And, V ｖｎ And, V ｗｎ Using these, the output voltage vector of the inverter is defined by equation (1-1). Also, the magnitude of the output voltage vector V o This is defined by equation (1-2).

[0016]

[0017] Figure 4 is a control block diagram of the control unit 10 (see Figure 1). The control unit 10 comprises a phase estimation unit 101, a voltage averaging unit 102, a voltage command generation unit 103, a comparison unit 104, a gain correction unit 105, a multiplier 106, a manipulated variable limiter 107, and a modulation rate command generation unit 108.

[0018] [Phase Estimation Unit] The phase estimation unit 101 estimates the phase of the AC power supply 20. The phase estimation unit 101 acquires the pulse transmitted from the phase detection unit 11 and calculates the phase. The method for estimating the phase is not particularly limited, and the voltage V across the capacitor 4 is also used. ｄｃ Detects voltage V ｄｃ By filtering the voltage V ｄｃ The pulsations may be extracted, and the frequency and phase may be calculated from the pulsation components.

[0019] [Voltage Averaging Section] The voltage averaging section 102 measures the voltage V ｄｃ The average value is calculated. The voltage averaging unit 102 calculates the average value of the detected voltage V ｄｃ Passing this through a low-pass filter, the voltage V ｄｃ Average voltage V ＤＣＡＶＥ The voltage averaging unit 102 may also average instantaneous values ​​from the phase estimation unit 101 when the phase is between 60n° and 60(n-m)° (n and m are integers), using the phase of the line voltage as a reference.

[0020] [Voltage Command Generation Unit] The voltage command generation unit 103 receives the power supply voltage estimation phase θ from the phase estimation unit 101. ＶＤＣ The voltage average value V is obtained from the voltage averaging unit 102. ＤＣＡＶＥ The voltage command generation unit 103 obtains the power supply voltage estimated phase θ and the voltage average value V. ＤＣＡＶＥ From this, an ideal three-phase full-wave rectifier is generated, and this ideal three-phase full-wave rectifier is used as a voltage command V ｄｃ ＊ Let's assume that.

[0021] [Comparison Unit] The comparison unit 104 processes the voltage command V ｄｃ ＊ The voltage detection unit 12 detected the detected value V ｄｃ It compares the two and extracts the difference. Specifically, the comparison unit 104 processes the voltage command V ｄｃ ＊ From the detected value V ｄｃ Subtract it.

[0022] [Gain Correction Unit] The gain correction unit 105 controls the resonance suppression gain k ＦＢ Corrects i. ｉｎｖ Resonance control is performed by controlling the resonance suppression gain k ＦＢCorrects i. ｉｎｖ Resonance suppression is performed by controlling i ｉｎｖ Since direct control is difficult, resonance suppression is performed by manipulating the output voltage vector of the inverter. In this case, i ｉｎｖ The magnitude of pulsation (resonance suppression gain k) assuming manipulation ＦＢ Controlling the output voltage vector with ) causes excessive pulsation. Therefore, the effective value of the motor current I ｒｍｓ Accordingly, the magnitude of the pulsation in the output voltage vector (resonance suppression gain k) ＦＢ ) is corrected by. Specifically, the gain correction unit 105 uses the resonance suppression gain k ＦＢ Using as the numerator, then the effective value of the motor current I ｒｍｓ Perform division with the denominator being [the value of the denominator].

[0023] [Manipulated Variable Limiter] The manipulated variable limiter 107 has defined upper and lower limits for the manipulated variable, so as not to interfere with the manipulated variable that controls the fundamental wave of the motor current, and so as not to output abnormal numerical results.

[0024] [Modulation Rate Command Generation Unit] The modulation rate command generation unit 108 (subtractor) generates a modulation rate command value K that controls the fundamental wave of the motor current. ｓ From ', modulation rate correction value K ｓ By subtracting '', the modulation rate command value K ｓ ＊ Outputs the magnitude V of the output voltage vector. ｏ The modulation rate command value K ｓ ＊ It is proportional to.

[0025] [Control Timing] Figure 5 is a diagram showing the timing of control. The control unit 10 receives a voltage command V ｄｃ ＊ It has a voltage command V ｄｃ ＊ and detected value V ｄｃ The difference is extracted and compensated for. The extracted difference represents the resonant component. The control unit 10 corrects the modulation rate based on the extracted resonant component, thereby reducing the pulsation caused by resonance. ｉｎｖ This is used to press against the DC link section and suppress resonance.

[0026] In Figure 5, the left vertical axis shows the magnitude of the DC voltage [V], and the right vertical axis shows the magnitude of the inverter output voltage vector [V]. The horizontal axis shows time [s]. In Figure 5, the detected value of the DC voltage is V. ｄｃ This is shown by a solid line. The command value V generated by the voltage command generation unit 103. ｄｃ ＊ These are shown by dotted lines. The instantaneous value of the output vector is shown by a dashed line, and the average value of the output voltage vector is shown by a dashed line. In Figure 5, the vertical dashed lines indicate the phase of the DC voltage command. The phase of the DC voltage command is the phase estimated by the phase estimation unit 101 (see Figure 4). In Figure 5, the vertical double-dotted lines indicate a portion of the timing when the output voltage vector is pulsating.

[0027] The control unit 10 (see Figure 4) uses the phase of one line voltage of the three-phase AC power supply as a reference and controls the output voltage vector so that it changes from positive to negative or negative to positive, based on the average value of the inverter's output voltage vector between the phases of n × (π / 3) and (n+1) × (π / 3), starting at a phase of n × (π / 3) (where n is an integer).

[0028] Here, a positive magnitude for the output voltage vector relative to the average value of the inverter's output voltage vector means that the magnitude of the output voltage vector is greater than the average value of the output voltage vector. Conversely, a negative magnitude for the output voltage vector relative to the average value of the inverter's output voltage vector means that the magnitude of the average output voltage vector is smaller.

[0029] Furthermore, the control unit 10 may control the inverter's output vector from positive to negative, or from negative to positive, at more timings than (π / 3) × n timings from the reference phase. In this embodiment, the output magnitude of the output voltage vector changes depending on the difference between the DC voltage command and the detected DC voltage, so the output line voltage vector pulsates in synchronization with the LC filter's resonance period. In the example shown in the figure, the output voltage vector pulsates one or more times in synchronization with the LC resonance period within the phase range n × (π / 3) ≤ θ ≤ n × (π / 3) + (π / 6).

[0030] [Details of gain correction] Next, the gain correction unit 105 adjusts the resonance suppression gain k ＦＢ Using this as the numerator, the RMS value of the motor current I ｒｍｓ Let's explain the point about dividing with as the denominator. Figure 6 is an example of a model that controls the motor control device 1. Figure 6 shows i ｉｎｖ ' represents the current value determined by motor control. The i shown in Figure 6 is an example of this. ｉｎｖ '' is the value of the current manipulated by resonance suppression control. Also, i ｉｎｖ And, i ｉｎｖ ' and i ｉｎｖ The relationship with '' is given by equation (1-3).

[0031]

[0032] Figure 7 shows the original equivalent control model of the motor control device 1. Figure 7 shows the output voltage V ０ When applied to the motor load i ｉｎｖ This is determined. Also, the motor load value changes depending on the motor's operating point. V in Figure 7 ０ ' represents the voltage value determined by the motor's operation. ０ '' is the voltage value manipulated by resonance suppression control. Also, V ０ and V ０ ' and V ０ The relationship with '' is given by equation (1-4).

[0033]

[0034] Here, considering the motor load, V ０ Consider the magnitude of the pulsation. V０ The magnitude of the pulsation is determined by the modulation rate correction value K of the inverter inside the control, ｓ so the resonance suppression gain correction value k ＦＢＶＤＣ can be changed to adjust the magnitude of the pulsation of V. The subtractor 203 and the resonance suppression gain 272 in FIG. 6 lead to Equation (1-5) according to Equation (1-3). ０

[0035]

[0036] Also, from FIG. 4, Equation (1-6) is derived.

[0037]

[0038] Further, the subtractor 203 and the gain 271 in FIG. 7 lead to Equation (1-7) according to Equation (1-4).

[0039] Here, assuming that the inverter efficiency is 1 and the motor power factor is 1, the motor current execution value I ｒｍｓ and the inverter current i ｉｎｖ and the inverter output voltage V ０ and the voltage V ｄｃ are related by the following Equation (1-8).

[0040]

[0041] By transforming Equation (1-8), the relational expression of the output voltage V ０ the voltage V ｄｃ the modulation rate command value K ｓ ＊ is derived as Equation (1-9).

[0042]

[0043] By applying Equation (1-5) and Equation (1-6) to Equation (1-8) and dividing both sides by V ｄｃ Equation (1-11) is derived.

[0044] Here, the second term on the left side and the second term on the right side of equation (1-11) are terms determined by the magnitude of the pulsation due to resonance suppression. Also, the first term on the left side and the first term on the right side are terms determined by the motor's operating point. If we extract the second term on the left side and the second term on the right side from the magnitude of the pulsation due to resonance suppression, we obtain the following equation (1-12).

[0045]

[0046] Thus, k ＦＢＶＤＣ The correction value is I ｒｍｓ It is inversely proportional to.

[0047] Figure 8 is a circuit diagram showing a simplified equivalent circuit of the motor control device 1 in Figure 1. Voltage V ｉｎ i is the maximum value among the absolute values ​​of the three line voltages of the three-phase AC power supply 20. Ｌ This is the current flowing through reactor 5. Voltage V ｓ This is the DC voltage output from the rectifier circuit 2. Voltage V ｄｃ This is the voltage across capacitor 4. Current i ｃ This is the current flowing through capacitor 4. Current i ｉｎｖ This is the current flowing through inverter circuit 3.

[0048] Figure 9 shows Comparative Example 1 of a control block that controls the circuit shown in Figure 8. This Comparative Example 1 is a control model that does not perform resonance suppression control. Comparative Example 1 comprises three subtractors 201, 202, and 204 and three transfer functions G1, G2, and G3. Subtractor 201 controls the current i Ｌ This is fed back through the transfer function G1, V ｉｎ Subtract from the voltage V ｓ It outputs the following. The subtractor 202 outputs voltage V ｄｃ The voltage V is fed back. ｓ Voltage V ｄｃ Subtracting this gives the voltage V Ｌ It outputs the following. The transfer function G2 is the voltage V Ｌ Enter the current i Ｌ It outputs the current i. Subtractor 204 outputs the current i Ｌ From, current i ｉｎｖ Subtracting this, the current i ｃIt outputs the following. The transfer function G3 is given by the current i ｃ Enter V ｄｃ Outputs.

[0049] This control block is organized to show the transfer function G which represents the frequency characteristics. ＮｏＤａｍｐ (s) = V ｄｃ / V ｉｎ When we find this, we get equation (1-13).

[0050]

[0051] Figure 10 shows Comparative Example 2 of the control block that controls the circuit shown in Figure 8. In this Comparative Example 2, the voltage V across the reactor 5 is different from that in Comparative Example 1. Ｌ and resonance suppression gain k ＦＢ Multiplying by this, we get the current i ｉｎｖ It differs in that it outputs [something]. Comparative Example 2 comprises three subtractors 201, 202, and 204, three transfer functions G1, G2, and G3, and a resonance suppression gain 272.

[0052] The subtractor 201 processes the current i Ｌ This is fed back through the transfer function G1, V ｉｎ Subtract from the voltage V ｓ It outputs the following. The subtractor 202 outputs voltage V ｄｃ The voltage V is fed back. ｓ Voltage V ｄｃ Subtracting this gives the voltage V Ｌ It outputs the following. The transfer function G2 is the voltage V Ｌ Enter the current i Ｌ Outputs the following. Gain 272 is for voltage V Ｌ and gain k ＦＢ Multiply by the current i ｉｎｖ It outputs the current i. Subtractor 204 outputs the current i Ｌ From, current i ｉｎｖ Subtracting this, the current i ｃ It outputs the following. The transfer function G3 is given by the current i ｃ Enter V ｄｃ Outputs.

[0053] The control block of this comparative example 2 is organized to show the transfer function G exhibiting frequency characteristics. ＶＬ (s) = V ｄｃ / V ｉｎ When we find this, we get equation (1-14).

[0054]

[0055] Figure 11 shows the control block of this embodiment. The control block of this embodiment comprises four subtractors 201, 202, 203, and 204, three transfer functions G1 to G3, and a gain 273. Subtractor 201 controls the current i Ｌ This is fed back through the transfer function G1, V ｉｎ Subtract from the voltage V ｓ It outputs the following. The subtractor 202 outputs voltage V ｄｃ The voltage V is fed back. ｓ Voltage V ｄｃ Subtracting this gives the voltage V Ｌ The subtractor 203 outputs the command value V. ｄｃ ＊ From the detected value V dｃ Subtract the value. Gain 273 is equal to the command value V ｄｃ ＊ From the detected value V dｃ The value obtained by subtracting the gain k ＦＢ Multiply by the current i ｉｎｖ It outputs the current i. Subtractor 204 outputs the current i Ｌ From, current i ｉｎｖ Subtracting this, the current i ｃ It outputs the following. The transfer function G3 is given by the current i ｃ Enter V ｄｃ Outputs.

[0056] The control block of this embodiment is organized to show the transfer function G exhibiting frequency characteristics. ＶＤＣ (s) = V ｄｃ / V ｉｎ When we find this, we get equation (1-15).

[0057]

[0058] Here, equations (1-13), (1-14), and (1-15) are used to show the frequency characteristics of the transfer function. Figure 12 shows the frequency characteristics of the transfer function when there is no power supply Z. The horizontal axis shows frequency [Hz], and the vertical axis shows gain [dB] and phase [deg]. Comparative Example 2 is shown by a solid line, this embodiment is shown by a dashed line, and Comparative Example 1 is shown by a dotted line. When there is no power supply Z, Comparative Example 1, which does not perform resonance suppression, shows an increase in gain at the resonant frequency. On the other hand, both Comparative Example 2, which performs resonance control, and this embodiment, which also performs resonance control, have a gain of 0 or less at the resonant frequency, thus suppressing the increase in gain.

[0059] Figure 13 shows the frequency characteristics of the transfer function when a power supply Z is present. Similar to Figure 12, Figure 13 shows frequency [Hz] on the horizontal axis and gain [dB] and phase [deg] on the vertical axis. Comparative Example 2 is shown by a solid line, this embodiment by a dashed line, and Comparative Example 1 by a dotted line. In the examples shown in the figure, the frequency characteristics of the transfer function are shown when the magnitude of the power supply Z is 1.5 [mH]. When a power supply Z is present, in both Comparative Examples 1 and 2, the resonant component at the resonant frequency is not attenuated, and a peak is observed. On the other hand, in the resonance control of this embodiment, the resonant component is attenuated. In the resonance control of this embodiment, since the gain characteristic is 0 [dB] or less, the maximum value of the DC voltage is less than or equal to the maximum value of the absolute values ​​of the multiple power line voltages.

[0060] Figure 14 shows the frequency characteristics of the transfer function of a second-order lag element. In Figure 14, the horizontal axis shows frequency [Hz], and the vertical axis shows gain [dB] and phase [deg]. The solid line shows the frequency characteristics when the damping coefficient is 10. The dashed line shows the frequency characteristics when the damping coefficient is 3, and the dashed line shows the frequency characteristics when the damping coefficient is 1. Furthermore, the long dashed line shows the frequency characteristics when the damping coefficient is 0.7, the dashed line shows the frequency characteristics when the damping coefficient is 0.4, and the dotted line shows the frequency characteristics when the damping coefficient is 0.1. As shown in Figure 14, the gain at the resonant frequency increases when the damping coefficient is 0.1 and when the damping coefficient is 0.4. On the other hand, when the damping coefficient is 0.7 or higher, the gain is 0 or less, and the gain at the resonant frequency does not increase.

[0061] Figure 15 shows the relationship between the magnitude of the power supply Z and the magnitude of the attenuation coefficient ξ. In Figure 15, the horizontal axis shows the power supply Z [mH] and the vertical axis shows the magnitude of the attenuation coefficient ξ. Comparative Example 2 is shown by a solid line and this embodiment is shown by a dashed line. Here, the attenuation coefficient ξ of Comparative Example 2 ＶL This can be expressed by equation (1-16), based on equation (1-14).

[0062]

[0063] Furthermore, the damping coefficient ξ of this embodiment ＶＤＣ This can be expressed by equation (1-17) based on equation (1-15).

[0064]

[0065] Figure 15 shows the resonant gain k. ＦＢWhen the power supply Z is uniquely determined, it is shown that in Comparative Example 2 and this embodiment, the damping coefficient ξ changes as the power supply Z increases. In the example shown in the figure, in this embodiment, the value is 0.7 or more when the power supply Z is 0, and the value of the damping coefficient ξ increases as the power supply Z increases. The value of the power supply Z changes depending on the commercial power supply environment and is often unknown. If the damping coefficient ξ is designed to be 0.7 when the power supply Z is 0, the damping coefficient ξ will be 0.7 or more for any power supply Z, and the resonance component can be suppressed. On the other hand, if the damping coefficient ξ is designed to be greater than 0.7 when the power supply Z is 0, the gain will be excessive when the power supply Z is 0, and the control will become unstable.

[0066] [DC Voltage Waveform] Figure 16 shows the DC voltage waveform. The upper table in Figure 16 shows the case without a power supply Z, and the lower table shows the case when the power supply Z is 0.5 [mH]. In the table, time [s] is shown on the horizontal axis and the magnitude of the DC voltage [V] is shown on the vertical axis. The DC voltage waveform of Comparative Example 2 is shown by a solid line, and the DC voltage waveform of this embodiment is shown by a dotted line.

[0067] As shown in the lower waveform of Figure 16, when the power supply Z is 0.5 [mH], the difference between the maximum and minimum DC voltages in Comparative Example 2 is larger than the difference between the maximum and minimum DC voltages in this embodiment. In the resonance control of this embodiment, as shown in Figure 13, the gain characteristic is 0 [dB] or less even when the power supply Z is 1.5 [mH], so the maximum DC voltage is less than or equal to the maximum absolute value of the multiple power supply line voltages. On the other hand, in Comparative Example 2, as shown in Figure 13, when there is a power supply Z, the gain becomes large at the resonant frequency, and the resonant component cannot be sufficiently attenuated. Therefore, in Comparative Example 2, as shown in the lower waveform of Figure 16, the maximum DC voltage is larger than the maximum absolute value of the multiple power supply line voltages.

[0068] [Harmonic Components] Figure 17 shows the harmonic components included in the DC voltage waveform in this embodiment. In Figure 17, the horizontal axis shows the phase [deg], and the vertical axis shows the ratio of the DC component to the harmonic component of the DC voltage. In this embodiment, since the capacitor 4 (see Figure 1) does not smooth the DC voltage, the DC voltage contains 6 × m-th order harmonic components (m is an integer greater than or equal to 0) with the frequency of the power line voltage as the fundamental wave. The DC voltage can be expressed by separating it into a DC component and a harmonic component using the following formula.

[0069]

[0070] The first term on the right-hand side of equation (1-18) is the DC component of the DC voltage. The second term on the right-hand side of equation (1-18) is the 6 × m harmonic component. Furthermore, the third term and subsequent terms on the right-hand side of equation (1-18) include the carrier component and other components. Figure 17 shows this DC component V. ｄｃ０ and harmonic component V ｄｃ６ｍ cos(6mω) ｇ t + θ ６ｍ The ratio to ) is shown, and the harmonic components up to m = 3 are shown.

[0071] Figure 17 shows the reference phase with a dotted line. Here, the reference phase is the phase of one line voltage of the AC power supply 20. In Figure 17, the 6th harmonic is shown with a solid line, the 12th harmonic with a dashed line, and the 18th harmonic with a dashed line. As shown in Figure 17, the ratio of the 6th harmonic to the DC component of the DC voltage is kept within 6%. The ratio of the 12th harmonic to the DC component of the DC voltage is kept within 2%. Furthermore, the ratio of the 18th harmonic to the DC component of the DC voltage is kept within 1%.

[0072] The DC voltage can be controlled within the ranges of equations (1-19) and (1-20) by a command value of a DC voltage having the amplitude and phase of the 6 × m-th harmonic component (where m is an integer greater than or equal to 0) of an ideal three-phase full-wave rectified waveform.

[0073]

[0074] [Effects] The motor control device 1 of this disclosure includes a rectifier circuit 2 that rectifies and outputs AC power output from a three-phase AC power supply 20, an inverter circuit 3 having a plurality of switching elements that converts the DC power output by the rectifier circuit 2 into AC power through the switching operation of the plurality of switching elements and supplies it to the motor, a capacitor 4 connected between the input nodes of the inverter circuit 3, which allows for voltage fluctuations output by the rectifier circuit 2 and has a capacitance value set to suppress ripple voltage caused by the switching operation of the inverter circuit 3, and a switch between the three-phase AC power supply 20 and the capacitor 4. A motor control device comprising a reactor 5 whose capacitance value is set to suppress ripple current caused by pulsation, and a capacitor 4 and a reactor 5 which function as a filter, controls the output voltage vector of the inverter circuit 3 to change from positive to negative or from negative to positive, based on the average value of the output voltage vector of the inverter circuit 3 when the phase Θ is in the range of n × π / 3 ≤ Θ < (n + 1) × π / 3 (n is an integer of 0 or more), with the phase Θ starting at π / 3 × n (where n is an integer of 0 or more), using the phase Θ of one line voltage of the three-phase AC power supply 20 as a reference. The motor control device improves the distortion of the power supply input current caused by the suppression control of LC resonance by the rectifier circuit 2, the capacitor 4 which allows DC voltage pulsation, the reactor 5 which forms an LC filter, and the inverter circuit 3. Here, the motor control device 1 controls the output voltage of the inverter circuit 3 so that the output voltage vector pulsates at least once in synchronization with the period of the LC resonance between the reactor 5 and the capacitor 4, within the range where the phase Θ of one line voltage of the three-phase AC power supply 20 satisfies n×π / 3≦Θ≦n×π / 3+π / 6 (where n is a non-negative integer), with respect to the phase Θ of one line voltage of the three-phase AC power supply 20. Here, when the motor control device 1 controls the magnitude of the output voltage vector of the inverter circuit 3 from positive to negative or negative to positive, it controls it so that the magnitude of the change in the output voltage vector of the inverter circuit 3 from positive to negative or negative to positive is inversely proportional to the effective value of the current flowing to the motor supplied with AC from the inverter circuit 3. In this case, the stability of the control or the performance of resonance suppression is prevented.Here, the motor control device 1 controls the output voltage of the inverter circuit 3 such that the DC voltage across the capacitors 4 contains 6 × m-th order (m is an integer greater than or equal to 0) harmonic components with the frequency of the line voltage as the fundamental wave, the amplitude of the 6 × m-th order (m is an integer greater than or equal to 0) harmonic being 6% or less of the DC component of the DC voltage, and the phase of the 6 × m-th order (m is an integer greater than or equal to 0) harmonic being π ± π / 16 with respect to the fundamental wave phase of the line voltage. In this case, the power factor of the input current on the power supply side is improved and power supply harmonics are suppressed. The motor control device 1 is characterized by controlling the output voltage of the inverter circuit 3 such that the transfer characteristic consisting of the DC voltage across the capacitors 4 and the maximum value among the absolute values ​​of the three line voltages of the three-phase AC power supply 20 becomes a second-order lag element, and the damping coefficient of the transfer function of the second-order lag element is greater than 0.7. In this case, the decrease in resonance suppression performance is suppressed. The motor control device 1 controls the output voltage of the inverter circuit 3 so that the DC voltage across the capacitors 4 is less than or equal to the maximum value among the three absolute values ​​of the line voltages of the three-phase AC power supply 20. In this case, the deterioration of resonance suppression performance is suppressed.

[0075] Although embodiments have been described above, the technical scope of this disclosure is not limited to the embodiments described above. It is clear from the claims that combinations of two or more of the above embodiments, as well as various modifications or improvements to the above embodiments, are also included in the technical scope of this disclosure.

[0076] 1...Motor control device, 2...Rectifier circuit, 3...Inverter circuit, 4...Capacitor, 5...Reactor, 10...Control unit, 11...Phase detection unit, 12...Voltage detection unit, 20...AC power supply, 21r, 21s, 21t...Power supply impedance, 65...Motor, 101...Phase estimation unit, 102...Voltage averaging unit, 103...Voltage command generation unit, 104...Comparison unit, 105...Gain correction unit, 106...Multiplier, 107...Operation variable limiter, 108...Modulation rate command generation unit

Claims

1. A motor control device comprising: a rectifier circuit that rectifies and outputs AC power output from a three-phase AC power supply; an inverter circuit having a plurality of switching elements that converts the DC power output from the rectifier circuit into AC power through the switching operation of the plurality of switching elements and supplies it to a motor; a DC link capacitor connected between the input nodes of the inverter circuit, which allows fluctuations in the voltage output from the rectifier circuit and has a capacitance value set to suppress ripple voltage caused by the switching operation of the inverter circuit; and a reactor between the three-phase AC power supply and the DC link capacitor, which has a capacitance value set to suppress ripple current caused by the switching operation, wherein the DC link capacitor and the reactor function as a filter, and the motor control device is characterized in that, with respect to the phase Θ of one line voltage of the three-phase AC power supply as a reference, the output voltage vector is controlled to change from positive to negative or from negative to positive, with respect to the average value of the output voltage vector of the inverter circuit when the phase Θ is within the range that satisfies equation 1, at a timing starting from phase Θ π / 3 × n (where n is an integer greater than or equal to 0) with respect to the phase Θ of one line voltage of the three-phase AC power supply. (Equation 1) n × π / 3 ≤ Θ < (n + 1) × π / 3 (where n is a non-negative integer) 2. The motor control device according to claim 1, characterized in that, with respect to the phase Θ of one of the line voltages of the three-phase AC power supply, the output voltage of the inverter circuit is controlled such that the output voltage vector pulsates once or more in synchronization with the period of LC resonance by the reactor and the DC link capacitor, within the range in which the phase Θ satisfies Equation 2. (Equation 2) n × π / 3 ≤ Θ ≤ n × π / 3 + π / 6 (where n is an integer greater than or equal to 0) 3. The motor control device according to claim 1, characterized in that when controlling the magnitude of the output voltage vector of the inverter circuit from positive to negative or negative to positive, the magnitude of the change in the output voltage vector of the inverter circuit from positive to negative or negative to positive is controlled to be inversely proportional to the effective value of the current flowing to the motor supplied with AC from the inverter circuit.

4. The motor control device according to claim 1, characterized in that the DC voltage between the DC link capacitors contains a 6 × m (where m is an integer greater than or equal to 0) harmonic component with the frequency of the line voltage as the fundamental wave, the amplitude of the 6 × m (where m is an integer greater than or equal to 0) harmonic is 6% or less of the DC component of the DC voltage, and the output voltage of the inverter circuit is controlled such that the phase of the 6 × m (where m is an integer greater than or equal to 0) harmonic is π ± π / 16 with respect to the fundamental wave phase of the line voltage.

5. The motor control device according to claim 1, characterized in that the transfer characteristic consisting of the DC voltage between the DC link capacitors and the maximum value among the three absolute values ​​of the line voltages of the three-phase AC power supply forms a second-order lag element, and the output voltage of the inverter circuit is controlled such that the damping coefficient of the transfer function of the second-order lag element is greater than 0.

7.

6. The motor control device according to claim 1 or 5, characterized in that the output voltage of the inverter circuit is controlled such that the DC voltage between the DC link capacitors is less than or equal to the maximum value of the three absolute values ​​of the line voltages of the three-phase AC power supply.

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