Power conversion apparatus

Abstract

[Problem] To provide a power conversion apparatus capable of achieving operation in a wider range by further increasing a modulation factor while an effect of suppressing common-mode noise is substantially maintained. [Solution] A power conversion apparatus comprises a control device 21 that controls the switching of switching elements. When a command voltage vector is inside a basic voltage space and outside a modulatable region, a modulation unit 35 of the control device 21 switches between and executes a first pulse width modulation for regulating the command voltage vector to be inside the modulatable region, and a second pulse width modulation for fixing the on / off states of upper and lower arm switching elements of two phases and modulating the on / off states of upper and lower arm switching elements of the other one phase.

Description

Power converter

[0001] This invention relates to a power conversion device that converts a DC voltage to an AC voltage.

[0002] Various pulse width modulation (PWM) methods have been proposed to suppress conducted noise propagating through power supplies, but these methods can be broadly divided into two categories. One method completely suppresses fluctuations in common-mode voltage, which are a source of common-mode noise, while the other method partially suppresses the noise while tolerating fluctuations in common-mode voltage.

[0003] One of the former methods is pulse width modulation, which outputs only odd voltage vectors or only even voltage vectors. This method makes it possible to completely suppress fluctuations in common-mode voltage within the carrier period. There is also pulse width modulation that switches between outputting only odd voltage vectors or only even voltage vectors depending on the electrical angular phase. This method can also significantly suppress fluctuations in common-mode voltage (see, for example, Patent Documents 1 and 2).

[0004] Another example of the latter method is pulse width modulation of two-phase modulation, in which the switching of one phase is fixed and the other two phases are switched. This method can suppress fluctuations in the common-mode voltage (see, for example, Patent Document 3).

[0005] While the former method (Patent Documents 1 and 2) is the most effective method for suppressing common-mode voltage fluctuations, it has the drawback of limiting the linear output region (the maximum amplitude at which the voltage vector can complete one rotation with a constant radius) and thus limiting the possible modulation rate due to restrictions on the voltage vector used. Therefore, it is difficult to apply to applications such as driving the motor of an electric compressor.

[0006] In contrast, the latter method (Patent Document 3) allows the linear output region to be used up to its normal maximum, and can achieve a high modulation rate, but its effect in suppressing common-mode voltage fluctuations is still inferior to that of the former method.

[0007] Therefore, in addition to the control described in Patent Documents 1 and 2 above, a method has been proposed in which, when the command voltage vector is outside the modulatorable region in which the modulation unit can modulate within the basic voltage space, the command voltage vector is modified to be within the modulatorable region, and this modified command voltage vector is used for output (Patent Document 4).

[0008] Japanese Patent Publication No. 5397448, Japanese Patent Publication No. 7394619, Japanese Unexamined Patent Publication No. 2022-33476, Japanese Unexamined Patent Publication No. 2024-40733

[0009] According to the method described in Patent Document 4, it is possible to increase the output voltage while suppressing the excitation of common-mode noise, as in the methods described in Patent Documents 1 and 2. However, there are limits to further increasing the modulation ratio to raise the output voltage, and there has been a desire for the development of a method that can raise the modulation ratio and output voltage even further.

[0010] The present invention was made to solve the aforementioned conventional technical problems, and aims to provide a power converter that can achieve a higher modulation rate and operate over a wider range while substantially maintaining the common-mode noise suppression effect.

[0011] The power conversion device of the present invention converts a DC voltage to an AC voltage and comprises an inverter circuit that applies phase voltages at the connection points of the upper and lower arm switching elements of each phase to a load, and a control device that controls the switching of each switching element. The control device includes a modulation unit that defines a predetermined region of the basic voltage space, which is the outputtable voltage vector region in the inverter circuit, as a moduloable region. The modulation unit is characterized by switching between and executing a first pulse width modulation that corrects the command voltage vector to be within the moduloable region when the command voltage vector is within the basic voltage space and outside the moduloable region, and a second pulse width modulation that fixes the ON / OFF state of two phase upper and lower arm switching elements and modulates the ON / OFF state of the other phase upper and lower arm switching element.

[0012] The power conversion device of the second invention is characterized in that, in addition to the present invention described above, the modulation unit switches between and performs pulse width modulation that outputs only odd voltage vectors in one control cycle and pulse width modulation that outputs only even voltage vectors in one control cycle, based on the electrical angular phase, in the first pulse width modulation.

[0013] The power conversion device of the third invention is characterized in that, in addition to the second invention described above, the modulation unit, in the first pulse width modulation, modulates the command voltage vector to a moduloable region obtained by superimposing a moduloable region for pulse width modulation that outputs only odd voltage vectors in one control cycle and a moduloable region for pulse width modulation that outputs only even voltage vectors in one control cycle, when the command voltage vector is in the fundamental voltage space and outside the moduloable region.

[0014] The power conversion device of the fourth invention is characterized in that, in addition to the present invention, the modulation unit calculates a three-phase modulation command value for generating the phase voltage of each phase, turns on the lower arm switching element of the phase with the smallest three-phase modulation command value to stop switching, or turns on the upper arm switching element of the phase with the largest three-phase modulation command value to stop switching, and switches between a first pulse width modulation and a second pulse width modulation according to the modulation rate and electrical angular phase of this two-phase modulation command value.

[0015] The power conversion device of the fifth invention is characterized in that, in addition to the fourth invention, the modulation unit performs a first pulse width modulation in an electrical angular phase where, depending on the modulation rate, it stops switching by turning on the lower arm switching element of only the phase in which the two-phase modulation command value is smallest, or by turning on the upper arm switching element of only the phase in which the two-phase modulation command value is largeest.

[0016] The power conversion device of the sixth invention is characterized in that, in addition to the fifth invention, the modulation unit performs pulse width modulation in the first pulse width modulation, where, in the electrical angular phase where the lower arm switching element of the phase with the minimum two-phase modulation command value is turned ON and switching is stopped, it outputs only even voltage vectors during one control cycle, and in the electrical angular phase where the upper arm switching element of the phase with the maximum two-phase modulation command value is turned ON and switching is stopped, it outputs only odd voltage vectors during one control cycle.

[0017] The power conversion device of the seventh invention, in addition to the sixth invention described above, is characterized in that, in the first pulse width modulation, when the command voltage vector is within the fundamental voltage space and outside the modulatorable region, the lower arm switching element of the phase with the minimum two-phase modulation command value is turned ON to stop switching, and the command voltage vector is modified to fit within the modulatorable region of pulse width modulation that outputs only even voltage vectors in one control cycle; and the upper arm switching element of the phase with the maximum two-phase modulation command value is turned ON to stop switching, and the command voltage vector is modified to fit within the modulatorable region of pulse width modulation that outputs only odd voltage vectors in one control cycle.

[0018] The power conversion device of the eighth invention is characterized in that, in addition to the fourth invention, the modulation unit performs a second pulse width modulation at an electrical angular phase in which the lower arm switching element or the upper arm switching element of the two phases of the two-phase modulation command value is turned ON and switching is stopped, depending on the modulation rate.

[0019] The power conversion device of the ninth invention is characterized in that, in each of the above inventions, the inverter circuit drives the motor by applying the phase voltage at the connection point of the upper and lower arm switching elements of each phase.

[0020] According to the present invention, a power conversion device that converts a DC voltage to an AC voltage includes an inverter circuit that applies phase voltages at the connection points of the upper and lower arm switching elements of each phase to a load, and a control device that controls the switching of each switching element. This control device includes a modulation unit that defines a predetermined region of the basic voltage space, which is the outputtable voltage vector region of the inverter circuit, as a moduloable region. This modulation unit switches between performing a first pulse width modulation that corrects the command voltage vector to be within the moduloable region when the command voltage vector is within the basic voltage space but outside the moduloable region, and a second pulse width modulation in which the ON / OFF states of two phase upper and lower arm switching elements are fixed and the ON / OFF state of the other phase upper and lower arm switching element is modulated. As a result, the first pulse width modulation corrects the voltage vector that is outside the moduloable region to be within the moduloable region before the modulation unit can perform modulation control, making it possible to achieve operation at a relatively high modulation rate while suppressing the excitation of common-mode noise.

[0021] On the other hand, when a high modulation ratio is required, the system switches to a second pulse width modulation that modulates the ON / OFF state of only one phase upper / lower arm switching element. This makes it possible to further increase the modulation ratio while maintaining a substantial common-mode noise suppression effect. As a result, it becomes possible to achieve operation over a wide range equivalent to that of general modulation methods while maintaining an excellent common-mode noise suppression effect, and is extremely effective, for example, when driving a motor as a load, as in the ninth invention.

[0022] In particular, the second invention, as in the third invention, is configured such that the modulation unit switches between pulse width modulation that outputs only odd voltage vectors in one control cycle and pulse width modulation that outputs only even voltage vectors in one control cycle, based on the electrical angular phase, when the command voltage vector is within the fundamental voltage space and outside the moduloable region, the command voltage vector is modified to fall within a moduloable region that superimposes the moduloable region for pulse width modulation that outputs only odd voltage vectors in one control cycle and the moduloable region for pulse width modulation that outputs only even voltage vectors in one control cycle. This makes it possible to increase the modulation ratio while achieving a high common-mode noise suppression effect with the first pulse width modulation.

[0023] Furthermore, as in the fourth invention, the modulation unit calculates a three-phase modulation command value for generating the phase voltage of each phase, and calculates a two-phase modulation command value that stops switching by turning on the lower arm switching element of the phase with the smallest three-phase modulation command value, or by turning on the upper arm switching element of the phase with the largest three-phase modulation command value, and switches between the first pulse width modulation and the second pulse width modulation according to the modulation rate and electrical angle phase of this two-phase modulation command value, thereby enabling switching between the first pulse width modulation and the second pulse width modulation using two-phase modulation.

[0024] In that case, as in the fifth invention, the first pulse width modulation is performed in an electrical angular phase where, depending on the modulation rate, the lower arm switching element is turned ON only for the phase in which the two-phase modulation command value is smallest, thereby stopping the switching, or the upper arm switching element is turned ON only for the phase in which the two-phase modulation command value is large, thereby stopping the switching.

[0025] On the other hand, as in the eighth invention, in the electrical angular phase where the lower arm switching element or the upper arm switching element of the two phases of the two-phase modulation command value is turned ON and switching is stopped, a second pulse width modulation is performed.

[0026] Furthermore, as in the sixth invention, in the first pulse width modulation, in the electrical angular phase where the lower arm switching element of the phase with the minimum two-phase modulation command value is turned ON and switching is stopped, pulse width modulation is performed that outputs only even voltage vectors in one control cycle. In the electrical angular phase where the upper arm switching element of the phase with the maximum two-phase modulation command value is turned ON and switching is stopped, pulse width modulation is performed that outputs only odd voltage vectors in one control cycle. By utilizing the link with two-phase modulation, it becomes possible to smoothly switch between pulse width modulation that outputs only even voltage vectors in one control cycle and pulse width modulation that outputs only odd voltage vectors in one control cycle.

[0027] Furthermore, in the first pulse width modulation as described in the seventh invention, when the command voltage vector is within the fundamental voltage space and outside the moduloable region, in the electrical angular phase where the lower arm switching element of the phase with the minimum two-phase modulation command value is turned ON and switching is stopped, the command voltage vector is modified to fit within the moduloable region of pulse width modulation that outputs only even voltage vectors in one control cycle. In the electrical angular phase where the upper arm switching element of the phase with the maximum two-phase modulation command value is turned ON and switching is stopped, the command voltage vector is modified to fit within the moduloable region of pulse width modulation that outputs only odd voltage vectors in one control cycle. This makes it possible to expand the modulation rate in the first pulse width modulation.

[0028] This is an electrical circuit diagram of a power converter according to one embodiment to which the present invention is applied. This figure shows an example of voltage and current when a motor is driven with two-phase modulation. This figure illustrates the linear output region of RSPWM with odd voltage vectors. This figure shows the PWM pattern of RSPWM with odd voltage vectors. This figure illustrates the linear output region of RSPWM with even voltage vectors. This figure shows the PWM pattern of RSPWM with even voltage vectors. This figure illustrates the linear output region of RSPWM with all voltage vectors. This figure shows the correspondence between RSPWM with all voltage vectors and each phase. This figure illustrates the common-mode voltage of RSPWM with all voltage vectors. This figure illustrates the case when a motor is driven with RSPWM with all voltage vectors. This figure shows an example of voltage and current when a motor is driven with RSPWM with all voltage vectors. This is an enlarged view of the 80° to 90° section of Figure 11. This figure shows the moduloable region of RSPWM with all voltage vectors. This figure shows the applicable range in voltage space for RSPWM using odd voltage vectors and RSPWM using even voltage vectors in RSPWM using the total voltage vector. This figure shows the command voltage vector correction method in voltage space in the first pulse width modulation. This figure explains the command voltage vector when an arbitrary voltage is output using the method in Figure 15. This figure shows the waveform in the case of Figure 16. This figure explains the command voltage vector when a motor is driven using the method in Figure 15. This figure shows the waveform in the case of Figure 18. This figure shows single-phase modulation (second pulse width modulation) in voltage space. This is a flowchart explaining the switching control between the first pulse width modulation and the second pulse width modulation by the line-to-line modulation calculation unit (modulation switching unit) that constitutes the modulation unit in Figure 1. This figure explains the waveform of the count value of each phase in the switching control between the first pulse width modulation and the second pulse width modulation. This figure shows the waveforms of two-phase modulation and RSPWM. This figure shows the relationship between two-phase modulation and RSPWM in electrical angular phase. This figure illustrates the command voltage vector when an arbitrary voltage is output by switching control between the first and second pulse width modulation. This figure illustrates the command voltage vector when a motor is driven by switching control between the first and second pulse width modulation. This figure shows the waveform in the case of Figure 26. This is an enlarged view of the 40° to 60° section of Figure 27.This diagram compares the noise levels of each modulation scheme.

[0029] Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

[0030] The power conversion device 1 in the embodiment to which the present invention is applied drives the motor 8 (load) of a so-called inverter-integrated electric compressor that constitutes the refrigerant circuit of a vehicle air conditioning system mounted on a vehicle such as an electric vehicle.

[0031] (1) In the circuit diagram 1 of the power converter 1, the power converter 1 of the embodiment includes a three-phase inverter circuit 28 and a control device 21. The inverter circuit 28 is a circuit that converts the DC voltage Vdc (voltage of the smoothing capacitor 32, which will be described later) of the DC power supply (vehicle battery: for example, 350V) 29 into a three-phase AC voltage and applies it to the motor 8. In this case, the motor 8 of the embodiment is an IPMSM (Internal Permanent Magnet Synchronous Motor).

[0032] In this embodiment, a normal mode choke coil 30 is connected to the upper arm power line (positive busbar) 10 of the DC power supply 29, and the aforementioned smoothing capacitor 32 is connected between the upper arm power line 10 and the lower arm power line (negative busbar) 15 of the DC power supply 29 downstream of the normal mode choke coil 30. The normal mode choke coil 30 and the smoothing capacitor 32 constitute an input LC filter.

[0033] The inverter circuit 28 has a U-phase half-bridge circuit 19U, a V-phase half-bridge circuit 19V, and a W-phase half-bridge circuit 19W. Each of the phase half-bridge circuits 19U to 19W has upper arm switching elements 18A to 18C and lower arm switching elements 18D to 18F, respectively. Furthermore, each of the switching elements 18A to 18F has a flywheel diode 31 connected in antiparallel. In this embodiment, each of the upper and lower arm switching elements 18A to 18F is composed of an insulated gate bipolar transistor (IGBT) with a MOS structure incorporated into its gate portion.

[0034] The collectors of the upper arm switching elements 18A to 18C of the inverter circuit 28 are connected to the upper arm power line (positive bus) 10 downstream of the smoothing capacitor 32. On the other hand, the emitters of the lower arm switching elements 18D to 18F of the inverter circuit 28 are connected to the lower arm power line (negative bus) 15 downstream of the smoothing capacitor 32.

[0035] In this case, the emitter of the upper arm switching element 18A of the U-phase half-bridge circuit 19U and the collector of the lower arm switching element 18D are connected in series, the emitter of the upper arm switching element 18B of the V-phase half-bridge circuit 19V and the collector of the lower arm switching element 18E are connected in series, and the emitter of the upper arm switching element 18C of the W-phase half-bridge circuit 19W and the collector of the lower arm switching element 18F are connected in series.

[0036] Furthermore, the connection point (U-phase voltage Vu) between the upper arm switching element 18A and the lower arm switching element 18D of the U-phase half-bridge circuit 19U is connected to the U-phase armature coil of the motor 8, the connection point (V-phase voltage Vv) between the upper arm switching element 18B and the lower arm switching element 18E of the V-phase half-bridge circuit 19V is connected to the V-phase armature coil of the motor 8, and the connection point (W-phase voltage Vw) between the upper arm switching element 18C and the lower arm switching element 18F of the W-phase half-bridge circuit 19W is connected to the W-phase armature coil of the motor 8.

[0037] (2) Control device 21 Next, the control device 21 is composed of a microcomputer having a processor. In this embodiment, the built-in vector control unit 25 derives the d-axis voltage command value Vd and the q-axis voltage command value Vq, and controls the ON / OFF state (switching) of each switching element 18A to 18F of the inverter circuit 28 based on these values. Specifically, it controls the gate voltage applied to the gates of each switching element 18A to 18F.

[0038] The control device 21 of the embodiment includes a vector control unit 25, a phase voltage command calculation unit 33, a line-to-line modulation calculation unit 34, a PWM signal generation unit 36, a gate driver 37, and current sensors 26A, 26B, and 26C composed of current transformers for measuring the motor currents (phase currents) of each phase flowing through the motor 8, namely the U-phase current Iu, the V-phase current Iv, and the W-phase current Iw. In the embodiment, the phase voltage command calculation unit 33 and the line-to-line modulation calculation unit 34 of the control device 21 constitute the modulation unit 35 in the present invention.

[0039] In the embodiment, each of the current sensors 26A, 26B, and 26C is connected to the line-to-line modulation calculation unit 34. The current sensor 26A measures the U-phase current Iu, the current sensor 26B measures the V-phase current Iv, and the current sensor 26C measures the W-phase current Iw. However, the U-phase current Iu may be measured by the current sensor 26A, the V-phase current Iv may be measured by the current sensor 26B, and the W-phase current Iw may be obtained by calculation from these.

[0040] In addition to measuring with the current sensors 26A to 26C as in the embodiment for the method of detecting the motor current of each phase, there are methods such as detecting the current value of the lower-arm power supply line 15 with a shunt resistor and the phase voltage command calculation unit 33 estimating from the current value and the operating state of the motor 8. Therefore, the method for detecting and estimating each phase current is not particularly limited.

[0041] (2-1) Vector control unit 25 The vector control unit 25 calculates the q-axis current command value Iqref by PI calculation from the difference between the speed command value ωrmref and the mechanical angular velocity ωrm of the motor 8. Then, the d-axis voltage command value Vd and the q-axis voltage command value Vq are calculated from the d-axis current command value Idref and the d-axis current Id, and the q-axis current command value Iqref and the q-axis current Iq, and output to the phase voltage command calculation unit 33.

[0042] Here, in order to rotate the motor 8 (IPMSM), it is necessary to generate torque τ. This torque τ is obtained by the following formula (I). Here, Id and Iq are the d-axis current and the q-axis current, Ld and Lq are the d-axis inductance and the q-axis inductance, kE is the generator constant of the motor 8, and P is the number of pole pairs.

[0043]

[0044] The d-axis and q-axis are two-dimensional axes, with the magnetic pole position of the motor 8 (IPMSM) being the d-axis and the coordinate system orthogonal to it being the q-axis. In this case, the output torque τ is proportional to the q-axis current Iq, so to simply control the torque τ, the q-axis current Iq is controlled. The d-axis voltage command value Vd and the q-axis voltage command value Vq, which are used to supply the d-axis current Id and q-axis current Iq to the motor 8, can be calculated using the following formula (II).

[0045]

[0046] Here, ωre is the electrical angular rotation speed of motor 8, p is the derivative term, and R is the phase resistance. From this equation (II) and the previously mentioned equation (I), if the d-axis current Id and q-axis current Iq to be supplied to motor 8, and the electrical angular rotation speed ωre of motor 8 are determined, then the d-axis voltage command value Vd and the q-axis voltage command value Vq to be applied to motor 8 can be determined.

[0047] As described above, the motor 8 (IPMSM) is driven by the half-bridge three-phase inverter circuit 28. If the voltages that the inverter circuit 28 applies to the three-phase motor 8 are the U-phase voltage Vu, V-phase voltage Vv, and W-phase voltage Vw as described above, then the d-axis voltage command value Vd, the q-axis voltage command value Vq, and the phase voltages Vu, Vv, and Vw of U, V, and W are related by the following formula (III).

[0048]

[0049] (2-2) Phase Voltage Command Calculation Unit 33 The phase voltage command calculation unit 33, which constitutes the modulation unit 35 in the present invention, uses the following formula (IV) to calculate the U phase voltage command value Vu based on the d-axis voltage command value Vd and the q-axis voltage command value Vq obtained from the vector control unit 25. * V-phase voltage command value Vv * W-phase voltage command value Vw * This calculates the following: that is, a dq-axis to three-phase conversion is performed. Note that the command voltage vector Vm and θm in equation (IV) are obtained from equation (V), respectively. Also, θ is the magnetic pole position relative to the U phase, and θm is the voltage phase difference with respect to the magnetic pole position. Furthermore, the phase used in the control here is denoted as the electrical angular phase θx, and is expressed as θx = θ + θm.

[0050]

[0051] (2-3) Line-to-line modulation calculation unit 34 The line-to-line modulation calculation unit 34, which is also part of the modulation unit 35, calculates and outputs the phase voltage command value Vu for each phase from the phase voltage command calculation unit 33. * , Vv * VW * From there, the following formula (VI) is used to calculate the PWM count values ​​Nu1 (U-phase PWM count value), Nv1 (V-phase PWM count value), and Nw1 (W-phase PWM count value) for each phase, which are normalized (corrected to 0 to 1) by the DC voltage Vdc. These PWM count values ​​Nu1, Nv1, and Nw1 are the three-phase modulation command values ​​in the present invention before line-to-line modulation.

[0052]

[0053] Furthermore, the line-to-line modulation calculation unit 34 performs line-to-line modulation using formula (VII) based on the PWM count values ​​Nu1, Nv1, and Nw1 of each phase. In formula (VII), Nu2, Nv2, and Nw2 are the PWM count values ​​of each phase after line-to-line modulation.

[0054]

[0055] In formula (VII), Nmod is the line-to-line modulation value for line-to-line modulation (two-phase modulation). When this line-to-line modulation value Nmod is 0 (Nmod = 0), the PWM count values ​​Nu2, Nv2, and Nw2 are not modulated, meaning they are the same as the PWM count values ​​Nu1, Nv1, and Nw1 before line-to-line modulation. On the other hand, when the line-to-line modulation value Nmod is set to the following formula (VIII), line-to-line modulation is performed.

[0056]

[0057] In formula (VII), min(Nu1, Nv1, Nw1) represents the minimum value of the phase among the PWM count values ​​Nu1, Nv1, and Nw1 (three-phase modulation command values). In formula (VII), this value Nmod1 is subtracted from all PWM count values ​​Nu1, Nv1, and Nw1 as the line-to-line modulation value Nmod. As a result, the minimum PWM count value of the phase becomes 0, and the lower arm switching element of that phase is fixed in the ON state. The upper and lower arm switching elements of the other two phases are then turned ON / OFF, so Nu2, Nv2, and Nw2 become the two-phase modulation command values. In this application, this is referred to as sub-two-phase modulation.

[0058] In formula (VII), max(Nu1, Nv1, Nw1) represents the maximum value of the phase among the PWM count values ​​Nu1, Nv1, and Nw1 (three-phase modulation command values ​​in this embodiment). In formula (VII), this value is subtracted from 1 to obtain Nmod2, which is the line-to-line modulation value Nmod and is added to all PWM count values ​​Nu1, Nv1, and Nw1. As a result, the maximum PWM count value of the phase becomes 1, and the upper arm switching element of that phase is fixed in the ON state. The upper and lower arm switching elements of the other two phases are then turned ON / OFF, so Nu2, Nv2, and Nw2 become the two-phase modulation command values. In this application, this is referred to as superimposed two-phase modulation.

[0059] In formula (VII), the colon (:) indicates that the conditions must be met. Therefore, when the absolute value of Nmod1 is less than the absolute value of Nmod2, Nmod1 is used for the line-to-line modulation value Nmod in formula (VII), and when the absolute value of Nmod1 is greater than or equal to the absolute value of Nmod2, Nmod2 is used for the line-to-line modulation value Nmod in formula (VII). In other words, superscript two-phase modulation and subscript two-phase modulation are switched and executed. In this application, this is referred to as superscript two-phase modulation.

[0060] Furthermore, in this embodiment, the line-to-line modulation calculation unit 34 has a modulation switching unit 40, and the switching control between the first pulse width modulation and the second pulse width modulation performed by this modulation switching unit 40 will be described in detail later. The first pulse width modulation and the second pulse width modulation will also be described in detail later.

[0061] (2-4) PWM signal generation unit 36 ​​The PWM signal generation unit 36 ​​receives the voltage vector and output time output by the line modulation calculation unit 34, and compares its magnitude with the carrier signal to generate and output PWM signals that serve as drive command signals for the U-phase half-bridge circuit 19U, V-phase half-bridge circuit 19V, and W-phase half-bridge circuit 19W of the inverter circuit 28.

[0062] The PWM signal generation unit 36 ​​adjusts the switching timing of each switching element 18A to 18F. Specifically, the PWM signal generation unit 36 ​​adds a dead time, and the switching timing of the upper and lower arm switching elements 18A to 18F is determined.

[0063] (2-5) Gate Driver 37 The gate driver 37 generates gate voltages for the switching elements 18A and 18D of the U-phase half-bridge circuit 19U, the gate voltages for the switching elements 18B and 18E of the V-phase half-bridge circuit 19V, and the gate voltages for the switching elements 18C and 18F of the W-phase half-bridge circuit 19W, based on the PWM signal output from the PWM signal generation unit 36.

[0064] Then, each switching element 18A to 18F of the inverter circuit 28 is driven ON / OFF based on the gate voltage output from the gate driver 37. That is, when the gate voltage is ON (a predetermined voltage value), the switching element operates ON, and when the gate voltage is OFF (zero), the switching element operates OFF. The gate driver 37 is a circuit for applying the gate voltage to the IGBTs based on the PWM signal when the switching elements 18A to 18F are the aforementioned IGBTs, and is composed of a photocoupler, logic IC, transistor, etc.

[0065] Then, the voltage at the connection point between the upper arm switching element 18A and the lower arm switching element 18D of the U-phase half-bridge circuit 19U is applied (output) to the U-phase armature coil of the motor 8 as the U-phase voltage Vu (phase voltage), the voltage at the connection point between the upper arm switching element 18B and the lower arm switching element 18E of the V-phase half-bridge circuit 19V is applied (output) to the V-phase armature coil of the motor 8 as the V-phase voltage Vv (phase voltage), and the voltage at the connection point between the upper arm switching element 18C and the lower arm switching element 18F of the W-phase half-bridge circuit 19W is applied (output) to the W-phase armature coil of the motor 8 as the W-phase voltage Vw (phase voltage).

[0066] In this example, with a two-level inverter, when the lower arm switches 18D to 18F are ON, 0V is applied to the motor 8, and when the upper arm switching elements 18A to 18C are ON, a DC voltage Vdc (input voltage) is applied to the motor 8. In other words, two levels of voltage are applied to the motor 8 as phase voltages.

[0067] The phase voltages Vun, Vvn, and Vwn, which are normalized to 0-1 as described later, are expressed as 0 or 1 by dividing the two-level voltages Vu, Vv, and Vw (0 or Vdc) by Vdc. Furthermore, while the common-mode voltage Vmid is generally expressed as (Vu + Vv + Vw) / 3, here it is expressed as Vmidn = (Vun + Vvn + Vwn) / 3 using the normalized phase voltages Vun, Vvn, and Vwn.

[0068] Furthermore, the modulation index kH is defined by the following formula (IX).

[0069]

[0070] Without line-to-line modulation (no line-to-line modulation is performed), the maximum output modulation index kHmax can be used up to 1.0. Furthermore, by using two-phase modulation, the maximum output modulation index kHmax can be used up to 2 ÷ √3. In the case of the first pulse width modulation described later, the maximum output modulation index kHmax can also be used up to approximately 2 ÷ √3. However, while overmodulation allows the maximum output modulation index kHmax to exceed 2 ÷ √3 even with two-phase modulation, this is not possible with the first pulse width modulation.

[0071] Furthermore, the voltages Vα and Vβ along the αβ axis can be calculated using the following formula (X). Here, Vu, Vv, and Vw are the voltages output by PWM, but here we use the average value of the phase voltages output during the carrier cycle for the calculation.

[0072]

[0073] Furthermore, when normalized phase voltages Vun, Vvn, and Vwn are used, Vαn and Vβn are given by the following formula (XI).

[0074]

[0075] (3) Operation of the Line-to-Line Modulation Calculation Unit 34 Next, an example of the operation of the line-to-line modulation calculation unit 34, which constitutes the modulation unit 35, will be described. (3-1) Common-Mode Noise Leakage First, with reference to Figure 2, the leakage of common-mode noise in a power converter that drives the motor of this type of electric compressor will be described. Figure 2 shows an example of voltage and current when the motor 8 is driven by two-phase modulation. The top row of Figure 2 shows the waveforms of the U-phase current Iu, V-phase current Iv, and W-phase current Iw. The second row from the top shows the normalized U-phase voltage Vun and the U-phase PWM count value Nu2 after line-to-line modulation, the third row from the top shows the normalized V-phase voltage Vvn and the V-phase PWM count value Nv2 after line-to-line modulation, and the second row from the bottom shows the normalized W-phase voltage Vwn and the W-phase PWM count value Nw2 after line-to-line modulation.

[0076] Furthermore, the bottom row shows the normalized common-mode voltage (neutral point potential of motor 8) Vmidn. Figure 2 shows the simulation results of motor control performed by current feedback, with the horizontal axis representing the electrical angular phase.

[0077] As is clear from Figure 2, the common-mode voltage Vmidn constantly fluctuates in accordance with the phase voltages Vun, Vvn, and Vwn of each phase of U, V, and W. Generally, the motor 8 and switching elements 18A to 18F have stray capacitance and are connected to the housing of the electric compressor with a small capacitance. Therefore, fluctuations in the common-mode voltage Vmidn cause common-mode current to leak from the housing, which becomes common-mode noise.

[0078] In this case, the common-mode voltage Vmidn is generated at the timing of changes in the phase voltages Vun, Vvn, and Vwn of each phase of the UVW system. The technique of suppressing fluctuations in the common-mode voltage Vmidn by synchronizing the switching timing of each phase of the UVW system is known as RSPWM (Remote State PWM).

[0079] (3-2) RSPWM Here, the U-phase voltage Vu, V-phase voltage Vv, and W-phase voltage Vw can be represented by eight fundamental voltage vectors V0 to V7. Of these, V1, V3, and V5 are odd voltage vectors, V2, V4, and V6 are even voltage vectors, and V0 and V7 are zero voltage vectors. The fundamental voltage space B enclosed by the fundamental voltage vectors V1 to V6 in vector control is a hexagon as shown in Figures 3 and 5. This fundamental voltage space B becomes the outputtable voltage vector region in the inverter circuit 28.

[0080] Furthermore, the maximum amplitude at which the voltage vector can complete a clean rotation (draw a circle) in this basic voltage space B is the linear output region. Theoretically, this linear output region is the inscribed circle of a hexagon, and when the aforementioned two-phase modulation is used, the inscribed circle of a hexagon can be output. As a result, the maximum possible modulation index kHmax is 2 ÷ √3, as previously mentioned.

[0081] RSPWM includes RSPWM using odd voltage vectors, RSPWM using even voltage vectors, and RSPWM using all voltage vectors, which switches between RSPWM using odd voltage vectors and RSPWM using even voltage vectors depending on the electrical angular phase.

[0082] (3-2-1) RSPWM with odd voltage vectors RSPWM with odd voltage vectors (odd RSPWM) is a pulse width modulation that outputs only the odd voltage vectors V1, V3, and V5 from the aforementioned basic voltage vectors during one control cycle, as shown in Figure 3. Its moduloable region is the triangle in Figure 3, and the linear output region (linear region in the figure) is the inscribed circle of the triangle. That is, the moduloable region is a part of the basic voltage space B (a predetermined region). Figure 4 shows the output patterns of the odd voltage vectors V1, V3, and V5 of RSPWM with odd voltage vectors.

[0083] As is clear from FIG. 4, according to this RS-PWM with odd voltage vectors, theoretically, it is possible to completely suppress the fluctuation of the common-mode voltage Vmid within the carrier period. However, since only the odd voltage vectors V1, V3, and V5 are output, the linear output region is limited, and the maximum achievable modulation ratio kHmax = 2 / 3.

[0084] (3-2-2) RS-PWM with even voltage vectors RS-PWM with even voltage vectors (even RS-PWM) is a pulse-width modulation that outputs only the even voltage vectors V2, V4, and V6 of the basic voltage vectors described above during one control period as shown in FIG. 5. Its modifiable region is a triangle in FIG. 5, and the linear output region (the linear region in the figure) is the inscribed circle of the triangle. FIG. 6 shows the output patterns of the even voltage vectors V2, V4, and V6 of RS-PWM with even voltage vectors.

[0085] As is clear from FIG. 6, also with this RS-PWM with even voltage vectors, theoretically, it is possible to completely suppress the fluctuation of the common-mode voltage Vmid within the carrier period. However, since only the even voltage vectors V2, V4, and V6 are output, in this case as well, the linear output region is limited, and the maximum achievable modulation ratio kHmax = 2 / 3.

[0086] (3-2-3) Operation of the modulation switching unit 40 Incidentally, the operation of the modulation switching unit 40 of the inter-phase modulation calculation unit 34 described above at this time will be explained. In the modulation switching unit 40, either the first pulse modulation described later based on RS-PWM or the second pulse-width modulation, which is a one-phase modulation in which two of the three-phase modulation command values Vu * , Vv * , Vw * cause the upper-arm switching elements 18A to 18C or the lower-arm switching elements 18D to 18F of two phases to be in the ON state and stop switching, is switched and executed.

[0087] Here, the modulation switching unit 40 calculates the PWM count values ​​Nu, Nv, and Nw for each phase of UVW, which are ultimately output to the PWM signal generation unit 36, by adding the line-to-line modulation sum value NmodRS, given by the following formula (XII), to the PWM count values ​​Nu2, Nv2, and Nw2 of each phase after line-to-line modulation.

[0088]

[0089] The line-to-line modulation sum NmodRS becomes the following formula (XIII) when correcting to RSPWM with even voltage vectors, and the following formula (XIV) when correcting to RSPWM with odd voltage vectors. Furthermore, in the case of single-phase modulation (second pulse width modulation), the following formula (XV) is used.

[0090]

[0091] In RSPWM using odd voltage vectors, the switching timing is adjusted as shown in Figure 4, and in RSPWM using even voltage vectors, the switching timing is adjusted as shown in Figure 6. In the case of single-phase modulation (second pulse width modulation), only the switching pulse width is specified, and the switching timing is arbitrary.

[0092] Here, we will explain the basis for the above formulas (XIII) and (XIV) concerning the line-to-line modulation sum NmodRS. The PWM count values ​​Nu, Nv, and Nw for each phase of UVW are between 0 and 1. Therefore, if all three phases of UVW are ON during one carrier cycle, the sum of the ON times of the three phases will be 3. That is, Nu + Nv + Vw = 3.

[0093] Next, considering RSPWM with even voltage vectors, as is clear from Figure 6, one of the three phases will always have an OFF time during one carrier period. That is, the sum of the OFF times is 1, and conversely, the sum of the ON times is 2, so we want to set the line-to-line modulation sum NmodRS so that Nu + Nv + Vw = 2. Therefore, (Nu2 + NmodRS) + (Nv2 + NmodRS) + (Nw2 + NmodRS) = 2, and rearranging the equation, we get NmodRS = (2 - (Nu2 + Nv2 + Nw2)) / 3, which is equation (XIII).

[0094] Similarly, in the case of RSPWM with odd voltage vectors, one of the three phases will always have an ON time during one carrier cycle. That is, the total ON time is 1. Then, calculating in the same way as in the case of RSPWM with even voltage vectors, we get equation (XIV) with NmodRS = (1 - (Nu2 + Nv2 + Nw2)) / 3.

[0095] (3-2-4) RSPWM with all voltage vectors Next, Figure 7 shows the linear output region of the RSPWM with all voltage vectors (the linear region in the figure), and Figure 8 shows the correspondence between the RSPWM with all voltage vectors and each phase. In Figure 7, odd RSPWM is the moduloable region of the RSPWM with odd voltage vectors, and even RSPWM is the moduloable region of the RSPWM with even voltage vectors.

[0096] As shown in Figure 8, one period of the electrical angle is divided into six phase regions (330° < θx ≤ 30°, 30° < θx ≤ 90°, 90° < θx ≤ 150°, 150° < θx ≤ 210°, 210° < θx ≤ 270°, 270° < θx ≤ 330°), and for each phase, RSPWM with odd voltage vectors and RSPWM with even voltage vectors are alternately switched. As a result, the linear output region by RSPWM is expanded compared to when each is performed individually (the innermost circle in Figure 7), as shown by the second innermost circle in Figure 7. Consequently, the maximum possible output modulation index kHmax = 4 / (3 * √3).

[0097] Figure 9 shows the phase voltage command values ​​Vu for each phase. * , Vv * VW * The common-mode voltage Vmid for RSPWM using all voltage vectors is shown. In this case, the common-mode voltage Vmid fluctuates when switching between RSPWM using odd voltage vectors and RSPWM using even voltage vectors.

[0098] Figures 10 and 11 show an example of driving motor 8 with RSPWM using the full voltage vector. Figure 10 plots the normalized output voltage on the αβ axis. Figure 11 shows the phase current and phase voltage waveforms of RSPWM using the full voltage vector. With RSPWM using the full voltage vector, the output can extend to the inscribed circles of two triangles, so this example shows the maximum modulation rate that can be output with RSPWM using the full voltage vector. Theoretically, the common mode voltage Vmidn occurs 12 times during two electrical angles, as shown in Figure 9. In Figure 11, there is one electrical angle rotation, so the fluctuation of the common mode voltage Vmidn should be 6 times, but more fluctuations occur. This is because the switching timing has not been synchronized due to the effect of dead time.

[0099] In this analysis, the motor 8 used was driven at 4800 rpm. Therefore, with RSPWM using the full voltage vector, it can only be driven from 0 to 4800 rpm. However, since the rotational speed changes when the motor 8 and DC voltage Vdc change, the comparison should be made only as a relative rotational speed.

[0100] Figure 12 is an enlarged view of the voltage waveform in Figure 11, specifically the 80° to 90° interval (excluding the current waveform). As mentioned earlier, although RSPWM is implemented using the entire voltage vector, cancellation fails due to the effects of dead time. In reality, the switching timing and speed change depending on the magnitude and direction of the current flowing through switching elements 18A to 18F. As a result, it is difficult to perfectly synchronize the switching, and fluctuations in the common-mode voltage Vmidn occur.

[0101] Furthermore, the characteristics of switching elements 18A to 18F are nonlinear, making it difficult to predict the current flowing around 0A. Therefore, it is difficult to correct the switching to be perfectly synchronized. As a result, in Figure 12, the switching synchronization between the V-phase and W-phase fails, and a common-mode voltage Vmidn is generated.

[0102] Thus, although it is difficult to completely suppress the common-mode voltage Vmidn, the number of occurrences of the common-mode voltage Vmidn is significantly reduced compared to the common-mode voltage Vmidn of two-phase modulation shown in Figure 2. As a result, the common-mode noise itself can also be significantly reduced.

[0103] Figure 13 shows the moduloable region Z3 of RSPWM using all voltage vectors. The moduloable region of RSPWM using all voltage vectors (hereinafter referred to as the "total voltage moduloable region") Z3 is a region that superimposes the moduloable region Z1 of RSPWM using odd voltage vectors and the moduloable region Z2 of RSPWM using even voltage vectors. In addition, there are a total of six regions X that are outside the range of the total voltage moduloable region Z3 (hereinafter referred to as the "non-modulation region"), surrounding the total voltage moduloable region Z3. These non-modulation regions X form an isosceles triangle.

[0104] Furthermore, circle Q3 in Figure 13 represents the linear output region when RSPWM using the full voltage vector is used alone. As mentioned above, circle Q3, which is the linear output region of RSPWM using the full voltage vector, is larger than circles Q1 and Q2, which are the linear output regions when RSPWM using odd voltage vectors or RSPWM using even voltage vectors is performed alone, but the modulation rate can only be used up to circle Q3.

[0105] A smaller available modulation rate means that the voltage that can be applied to the motor 8 is smaller, which means that it cannot be driven at high rotational speeds, i.e., the range of usable rotational speeds is narrower.

[0106] (3-2-5) First pulse width modulation The method described in Patent Document 4 above was developed to enable output beyond this linear output region (circle Q3) and utilize the entire range of the full voltage modulation region Z3 in Figure 13. In the present invention, this is referred to as the first pulse width modulation. Below, an overview of the command voltage vector modification operation performed in the first pulse width modulation of the present invention as described in Patent Document 4 will be explained.

[0107] Furthermore, Figure 14 shows the phase region odd to which RSPWM using odd voltage vectors is applied, and the phase region even to which RSPWM using even voltage vectors is applied, within the full voltage modulation region Z3 of RSPWM using all voltage vectors. Therefore, the full voltage modulation region Z3 is also a part of the fundamental voltage space B (a predetermined region).

[0108] In this first pulse width modulation, the line-to-line modulation calculation unit 34 determines whether the command voltage vector Vm is within the range of the full voltage modulation region Z3. If the command voltage vector Vm is within the range of the full voltage modulation region Z3, no modification of the command voltage vector Vm is necessary, so the command voltage vector Vm is set to = modified command voltage vector Vm'.

[0109] On the other hand, as shown in Figure 15, if the command voltage vector Vm is outside the range of the full voltage modulation region Z3, i.e., in the non-modulation region X, a command voltage vector is calculated that is close to the command voltage vector Vm (approximate range) but within the range of the full voltage modulation region Z3, and this is designated as the corrected command voltage vector Vm'.

[0110] Furthermore, the non-modulation region X refers to the region within the hexagonal fundamental voltage space B enclosed by the fundamental voltage vectors V1 to V6 in vector control, and outside the range of the fully modulated region Z3. The above approximate ranges include, for example, the range where the vector length N' of the modified command voltage vector Vm' is 0.3N ≤ N' ≤ 1.7N, 0.5N ≤ N' ≤ 1.5N, or 0.7N ≤ N' ≤ 1.3N, relative to the vector length N of the command voltage vector Vm.

[0111] Furthermore, the approximate ranges include, for example, the range in which the electrical angular phase θx' of the corrected command voltage vector Vm' is θx - 120° ≤ θx' ≤ θx + 120°, θx - 90° ≤ θx' ≤ θx + 90°, θx - 60° ≤ θx' ≤ θx + 60°, and θx - 45° ≤ θx' ≤ θx + 45°.

[0112] In the present invention, as described later, when the two-phase modulation command values ​​Nu2, Nv2, and Nw2 are subscript two-phase modulation, RSPWM is executed using even voltage vectors, and when they are superscript two-phase modulation, RSPWM is executed using odd voltage vectors. Therefore, the line-to-line modulation calculation unit 34, while maintaining the above approximation range, basically corrects the command voltage vector Vm to within the modulatorable region Z2 of RSPWM using even voltage vectors in the electrical angular phase of subscript two-phase modulation, and outputs a corrected command voltage vector Vm' in which the command voltage vector Vm is corrected to within the modulatorable region Z1 of RSPWM using odd voltage vectors in the electrical angular phase of superscript two-phase modulation.

[0113] Here, Figures 16 and 17 show the waveforms when no feedback control is performed and an arbitrary voltage is output. Therefore, unlike Figure 18 which will be described later, an arbitrary voltage command is continuously issued. With the first pulse width modulation, as shown in Figure 16, the modified command voltage vector Vm', which is a modified version of the command voltage vector Vm, takes the shape of a flower petal and is wider than the inscribed circle Q3 of the two triangles.

[0114] However, since the correction command voltage vector Vm' used is inside a triangle, the common-mode voltage Vmid can be canceled out by RSPWM. In this case, the applied voltage is no longer a circle (it's like a flower petal), so the motor current tends to be distorted, but by controlling it as in the first pulse width modulation, it becomes possible to drive the motor while minimizing the distortion of the motor current.

[0115] Figures 18 and 19 show an example of driving the motor 8 using feedback control with the first pulse width modulation. Although the phase currents Iu, Iv, and Iw are distorted, a high modulation rate can be output, and the common-mode voltage Vmidn is suppressed to the same level as RSPWM using the total voltage vector. Note that in Figure 18, when plotting along the sides of a triangle, it is a two-phase modulation state where the switching of one phase is stopped, and the remaining two phases are operating, synchronizing the rising and falling edges to cancel out fluctuations in the common-mode voltage Vmidn.

[0116] As shown in this figure, there is a limit to how far we can continue driving with RSPWM while distorting the phase current to some extent, as shown in Figure 18. If we try to increase the modulation rate any further, the current distortion will increase.

[0117] Furthermore, in Figure 18, there are cases where the voltage is output outside the triangle; this is due to the effect of dead time error, and the voltage command (corrected command voltage vector Vm') is designed so that the voltage is output inside the triangle. Note that the motor drive used in the analysis of each figure is conditional on 6400 rpm. Therefore, by using the first pulse width modulation, it is possible to drive from 0 to 6400 rpm, which significantly expands the driving range compared to RSPWM using the full voltage vector, but it still has the problem of having a narrower driving range than two-phase modulation, which can further expand the driving range by using overmodulation.

[0118] (3-3) Single-phase modulation as a second pulse width modulation Next, Figure 20 shows the voltage space in the case of single-phase modulation as a second pulse width modulation in the present invention. In single-phase modulation, the ON / OFF states of the upper and lower arm switching elements of two of the three phases are fixed, and the ON / OFF state of the upper and lower arm switching elements of the remaining one phase is modulated (PWM).

[0119] Here, Figure 20 shows the results when the upper arm switching element 18A of the U-phase half-bridge circuit 19U is fixed ON and the lower arm switching element 18D is fixed OFF, while the upper arm switching element 18C of the W-phase half-bridge circuit 19W is fixed OFF and the lower arm switching element 18F is fixed ON, and the upper and lower arm switching elements 18B and 18E of the V-phase half-bridge circuit 19V are turned ON / OFF to change the PWM pulse width.

[0120] In the upper part of Figure 20, the OFF interval is slightly longer with respect to the V-phase pulse width. Also, in the lower part of Figure 20, the ON interval is slightly longer with respect to the V-phase pulse width. Thus, when only the V-phase pulse is changed, the combined voltage vector moves along the sides of the outer hexagon (fundamental voltage space B).

[0121] In this way, with single-phase modulation where only one phase is operated by PWM, the motor applied voltage can be placed on the sides of the outer hexagon (fundamental voltage space B). In that case, the voltage that can be applied to the motor 8 is higher than the inscribed circle (linear output region Z3 of RSPWM due to the total voltage vector), but the voltage is distorted, so the current waveform of the motor 8 moves away from a sine wave. As a result, the efficiency of the motor 8 decreases.

[0122] On the other hand, since single-phase modulation involves PWM operation (switching) of only one phase, the switching losses of the inverter circuit 8 can be significantly reduced, resulting in improved inverter efficiency. In addition, because the number of PWM operations (switching) is reduced, the fluctuation of the common-mode voltage is reduced (twice per carrier cycle), and the resulting common-mode noise can also be reduced.

[0123] Furthermore, while the first pulse width modulation method described above (the method in Patent Document 4) ideally eliminates common-mode noise, in reality, as mentioned above, it is not possible to eliminate fluctuations in common-mode voltage due to the effects of dead time, and common-mode noise is generated. Therefore, the first pulse width modulation is a common-mode noise suppression method that is equivalent to or worse than the second pulse width modulation method (single-phase modulation). Consequently, it can be said that using the second pulse width modulation method (single-phase modulation) does not increase common-mode noise.

[0124] (3-4) Switching control between the first pulse width modulation and the second pulse width modulation In this invention, the first pulse width modulation and the second pulse width modulation are switched and executed depending on the modulation rate. An example of the switching control between the first pulse width modulation and the second pulse width modulation in this invention will be described below with reference to Figures 21 to 29. Figure 21 is a flowchart illustrating the operation of the line-to-line modulation calculation unit 34 and its modulation switching unit 40 that constitute the modulation unit 35, and Figure 22 shows the waveforms of the PWM count values ​​Nu, Nv, and Nw for each phase of UVW output from the line-to-line modulation calculation unit 34 to the PWM signal generation unit 36.

[0125] In step S1 of Figure 21, the line-to-line modulation calculation unit 34 calculates the two-phase modulation command values ​​Nu2, Nv2, and Nw2 for the upper and lower two-phase modulation described above. Next, in step S2, the modulation switching unit 40 of the line-to-line modulation calculation unit 34 determines whether the two-phase modulation command values ​​Nu2, Nv2, and Nw2 for two of the UVW phases are fixed to 1 or 0. That is, it determines whether the ON / OFF state of the upper and lower arm switching elements of the two phases is fixed.

[0126] If the result at this point is negative, the process proceeds to step S3, where it is determined whether the current electrical angular phase is the aforementioned subscript two-phase modulation. If it is the aforementioned subscript two-phase modulation, the process proceeds to step S4 to execute the aforementioned odd-numbered voltage vector RSPWM. If it is the aforementioned subscript two-phase modulation, the process proceeds to step S5 to execute the aforementioned even-numbered voltage vector RSPWM. Furthermore, the aforementioned first pulse width modulation is executed by correcting the aforementioned command voltage vector Vm (generating the corrected command voltage vector Vm').

[0127] Here, Figures 23 and 24 show a comparison of the two-phase modulation and RSPWM methods. As shown in Figures 23 and 24, the phase at which the upper and lower two-phase modulation are switched in upper and lower two-phase modulation is the same as the phase (electrical angular phase θx) at which the odd-voltage vector RSPWM and even-voltage vector RSPWM are switched in total power vector RSPWM.

[0128] By utilizing this characteristic, it is possible to calculate whether it is preferable to use RSPWM with odd voltage vectors or RSPWM with even voltage vectors, based on two-phase modulation.

[0129] Furthermore, the switching to single-phase modulation, which will be described later, can be performed by fixing the voltage vector of one phase to 1 or 0, and ensuring that the voltage vector of the other phase also exceeds the pulse width limit of 0 to 1. Therefore, control can be performed by first calculating the two-phase modulation command values ​​Nu2, Nv2, and Nw2 in advance, then confirming that the system is in a single-phase modulation state, thereby simplifying the calculations.

[0130] When the modulation rate is low, the first pulse width modulation is performed as described above. However, when the modulation rate increases, as shown in the phase region Y (around 30°) of Figure 22, for example, the count value Nu of the U phase becomes 1 and the count value Nw of the W phase becomes 0, resulting in a state where only the V phase is subjected to PWM. In other words, it becomes a single-phase modulation state.

[0131] In that case, the modulation switching unit 40 proceeds from step S2 to step S6 to perform a second pulse width modulation, i.e., single-phase modulation. By switching between the first pulse width modulation and the second pulse width modulation depending on the modulation rate, it becomes possible to output voltages not only inside the two triangles as shown in Figure 25, but also on the sides of the outer hexagon. The voltages on the sides of this hexagon represent a single-phase modulation state in which two phases of PWM among the PWM of each phase (UVW) are stopped. Note that Figure 25 also shows a state in which an arbitrary voltage command is continuously being issued.

[0132] By switching between the first pulse width modulation (RSPWM using the corrected total power vector) and the second pulse width modulation (single-phase modulation) as needed within one electrical angle period, i.e., according to the modulation rate and electrical angle phase, a high modulation rate can be output, and distortion of the motor current can be suppressed.

[0133] Figures 26 and 27 show an example of driving the motor 8 using the switching control between the first and second pulse width modulation described above. From each figure, it can be seen that the modulation rate can be increased and the current distortion is small. In addition, the number of fluctuations in the common-mode voltage Vmidn is about the same as that of RSPWM using the total voltage vector.

[0134] Looking at the command voltage vector in Figure 26, we see that the point where the command voltage vector on the side of the hexagon, which is in phase when single-phase modulation is being performed, is selected is different from that in Figure 25. This is because in Figure 26, the voltage command is issued while current control is performed by feedback control, so the phase of switching between the second pulse width modulation (single-phase modulation) and the first pulse width modulation is different from that in Figure 25.

[0135] The motor drive conditions used in this analysis were 7200 rpm. That is, by switching between the first and second pulse width modulation, it became possible to drive at speeds above 6400 rpm (an example of 7200 rpm is shown here). Furthermore, it is possible to drive at even higher rotational speeds; by increasing the ratio of the second pulse width modulation, it is possible to raise the speed to around 7600 rpm. This is comparable to two-phase modulation using overmodulation.

[0136] Figure 28 shows an enlarged waveform of the 40° to 60° section of Figure 27 (current waveform omitted). Figure 28 shows the region of single-phase modulation (second pulse width modulation) where only the W phase is switched. In the 58° to 60° region, only the V phase has stopped switching, and the U and W phases are operating to cancel each other's switching (first pulse width modulation). However, it can be seen that the cancellation operation fails, and a common-mode voltage Vmidn is generated.

[0137] Thus, when the generation of common-mode voltage is canceled by the first pulse width modulation, cancellation failures occur approximately once per carrier cycle. On the other hand, in the case of single-phase modulation (second pulse width modulation), only one phase switches, and in that case, a common-mode voltage Vmidn is generated.

[0138] Thus, when a second pulse width modulation (single-phase modulation) is performed, and when the cancellation of common-mode voltages fails with the first pulse width modulation, the number of common-mode voltages generated is roughly the same. Therefore, the amount of common-mode noise generated by the second pulse width modulation and the first pulse width modulation is roughly equal.

[0139] Figure 29 shows the FFT results for the common-mode voltage Vmid = (Vu + Vv + Vw) / 3. In Figure 29, the horizontal axis represents frequency, and the vertical axis represents the level of the common-mode voltage, i.e., the level of common-mode noise. Therefore, the smaller the value on the vertical axis in Figure 29, the lower the amount of noise.

[0140] In the 0.3 MHz to 3 MHz range shown in Figure 29, the noise level of the first pulse width modulation alone (indicated as RSPWM) and the switching control of the first and second pulse width modulations according to the present invention (indicated as in the present invention) are approximately the same, with only the two-phase modulation exhibiting a significantly higher noise level.

[0141] For example, at 1 MHz, the common-mode voltage (common-mode noise) is approximately 125 dBuV in the case of two-phase modulation, while it is approximately 122 dBuV in the case of the first pulse-width modulation alone and the switching control of the first and second pulse-width modulations according to the present invention. From this, it can be said that by performing the first pulse-width modulation during low-speed driving and switching control of the first and second pulse-width modulations when the rotational speed increases, a wide operating range can be achieved, and the common-mode voltage (common-mode noise) can be reduced across the entire operating range.

[0142] As described in detail above, in the present invention, the modulation unit 35 switches between performing a first pulse width modulation, which corrects the command voltage vector to be within the modulatory region, and a second pulse width modulation of single-phase modulation, when the command voltage vector is within the basic voltage space B and outside the modulatory region. As a result, the first pulse width modulation corrects the voltage vector that is outside the modulatory region to be within the modulatory region, and then the modulation unit 35 can perform modulation control. This makes it possible to achieve operation at a relatively high modulation rate while suppressing the excitation of common-mode noise.

[0143] On the other hand, when a high modulation ratio is required, the system switches to a second pulse width modulation, making it possible to further increase the modulation ratio while maintaining the effective common-mode noise suppression effect. This makes it possible to achieve operation over a wide range equivalent to that of general modulation methods while maintaining an excellent common-mode noise suppression effect, and is extremely effective, for example, when driving a motor 8 as a load, as in the embodiment.

[0144] In particular, as in the embodiment, the modulation unit 35 switches between executing RSPWM using odd voltage vectors and RSPWM using even voltage vectors based on electrical angular phase in the first pulse width modulation. Furthermore, when the command voltage vector is within the basic voltage space B and outside the moduloable region, the command voltage vector is modified to fall within a moduloable region Z3, which is a superposition of the moduloable region Z1 for RSPWM using odd voltage vectors and the moduloable region Z2 for RSPWM using even voltage vectors. This makes it possible to increase the modulation rate while achieving a high common-mode noise suppression effect with the first pulse width modulation.

[0145] Furthermore, as in the embodiment, the modulation unit 35 calculates a three-phase modulation command value for generating the phase voltage of each phase, and calculates a two-phase modulation command value to stop switching by turning on the lower arm switching element of the phase with the smallest three-phase modulation command value, or turning on the upper arm switching element of the phase with the largest three-phase modulation command value, and switches between the first pulse width modulation and the second pulse width modulation according to the modulation rate and electrical angle phase of this two-phase modulation command value, thereby enabling switching between the first pulse width modulation and the second pulse width modulation using two-phase modulation.

[0146] Furthermore, as in the embodiment, in the first pulse width modulation, by executing RSPWM with even voltage vectors in the electrical angular phase of subscripted two-phase modulation and RSPWM with odd voltage vectors in the electrical angular phase of superscripted two-phase modulation, it becomes possible to smoothly switch between RSPWM with even voltage vectors and RSPWM with odd voltage vectors by utilizing the link with two-phase modulation.

[0147] Furthermore, in the first pulse width modulation as shown in the embodiment, when the command voltage vector is within the fundamental voltage space B and outside the moduloable region, the command voltage vector is modified to fall within the moduloable region Z2 of RSPWM with even voltage vectors in the electrical angular phase of subscript two-phase modulation, and the command voltage vector is modified to fall within the moduloable region Z1 of RSPWM with odd voltage vectors in the electrical angular phase of superscript two-phase modulation, thereby enabling an expansion of the modulation rate in the first pulse width modulation.

[0148] In this embodiment, the driving of the motor (load) of an electric compressor was used as an example, but the method is not limited to this and is also effective when driving motors other than the motor of an electric compressor.

[0149] 1 Power conversion device 8 Motor 18A-18F Upper and lower arm switching element 21 Control device 25 Vector control unit 28 Inverter circuit 29 DC power supply 33 Phase voltage command calculation unit 34 Line-to-line modulation calculation unit 35 Modulation unit 36 ​​PWM signal generation unit 37 Gate driver 40 Modulation switching unit

Claims

1. A power conversion device for converting a DC voltage to an AC voltage, comprising: an inverter circuit that applies phase voltages at the connection points of the upper and lower arm switching elements of each phase to a load; and a control device that controls the switching of each switching element, wherein the control device comprises a modulation unit that defines a predetermined region of the basic voltage space, which is the outputtable voltage vector region of the inverter circuit, as a moduloable region, and the modulation unit is characterized by switching between and executing a first pulse width modulation that corrects the command voltage vector to the moduloable region when the command voltage vector is within the basic voltage space and outside the moduloable region, and a second pulse width modulation that fixes the ON / OFF state of the upper and lower arm switching elements of two phases and modulates the ON / OFF state of the upper and lower arm switching element of the other phase.

2. The power conversion device according to claim 1, characterized in that the modulation unit switches between and performs pulse width modulation that outputs only odd voltage vectors in one control cycle and pulse width modulation that outputs only even voltage vectors in one control cycle, based on the electrical angular phase, in the first pulse width modulation.

3. The power converter according to claim 2, wherein, in the first pulse width modulation, when the command voltage vector is within the basic voltage space and outside the moduloable region, the modulation unit modulates the command voltage vector to a moduloable region obtained by superimposing a moduloable region for pulse width modulation that outputs only odd voltage vectors in one control cycle and a moduloable region for pulse width modulation that outputs only even voltage vectors in one control cycle.

4. The power conversion device according to claim 1, wherein the modulation unit calculates a three-phase modulation command value for generating the phase voltage of each phase, and calculates a two-phase modulation command value that stops switching by turning on the lower arm switching element of the phase in which the three-phase modulation command value is smallest, or by turning on the upper arm switching element of the phase in which the three-phase modulation command value is largest, and switches between the first pulse width modulation and the second pulse width modulation according to the modulation rate and electrical angular phase of the two-phase modulation command value.

5. The power conversion device according to claim 4, characterized in that the modulation unit performs the first pulse width modulation in an electrical angular phase where, depending on the modulation rate, it stops switching by turning on the lower arm switching element only for the phase in which the two-phase modulation command value is smallest, or stops switching by turning on the upper arm switching element only for the phase in which the two-phase modulation command value is largest.

6. The power conversion device according to claim 5, characterized in that, in the first pulse width modulation, the modulation unit performs pulse width modulation that outputs only even voltage vectors during one control cycle in the electrical angular phase in which the lower arm switching element of the phase with the minimum two-phase modulation command value is turned ON and switching is stopped, and performs pulse width modulation that outputs only odd voltage vectors during one control cycle in the electrical angular phase in which the upper arm switching element of the phase with the maximum two-phase modulation command value is turned ON and switching is stopped.

7. The power converter according to claim 6, wherein, in the first pulse width modulation, when the command voltage vector is within the basic voltage space and outside the modulatorable region, the modulation unit corrects the command voltage vector to within the modulatorable region of pulse width modulation that outputs only even voltage vectors during one control cycle in the electrical angular phase in which the lower arm switching element of the phase with the minimum two-phase modulation command value is turned ON and switching is stopped, and the modulation unit corrects the command voltage vector to within the modulatorable region of pulse width modulation that outputs only odd voltage vectors during one control cycle in the electrical angular phase in which the upper arm switching element of the phase with the maximum two-phase modulation command value is turned ON and switching is stopped.

8. The power conversion device according to claim 4, characterized in that the modulation unit performs the second pulse width modulation at an electrical angular phase in which the lower arm switching element or the upper arm switching element of two phases of the two-phase modulation command value is turned ON and switching is stopped, depending on the modulation rate.

9. The power conversion device according to any one of claims 1 to 8, characterized in that the inverter circuit drives the motor by applying the phase voltage at the connection point of the upper and lower arm switching elements of each phase.

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