Integrated circuit, semiconductor device

Abstract

The present application provides an integrated circuit for switching a switching element while suppressing noise. The integrated circuit is an integrated circuit for switching a second switching element of a switching circuit including a first switching element on a power supply side and the second switching element on a ground side connected in series, a first backflow diode connected in parallel with the first switching element, and a second backflow diode connected in parallel with the second switching element, wherein the integrated circuit includes a detection circuit that detects a load current flowing through a load of the switching circuit, and a drive circuit that controls the size of a current for charging a gate capacitance of the second switching element according to the size of the load current when a drive signal is one logic level, and causes the second switching element to be turned off when the drive signal is another logic level.

Description

TECHNICAL FIELD

[0001] The present application relates to an integrated circuit and a semiconductor device. BACKGROUND

[0002] In a power conversion circuit such as an inverter circuit, a desired voltage is generated by switching a switching element provided in an upper arm on the power supply side and a switching element provided in a lower arm on the ground side (for example, Patent Literature 1).

[0003] PRIOR ART DOCUMENTS

[0004] PATENT LITERATURE

[0005] Patent Literature 1: Japanese Patent Application Laid-Open No. 2013-146008 SUMMARY

[0006] PROBLEMS TO BE SOLVED BY THE INVENTION

[0007] However, if the switching element of the lower arm is turned on, a reverse bias voltage is generated in the freewheeling diode of the switching element of the upper arm. Moreover, in general, the smaller the current flowing through the freewheeling diode of the upper arm, the greater the rate of change of the reverse bias voltage. As a result, if the current flowing through the freewheeling diode of the upper arm becomes small, a large noise is generated when the switching element of the lower arm is turned on.

[0008] The present application has been achieved in view of the above-described problems, and aims to provide an integrated circuit in which a switching element is switched while suppressing a noise.

[0009] SOLUTION TO PROBLEM

[0010] The main application for solving the above-described problems is an integrated circuit that is an integrated circuit that switches a second switching element of a switching circuit that includes a first switching element on the power supply side and the second switching element on the ground side connected in series, a first freewheeling diode connected in parallel to the first switching element, and a second freewheeling diode connected in parallel to the second switching element, wherein the integrated circuit includes a detection circuit that detects a load current flowing through a load of the switching circuit, and a drive circuit that controls the size of a current that charges a gate capacitance of the second switching element according to the size of the load current when a drive signal is one logic level, and turns off the second switching element when the drive signal is another logic level.

[0011] EFFECT OF THE INVENTION

[0012] According to the present application, it is possible to provide an integrated circuit in which a switching element is switched while suppressing a noise. BRIEF DESCRIPTION OF DRAWINGS

[0013] Figure 1 is a graph showing one example of the power module 10.

[0014] Figure 2 is a graph showing one example of the waveforms of the voltage VR and the current IF.

[0015] Figure 3 is a graph showing one example of the waveforms of the voltage VR and the current IF.

[0016] Figure 4 is a graph showing one example of the LVIC 25.

[0017] Figure 5 is a graph showing the relationship between the inductor current IL and the rate of change of the voltage VR.

[0018] Figure 6 is a graph showing one example of the LVIC 22.

[0019] Figure 7 is a graph showing one example of the detection circuit 51.

[0020] Figure 8 is a graph for explaining the operation of the detection circuit 51.

[0021] Figure 9 is a graph showing the main waveforms of the power module 10 when the IGBT 31 is turned on.

[0022] Figure 10 is a graph showing the relationship between the inductor current IL and the rate of change of the voltage VR.

[0023] Figure 11 is a graph showing one example of the detection circuit 51 and the setting circuit 70b.

[0024] Figure 12 is a graph showing one example of the power module 200.

[0025] Figure 13 is a graph showing one example of the power module 210. DETAILED DESCRIPTION

[0026] According to the description and the drawings of this specification, at least the following matters become clear.

[0027] === This Embodiment ===

[0028] Outline of Power Module 10

[0029] Figure 1Fig. 1 is a diagram showing the structure of a power module 10 according to an embodiment of the present application. The power module 10 includes power semiconductors for power conversion and a drive circuit, such as a semiconductor device that drives an inverter 11, and includes a half-bridge circuit 20, an HVIC 21, an LVIC 22, terminals IN1, IN2, COM, P, U, and N.

[0030] The half-bridge circuit 20 is configured to include IGBTs (Insulated Gate Bipolar Transistors) 30 and 31 and FWDs (Free Wheeling Diodes) 32 and 33. In addition, the half-bridge circuit 20 corresponds to a "switching circuit".

[0031] The IGBT 30 is a switching element provided in an upper arm on the power supply side, and the IGBT 31 is a switching element provided in a lower arm on the ground side in series with the IGBT 30. In addition, in the present embodiment, the gate voltage of the IGBT 30 is set to "voltage Vgl", and the gate voltage of the IGBT 31 is set to "voltage Vg2". The IGBT 30 corresponds to a "first switching element", and the IGBT 31 corresponds to a "second switching element".

[0032] The FWD 32 is connected in parallel with the IGBT 30 and is a free-wheeling diode that returns the energy of the inductor 11. The cathode of the FWD 32 is connected to the collector electrode of the IGBT 30, and the anode is connected to the emitter electrode of the IGBT 30.

[0033] The FWD 33 is a free-wheeling diode connected in parallel with the IGBT 31, and the cathode is connected to the collector electrode of the IGBT 31, and the anode is connected to the emitter electrode of the IGBT 31. In addition, the FWD 32 corresponds to a "first free-wheeling diode", and the FWD 33 corresponds to a "second free-wheeling diode".

[0034] A power supply voltage Vdc is applied to the terminal P, and the inductor 11 is provided as a load between the terminal P and the terminal U. Therefore, the inductor 11 and the half-bridge circuit 20 constitute a chopper circuit. In addition, the inductor 11 corresponds to a "load of a switching circuit".

[0035] A capacitor 12 for stabilizing the power supply voltage Vdc is connected between the terminal P and the terminal N, and a resistor 13 for detecting the inductor current IL of the inductor 11 is connected between the terminal N and the ground. In addition, in the present embodiment, the voltage generated in the resistor 13 is set to "voltage Vs", and the inductor current IL corresponds to a "load current".

[0036] Here, a load is not connected between the terminal U and the terminal N, but it is assumed that the FWD 33 operates as a free-wheeling diode in the case where an inductor is connected between the terminal U and the terminal N.

[0037] HVIC 21 (High Voltage Integrated Circuit: high voltage integrated circuit) is an integrated circuit for switching the IGBT 30 of the upper arm. The HVIC 21 switches the IGBT 30, for example, based on a drive signal Vdr1 input from a microcomputer (not shown) via a terminal IN1.

[0038] LVIC 22 (Low Voltage Integrated Circuit: low voltage integrated circuit) is an integrated circuit for switching the IGBT 31 of the lower arm. The LVIC 22 switches the IGBT 31, for example, based on a drive signal Vdr2 input from a microcomputer (not shown) via a terminal IN2.

[0039] The terminal COM is a terminal to which a ground voltage is applied, for example, connected to a case (not shown) of the power module 10 or the like.

[0040] <<Waveform of reverse bias voltage of FWD 32>>

[0041] Here, the change of the reverse bias voltage of the FWD 32 is described with reference to Figure 2 , Figure 3 . In addition, in the present embodiment, the voltage of the cathode electrode with the anode electrode of the FWD 32 of the upper arm as a reference, that is, the voltage after subtracting the voltage of the terminal U from the voltage of the terminal P is referred to as "voltage VR". Therefore, when the voltage VR becomes a voltage "positive", the reverse bias voltage is applied to the FWD 32.

[0042] Here, the current flowing in the FWD 32 is set to "current IF", and the direction of the current flowing in the forward direction of the FWD 32 is set to the "positive" direction. In addition, the power supply voltage Vdc is, for example, "300 V", and the description is made when the voltage of the terminal U is "0 V" and the voltage of the terminal P is "300 V". Figure 2 , Figure 3 . The IGBT 31 is driven by the LVIC 25 shown in Figure 4 instead of the LVIC 22 of Figure 1 .

[0043] The LVIC 25 is a drive circuit of a voltage drive type in which "0 V" and "15 V" are applied to the gate electrode of the IGBT 31, and is configured to include switches SW1 and SW2. In addition, the switch SW1 is configured by a PMOS transistor, for example, and the switch SW2 is configured by an NMOS transistor, for example.

[0044] Further, for example, when the drive signal Vdr2 is at a high level (hereinafter referred to as "H level"), the switch SW1 is turned on, and the switch SW2 is turned off. As a result, a prescribed voltage Vcc (for example, "15 V") is applied to the gate electrode of the IGBT 31 via the on-resistance of the switch SW1.

[0045] On the other hand, when the drive signal Vdr2 is at a low level (hereinafter referred to as "L level"), the switch SW1 is turned off, and the switch SW2 is turned on. As a result, a ground voltage (for example, "0 V") is applied to the gate electrode of the IGBT 31 via the on-resistance of the switch SW1.

[0046] Figure 2 is a graph showing one example of the waveform of the voltage VR when the current IF of the FWD 32 is small (for example, 2 A). Also, here, the IGBT 30 of the upper arm is always "turned off", and the IGBT 31 is "turned on" at time tl.

[0047] First, at time t0 to time tl, both the IGBT 30 and the IGBT 31 are "turned off", and thus the inductor current IL that flowed before time t0 continues to flow via the FWD 32. Figure 1 In the middle, although not particularly shown, for example, the FWD 32 and the wiring have a resistance component. Thus, the current IF gradually decreases as time approaches tl.

[0048] Further, for example, when the IGBT 31 is "turned on" at time tl, the current IF becomes substantially zero after passing through the reverse recovery time of the FWD 32. At this time, the voltage VR substantially rises to "300 V" after "time Ta" from time tl. As a result, a reverse bias voltage of "300 V" is applied to the FWD 32 for "time Ta".

[0049] Figure 3 is a graph showing one example of the waveform of the voltage VR when the current IF of the FWD 32 is large (for example, 28 A).

[0050] Here, as with Figure 2 Also, the IGBT 30 is always "turned off", and the IGBT 31 is "turned on" at time tl l. Also, Figure 2 As with Figure 3 the horizontal axis is the same, and thus the period from time t0 to t2 is equal to the period from time tlO to tl2, for example, 5 μs.

[0051] First, at time tlO to time tl l, both the IGBT 30 and the IGBT 31 are "turned off", and thus the inductor current IL that flowed before time tlO flows via the FWD 32 and gradually decreases as time passes.

[0052] Furthermore, for example, at time t11, when IGBT31 becomes "on", the current IF becomes essentially zero after the reverse recovery time of FWD32. At this time, the voltage VR rises from time t11 to approximately "300V" after time "Tb". As a result, a reverse bias voltage of "300V" is applied to FWD32 at "time Tb".

[0053] Here, if we take Figure 2 The rise time of voltage VR, i.e., "time Ta", is related to... Figure 3 Comparing the rise time (time Tb) of the voltage VR, the rise time Tb is longer. Therefore, if the inductor current IL decreases, the rise time of the voltage VR becomes shorter; conversely, if the inductor current IL increases, the rise time of the voltage VR becomes longer. This phenomenon is because, for example, when the inductor current IL is smaller and there are fewer charge carriers in the FWD32, the depletion layer diffuses faster; while when the inductor current IL is larger and there are more charge carriers in the FWD32, the depletion layer diffuses more slowly.

[0054] Figure 5 This is a graph showing the relationship between the inductor current IL and the rate of change of voltage VR during the rise when using LVIC25. Furthermore, the "rate of change of voltage VR" represents the proportion of the rise time of voltage VR, which is obtained, for example, by dividing the value of voltage VR during its rise ("300V") by the time it takes for voltage VR to change from "0V" to "300V".

[0055] like Figure 5 As shown, it can be seen that as the inductor current IL increases, the rate of increase of the voltage VR tends to decrease. Therefore, as... Figure 4 As shown, when using a typical voltage-driven LVIC25, the smaller the inductor current IL, the greater the noise generated through inductor 11 and capacitor 12 when IGBT31 is turned on.

[0056] Figure 1 To suppress this noise, the LVIC22 shown keeps the IGBT31 on for a longer period of time when the inductor current IL is small.

[0057] Therefore, even when the inductor current IL is small and the voltage VR rises rapidly, the sharp change in voltage VR can be suppressed by extending the conduction time of IGBT31. Furthermore, when the inductor current IL is large and the voltage VR rise time is long, shortening the conduction time of IGBT31 can prevent the voltage VR rise time from becoming excessively long.

[0058] <<<The Structure of LVIC22>>>

[0059] Figure 6is a diagram showing one example of the structure of the LVIC 22. The LVIC 22 is an integrated circuit that turns the IGBT 31 on and off based on the drive signal Vdr2 while preventing the voltage VR that prevents the reverse bias of the FWD 32 from becoming steep, and is configured by a filter 50, a detection circuit 51, and a drive circuit 52.

[0060] The filter 50 is a circuit that suppresses noise of the voltage Vs corresponding to the inductor current IL and outputs as the voltage Va. For example, the filter 50 is a low-pass filter including a resistor and a capacitor (not shown). Thus, the filter 50 suppresses noise of a frequency higher than the switching frequency of the IGBT 31 among the noise included in the voltage Vs.

[0061] The detection circuit 51 is a circuit that acquires the voltage Va output from the filter 50 based on the drive signal Vdr2. The voltage Va is the voltage Vs from which the noise is removed, and thus the detection circuit 51 detects the inductor current IL as the voltage. Details of the detection circuit 51 will be described later.

[0062] The drive circuit 52 turns the IGBT 31 on during a period corresponding to the detection result of the detection circuit 51 when the drive signal Vdr2 is at the "H" level, and turns the IGBT 31 off when the drive signal Vdr2 is at the "L" level. In addition, the "H" level corresponds to "one logic level", and the "L" level corresponds to "another logic level".

[0063] <<Details of the Detection Circuit 51>>

[0064] Figure 7 is a diagram showing one example of the structure of the detection circuit 51. The detection circuit 51 is a sample-and-hold circuit that samples and holds the voltage Va based on the drive signal Vdr2, and is configured by a delay circuit 60, an edge detection circuit 61, a switch 62, a capacitor 63, and a unity-gain buffer 64.

[0065] The delay circuit 60 and the edge detection circuit 61 are circuits that generate the timing of sampling the voltage Va. Specifically, the delay circuit 60 delays the drive signal Vdr2 by a "predetermined time Tx" and outputs as a signal Vd. In addition, the "predetermined time Tx" delayed in the delay circuit 60 is, for example, a time from when the drive signal Vdr2 is to turn the IGBT 31 off and become the "L" level to when the IGBT 31 actually becomes off. Details will be described later, but by sampling the current of the IGBT 31 at such timing, the detection circuit 51 can acquire the maximum value of the inductor current IL.

[0066] The edge detection circuit 61 detects the falling edge of the signal Vd and outputs a pulse signal Vp to the switch 62.

[0067] One end of the switch 62 is applied with the voltage Va, and is "turned on" only during the period when the pulse signal Vp is input. As a result, the voltage of the capacitor 63 becomes the voltage Va when the switch 62 is "turned on".

[0068] The unity gain buffer is a buffer circuit which outputs the voltage of the capacitor 63 as the voltage Vb.

[0069] Figure 8 is a diagram for explaining the operation of the detection circuit 51. Also, hereinafter, in the present embodiment, the noise component of the voltage Vs is omitted for convenience of explanation, and thus the voltage Vs is equal to the voltage Va.

[0070] When the drive signal Vdr2 becomes the "H" level at the time t30, the IGBT 31 is "turned on" by the drive circuit 52, for example, at the time t31. As a result, the inductor current IL flowing through the inductor 11 also gradually increases, and thus the voltage Va (= Vs) rises after the time t31.

[0071] Also, if the drive signal Vdr2 becomes the "L" level at the time t32, the signal Vd also becomes the "L" level from the time t32 to the time t33 after the delay of the "predetermined time Tx". As a result, the pulse signal Vp is output, and thus the switch 62 is "turned on", and the voltage Va is sampled.

[0072] Here, the time t33 is the timing when the IGBT 31 is turned off, and thus the current value of the inductor current IL is the largest. Thus, the voltage value representing the peak current of the IGBT 31 when the IGBT 31 is turned on is held in the capacitor 63. Also, the operation from the time t30 to the time t33 is repeated from the time t34 to the time t37 after the time t33.

[0073] In this way, the detection circuit 51 detects and holds the peak value of the current flowing in the IGBT 31 each time the IGBT 31 is turned on. Also, the current flowing in the IGBT 31 is the same as the inductor current IL when the IGBT 31 is turned on. Thus, the detection circuit 51 detects the peak value of the inductor current IL of the inductor 11 as a load each time the IGBT 31 is turned on.

[0074] <<Details of the drive circuit 52>>

[0075] The drive circuit 52 is a circuit that charges the gate capacitance of the IGBT 31 with a "source current Is (described later)" corresponding to the magnitude of the inductor current IL when the IGBT 31 is turned on, and applies a prescribed voltage to the gate electrode of the IGBT 31 when the IGBT 31 is turned off. That is, the drive circuit 52 current-drives the IGBT 31 when the IGBT 31 is turned on, and voltage-drives the IGBT 31 when the IGBT 31 is turned off. In addition, the gate electrode of the IGBT 31 corresponds to the "control electrode", and the gate capacitance of the IGBT 31 corresponds to the "gate capacitance of the second switching element".

[0076] As shown in FIG. 2, the drive circuit 52 includes a setting circuit 70a, a current generation circuit 71, a switch 72, an NMOS transistor 73, and a switching circuit 74. Figure 6

[0077] The setting circuit 70a is a circuit that sets the current value of the source current Is generated by the current generation circuit 71 (described later) based on the detection result of the detection circuit 51. Specifically, the setting circuit 70a outputs setting data SET that increases the source current Is as the voltage Vb increases to the current generation circuit 71. In addition, the setting data SET is, for example, 3-bit data. Here, although the setting circuit 70a is used, the setting circuit 70b described later can be used instead of the setting circuit 70a.

[0078] The current generation circuit 71 is a circuit that generates the source current Is based on the setting data SET, and includes a current source 100, PNP transistors 101 to 104, switches 110 to 112, and resistors 120 to 122.

[0079] The current source 100 generates a prescribed sink current, and the PNP transistor 101 and the PNP transistors 102 to 104 constitute a current mirror circuit.

[0080] The switches 110 to 112 are a switch group that turns on and off according to the setting data SET, and the resistors 120 to 122 are respectively provided between the switches 110 to 112 and the switch 72. Also, in the present embodiment, the current flowing from the PNP transistors 102 to 104 becomes the "source current Is".

[0081] ​When the inductor current IL is small, for example, "1 A", the voltage Vb is also small, and therefore, for example, only the switch 110 is on among the switches 110 to 112. Then, when the inductor current IL increases from "1 A" to, for example, "5 A", the voltage Vb becomes large. As a result, for example, the switches 110 and 111 are on among the switches 110 to 112. In this way, the current generation circuit 71 generates the source current Is that becomes larger as the inductor current IL increases, or becomes smaller as the inductor current IL decreases.

[0082] The switch 72 is an element for supplying the source current Is to the gate electrode of the IGBT 31 when the IGBT 31 is on, and the NMOS transistor 73 is an element for applying a ground voltage ("0 V") to the gate electrode of the IGBT 31 when the IGBT 31 is off. In addition, the on-resistance of the NMOS transistor 73 of the present embodiment is designed to be sufficiently small. Therefore, if the NMOS transistor 73 is on, the charge of the gate capacitance of the IGBT 31 is discharged in a short time. In addition, the switch 72 corresponds to a "first switch", and the NMOS transistor 73 corresponds to a "second switch".

[0083] The switch circuit 74 makes the switch 72 "on" and the NMOS transistor 73 "off when the drive signal Vdr2 is at the "H" level to make the IGBT 31 "on". As a result, the "source current Is" is supplied to the gate electrode of the IGBT 31, and as a result, the IGBT 31 is "on".

[0084] The switch circuit 74 makes the switch 72 "off and the NMOS transistor 73 "on when the drive signal Vdr2 is at the "L" level to make the IGBT 31 "off. As a result, the gate capacitance of the IGBT 31 is discharged, and the IGBT 31 is "off. Thus, the switch circuit 74 makes the IGBT 31 off regardless of the inductor current IL.

[0085] << Waveform when the IGBT 31 is "on"

[0086] Figure 9 is a graph showing one example of the waveforms when the IGBT 31 is "on". In addition, Figure 8 In the graph, the "dotted line" is one example of the waveforms when the IGBT 31 is "on" using the LVIC 25 instead of the LVIC 22, and the "solid line" is one example of the waveforms when the IGBT 31 is "on" using the LVIC 22. Figure 4

[0087] << Case where the LVIC 25 makes the IGBT 31 "on" (dotted line)

[0088] ​First, let's describe the waveform of the main node when the voltage-driven LVIC25 turns on IGBT31 at time t40, starting from the state where both IGBT30 and 31 are "off". (The waveform is shown as "dashed line" here.)

[0089] At time t40, when switch SW1 is turned on, a voltage Vcc (e.g., "15V") is applied to the gate electrode of IGBT31 through the on-resistance of switch SW1. As a result, an inrush current flows through the gate electrode of IGBT31, and the gate capacitance of IGBT31 is charged.

[0090] Then, as voltage Vg2 rises, the current supplied from LVIC25 to the gate electrode of IGBT31 decreases because the difference between voltage Vcc and voltage Vg2 becomes smaller. However, during this period, the gate capacitance of IGBT31 is also charged, thus causing voltage Vg2 to rise.

[0091] At time t41, when IGBT31 becomes "on", the inductor current IL increases, and therefore, the voltage Vs also rises. At this time, the voltage Vce between the emitter and collector electrodes of IGBT31 decreases from time t41, and by time t42 it has dropped to approximately "0V".

[0092] at this time, Figure 1 The voltage VR of FWD32 on the upper arm side is VR = Vdc - Vce. Therefore, starting from the moment t41 when IGBT31 becomes "on", it increases as the voltage Vce decreases, and at moment t42, it rises to approximately the voltage Vdc ("300V"). Thus, here, from moment t41 to moment t42, the voltage VR increases.

[0093] <<The case where LVIC22 enables IGBT31 to "conduct" (solid line)>>

[0094] Next, the waveform of the main node (the waveform shown as "solid line") when LVIC22 turns on IGBT31 at time t51, starting from the state where both IGBT30 and 31 are "off".

[0095] Additionally, the drive signal Vdr2, which is to "turn off" the IGBT31, occurs at time t50, prior to time t51. Figure 8 (Not shown in the diagram) The timing starts from when the signal reaches the "L" level until after a "predetermined time Tx". Therefore, at time t50, the detection circuit 51 maintains a voltage Va corresponding to the peak value of the inductor current IL before the IGBT 31 is "cut off". Furthermore, here, the peak value of the inductor current IL at time t50 is set as the "current value I0".

[0096] The setting circuit 70a outputs setting data SET corresponding to the "current value I0" at the time t50. Therefore, the current generating circuit 71 generates the source current Is corresponding to the "current value I0" when the switch 72 is on. Also, at this timing, the switch 72 is off, so the source current Is is not generated and is not supplied to the gate electrode of the IGBT 31.

[0097] When the drive signal Vdr2 (not shown in the figure) to be made "on" the IGBT 31 becomes the "H" level, at the time t51, the switching circuit 74 makes the switch 72 "on" and makes the NMOS transistor 73 "off". As a result, the prescribed source current Is corresponding to the "current value I0" is supplied to the gate electrode of the IGBT 31. Also, here, the prescribed source current Is is set to the "current value II". Figure 9

[0098] Then, the gate capacitance of the IGBT 31 is charged by the source current Is of the "current value II", so the gate voltage Vg2 gradually increases. Then, at the time t52, when the IGBT 31 becomes "on", the inductor current IL increases and the voltage Vs also rises. At this time, the voltage Vce between the emitter electrode and the collector electrode of the IGBT 31 gradually decreases from the time t52 and substantially decreases to "0 V" at the time t53.

[0099] Further, Figure 1 The voltage VR of the FWD 32 on the upper arm side shown in the figure becomes VR = Vdc - Vce, so it gradually increases as the voltage Vce decreases from the time t52 when the IGBT 31 becomes "on" and substantially rises to the voltage Vdc (= "300 V") at the time t53.

[0100] Here, in the case of using the general voltage drive type LVIC 25, as shown by the "dotted line", the voltage VR rises during the period from the time t41 to the time t42. On the other hand, in the case of using the LVIC 22 of the present embodiment, the voltage VR rises during the period from the time t52 to the time t53. Here, the period from the time t52 to the time t53 is longer than the period from the time t41 to the time t42. Therefore, as shown in the lowermost part of the figure, the sharp change in the voltage VR can be suppressed when using the LVIC 25, so the generation of noise can be suppressed. Figure 8

[0101] However, as explained in the "Background Art" section, when the inductor current IL decreases, the carrier of the FWD 32 decreases and the depletion layer of the FWD 32 spreads more quickly. Therefore, in general, when the inductor current IL decreases, the rise time of the voltage VR becomes shorter and the noise becomes larger. Figure 2 Figure 3 Therefore, in general, when the inductor current IL decreases, the rise time of the voltage VR becomes shorter and the noise becomes larger.​​​

[0102] However, when the inductor current IL becomes small, for example, the peak current of the IGBT 31 at the time t50 is smaller than the "current value I0". Thus, the voltage Va sampled at the time t50 also becomes small.

[0103] In this case, the current value of the source current Is is smaller than the "current value II", and thus the time for the gate capacitance of the IGBT 31 to be charged becomes long. Thus, in the present embodiment, even when the inductor current IL becomes small, the voltage VR can be prevented from changing sharply.

[0104] Further, when the inductor current IL becomes large, the number of carriers of the FWD 32 becomes large, and the depletion layer of the FWD 32 spreads slowly. At this time, the rise time of the voltage VR becomes longer than necessary, and the efficiency of the switching can be lowered.

[0105] However, in the present embodiment, as the inductor current IL increases, the current value of the source current Is also becomes large. Thus, the time for the gate capacitance of the IGBT 31 to be charged becomes short, and thus the drive circuit 52 can prevent the rise time of the voltage VR from becoming longer than necessary.

[0106] Comparison between LVIC 22 and LVIC 25

[0107] Figure 10 is a graph showing the relationship between the inductor current IL and the rate of change of the voltage VR. In addition, Figure 10 The "solid line" of is an example when the LVIC 22 turns the IGBT 31 "on", and the "dotted line" is an example when the LVIC 25 turns the IGBT 31 "on".

[0108] Thus, compared with the case where the IGBT 31 is turned on by the LVIC 25 of the general voltage drive type, by using the LVIC 22 of the present embodiment, the rate of change of the reverse bias voltage of the FWD 32, that is, the voltage VR can be reduced. As a result, the LVIC 22 can suppress the noise generated by the voltage VR. The LVIC 22 can also substantially fix the voltage VR even when the inductor current IL changes.

[0109] Other Embodiments

[0110] Other Embodiments of the Setting Circuit

[0111] Figure 11is a view showing one example of the setting circuit 70b. As with the setting circuit 70a, the setting circuit 70b is also a circuit that sets the current value of the source current Is generated by the current generation circuit 71 based on the detection result of the detection circuit 51. Specifically, the setting circuit 70b compares the voltage Vb sampled and held by the detection circuit 51 with the voltage Va during the next sampling period, and causes the source current Is to change based on the comparison result.

[0112] The setting circuit 70b is configured by including an edge detection circuit 150, a comparator 151, a D flip-flop 152, and a counter 153.

[0113] The edge detection circuit 150 detects the falling edge of the drive signal Vdr2, and outputs a pulse signal.

[0114] The comparator 151 (comparison circuit) compares the voltage Vb sampled and held by the detection circuit 51 with the voltage Va during the next sampling period. The comparator 151 outputs the comparison result Vc at the "L" level when the voltage Va is smaller than the voltage Vb, and outputs the comparison result Vc at the "H" level when the voltage Va is larger than the voltage Vb. Figure 8 In this case, the comparator 151 compares the voltage Vb sampled and held at the time t33 with the voltage Va, for example.

[0115] The D flip-flop 152 (holding circuit) holds the comparison result Vc of the comparator 151 at the timing when the drive signal Vdr2 becomes "L" based on the pulse signal from the edge detection circuit 150. For example, the D flip-flop 152 holds the comparison result Vc of the voltage Va with the voltage Vb held at the time t33 at the time t36.

[0116] The counter 153 is a circuit that causes the count value corresponding to the above-described "setting data SET" to change based on the comparison result Vc. Specifically, the counter 153 causes the count value (i.e., the "setting data SET") to change so that the source current Is becomes larger in the case where the comparison result Vc is at the "H" level, i.e., in the case where the inductor current IL increases. Further, the counter 153 causes the count value to change so that the source current Is becomes smaller in the case where the comparison result Vc is at the "L" level, i.e., in the case where the inductor current IL decreases. In addition, the counter 153 is set with an initial value so that a prescribed source current Is is generated. In the case where the setting circuit 70b is used instead of the setting circuit 70a, the same effects as in the case where the setting circuit 70a is used can be obtained.

[0117] Further, the timing when the IGBT 31 is turned off, i.e., the timing when the signal Vd becomes the "L" level (e.g., the time t33 of FIG. 15) corresponds to the "first timing", and the timing when the drive signal Fdr2 becomes the "L" level (e.g., the time t34 of FIG. 15) corresponds to the "second timing". Figure 8 Figure 8 ​The timing t36) corresponds to the "second timing".

[0118] Further, the voltage Va corresponding to the inductor current IL corresponds to the "first voltage", and the voltage Vb corresponds to the "second voltage".

[0119] == Power Module 200 ==

[0120] Figure 12 is a diagram showing one example of the power module 200. In the power module 200, the IGBT 35 is used in place of the IGBT 31 of the power module 10 of Figure 1 , and the resistor 36 is used in place of the resistor 13. In addition, in the power module 200, the blocks marked with the same reference numerals are the same as in the power module 10 of Figure 1 and Figure 12

[0121] The IGBT 35 is an element including an IGBT of a large size for controlling the inductor current IL and an IGBT of a small size for current detection, and the current from the IGBT for current detection is supplied to the resistor 36. Thus, in the case of using the IGBT 35, the voltage Vs corresponding to the inductor current IL as a load and the current flowing through the IGBT 35 is generated in the resistor 36. As a result, the detection circuit 51 of the power module 200 can detect the inductor current IL, and thus the power module 200 functions to have the same effect as the power module 10.

[0122] == Power Module 210 ==

[0123] Figure 13 is a diagram showing one example of the power module 210. The power module 210 is a semiconductor device for driving a three-phase motor 220, and includes a bridge circuit 300 for power conversion, HVICs 301 to 303, an LVIC 304, and terminals HU, HV, HW, LU, LV, LW, P, U, V, W, NU, NV, and NW.

[0124] The bridge circuit 300 is configured to include IGBTs 400 to 405, FWDs 410 to 415, and resistors Ru, Rv, and Rw. In addition, the bridge circuit 300 corresponds to the "switching circuit".

[0125] The IGBTs 400 and 401 are switching elements of the U phase, and the IGBTs 400 and 401 are respectively provided with the FWDs 410 and 411.

[0126] The IGBTs 402 and 403 are switching elements of the V phase, and the IGBTs 402 and 403 are respectively provided with the FWDs 412 and 413.

[0127] The IGBTs 404 and 405 are switching elements of the W phase, and the IGBTs 404 and 405 are respectively provided with the FWDs 414 and 415.​

[0128] In this embodiment, IGBTs 400, 402, and 404 each correspond to the "first switching element", and IGBTs 401, 403, and 405 each correspond to the "second switching element". In addition, FWDs 410, 412, and 414 each correspond to the "first return diode", and FWDs 411, 413, and 415 each correspond to the "second return diode".

[0129] Resistors Ru, Rv, and Rw are the resistors that detect the current flowing through the switching elements of phases U, V, and W, respectively. Furthermore, the voltages generated in resistors Ru, Rv, and Rw are designated as "Voltage Vsu", "Voltage Vsv", and "Voltage Vsw", respectively.

[0130] A power supply voltage Vdc is applied to terminal P. A three-phase motor 220 is installed as a load at terminals U, V, and W. Terminals NU, NV, and NW are grounded.

[0131] HVIC301 to 303 are integrated circuits used to switch the IGBTs 400, 402, and 404 of the upper arm using signals from a microcomputer (not shown) input via terminals HU, HV, and HW.

[0132] The LVIC304 is an integrated circuit used to switch the IGBTs 401, 403, and 405 in the lower arm using drive signals Vdru, Vdrv, and Vdrw from a microcomputer (not shown) input via terminals LU, LV, and LW.

[0133] Although the internal structure diagram of LVIC304 is omitted, LVIC304 contains 3 phases. Figure 6 The LVIC22 comprises three circuits: filter 50, detection circuit 51, and drive circuit 52. Specifically, the LVIC304 includes a filter 50, detection circuit 51, and drive circuit 52 for the U-phase, a filter 50, detection circuit 51, and drive circuit 52 for the V-phase, and a filter 50, detection circuit 51, and drive circuit 52 for the W-phase. As a result, the LVIC304 can suppress changes in the reverse bias voltage of the upper arms FWD410, 412, and 414 even when driving the three-phase motor 220. Therefore, noise generated in the power module 210 can be reduced.

[0134] ===Summary===

[0135] The power module 10, 200, 210 of the present embodiment has been described above. In general, when the inductor current IL decreases, the change in the reverse bias voltage VR of the FWD 32 of the upper arm becomes sharp. However, the drive circuit 52 controls the magnitude of the current that charges the gate capacitance of the IGBT 31 in accordance with the magnitude of the inductor current IL. Specifically, the drive circuit 52 decreases the source current Is as the inductor current IL decreases, and thus the period during which the gate capacitance of the IGBT 31 is charged becomes longer. As a result, the drive circuit 52 turns on the IGBT 31 for a period that is longer than in the case where the inductor current IL is large (for example, the period from time t52 to t53 in FIG. 8). Thus, even in the case where the inductor current IL decreases, the change in the voltage VR can be prevented from becoming sharp, and noise can be suppressed. Figure 9

[0136] In addition, as a structure in which the IGBT 31 is turned on for a period that becomes longer as the inductor current IL decreases, a voltage-driven LVIC can also be used. For example, if the on resistance of the switch SW1 of the LVIC 25 is increased as the inductor current IL decreases, the same effect as in the present embodiment can be obtained. However, in the case where the voltage-driven LVIC 25 is used, a surge current (for example, the current at time t40 in FIG. 8) can sometimes occur when the IGBT 31 is turned on. The LVIC 22 is a current-driven integrated circuit, and thus a surge current can be prevented from occurring when the IGBT 31 is driven, and thus switching noise can be reduced. Figure 9

[0137] In addition, the drive circuit 52 uses the NMOS transistor 73 when the IGBT 31 is turned off. Thus, the gate capacitance of the IGBT 31 can be discharged immediately.

[0138] In addition, the detection circuit 51 samples and holds the voltage Va at a predetermined timing at which the IGBT 31 is turned off. In this way, the timing at which the sampling is performed is determined in advance, and thus the source current Is can be generated with high accuracy. In addition, the timing at which the IGBT 31 is turned off is a timing at which the inductor current IL and the current flowing through the IGBT 31 become the largest.

[0139] In addition, the source current Is can also be adjusted on the basis of whether the voltage Va is larger than the sampled voltage Vb. Even with this method, the LVIC 22 can suppress the change in the reverse bias voltage VR of the FWD 32.

[0140] In addition, in the present embodiment, the filter 50 that suppresses noise of the voltage Vs is provided, and the output of the filter 50 is detected. Thus, the drive circuit 52 can generate the source current Is corresponding to the inductor current IL with high accuracy.

[0141] ​​Moreover, the switching element included in the half-bridge circuit 20, the bridge circuit 300, for example, is an IGBT, but can be a MOS transistor, for example. In this case, the return flow diode for the switching element is realized by the body diode of the MOS transistor.

[0142] The above-described embodiments are for easy understanding of the present application, and are not intended to limit and explain the present application. Moreover, the present application can be changed and improved without departing from the idea thereof, and equivalent applications of the present application are of course included in the present application.

[0143] The current generation circuit 71 of the present embodiment adjusts the source current Is by switching the on and off of the switching switches 110 to 112, but is not limited thereto. For example, it can be a voltage-controlled current source that changes the current value of the source current Is using the voltage Vb of the detection circuit 51.

[0144] Moreover, as the circuit for power conversion, it can not be the half-bridge circuit 20, but can be an H-bridge circuit or another circuit, for example. In this case, the H-bridge circuit corresponds to the "switching circuit".

[0145] Label Explanation

[0146] 10, 200, 210 Power module

[0147] 11 Inductor

[0148] 12, 63, 221 Capacitor

[0149] 13, 36, 120 to 122, Ru, Rv, Rw Resistor

[0150] 20 Half-bridge circuit

[0151] 21, 301 to 303 HVIC

[0152] 22, 25, 304 LVIC

[0153] 30, 31, 35, 400 to 405 IGBT

[0154] 32, 33, 410 to 415 FWD

[0155] 50 Filter

[0156] 51 Detection circuit

[0157] 52 Drive circuit

[0158] 60 Delay circuit

[0159] 61, 150 Edge detection circuit

[0160] 62, 72, 110-112, SW1, SW2 switches

[0161] 64 unity gain buffer

[0162] 70 setting circuit

[0163] 71 current generating circuit

[0164] 73 NMOS transistor

[0165] 100 current source

[0166] 101-104 PNP transistors

[0167] 151 comparator

[0168] 152 D flip-flop

[0169] 153 counter

[0170] 220 three-phase motor

[0171] 300 bridge circuit

Claims

1. An integrated circuit that switches a second switching element of a switching circuit, the switching circuit including a first switching element on a power supply side and the second switching element on a ground side connected in series, a first freewheeling diode connected in parallel with the first switching element, and a second freewheeling diode connected in parallel with the second switching element, the integrated circuit characterized by comprising: a detection circuit that detects a load current flowing through a load of the switching circuit; and a drive circuit that controls a size of a current that charges a gate capacitance of the second switching element in accordance with a size of the load current when a drive signal is one logic level, and causes the second switching element to be turned off when the drive signal is another logic level, the detection circuit being a sample-and-hold circuit that samples and holds a voltage corresponding to the load current at a first timing, the drive circuit including: a comparison circuit that compares a size of a first voltage corresponding to the load current with a size of a second voltage held by the sample-and-hold circuit; a holding circuit that holds a comparison result of the comparison circuit at a second timing; a current generation circuit that generates a source current that becomes larger when the comparison result indicating that the first voltage is larger than the second voltage is held in the holding circuit, and becomes smaller when the comparison result indicating that the first voltage is smaller than the second voltage is held in the holding circuit; and a switch that supplies the source current to a control electrode of the second switching element when the drive signal is the one logic level.

2. The integrated circuit according to claim 1, characterized by comprising a second switch that applies a ground voltage to the control electrode of the second switching element when the drive signal becomes the another logic level.

3. The integrated circuit according to claim 1 or 2, characterized by further comprising a filter that suppresses noise of the voltage corresponding to the load current and outputs the voltage to the detection circuit.

4. The integrated circuit according to claim 1 or 2, characterized in that the first switching element and the second switching element are each an insulated gate bipolar transistor.

5. The integrated circuit according to claim 1 or 2, characterized in that the first switching element and the second switching element are each a MOS transistor, the first freewheeling diode and the second freewheeling diode are each a body diode of the MOS transistor.

6. A semiconductor device that includes: a switching circuit including a first switching element on a power supply side and a second switching element on a ground side connected in series, a first freewheeling diode connected in parallel with the first switching element, and a second freewheeling diode connected in parallel with the second switching element; and an integrated circuit that switches the second switching element of the switching circuit, the semiconductor device characterized in that the integrated circuit includes: a detection circuit that detects a load current flowing through a load of the switching circuit; and a drive circuit that controls a size of a current that charges a gate capacitance of the second switching element in accordance with a size of the load current when a drive signal is one logic level, and causes the second switching element to be turned off when the drive signal is another logic level, the detection circuit being a sample-and-hold circuit that samples and holds a voltage corresponding to the load current at a first timing, the drive circuit including: a comparison circuit that compares a size of a first voltage corresponding to the load current with a size of a second voltage held by the sample-and-hold circuit; a holding circuit that holds a comparison result of the comparison circuit at a second timing; a current generation circuit that generates a source current that becomes larger when the comparison result indicating that the first voltage is larger than the second voltage is held in the holding circuit, and becomes smaller when the comparison result indicating that the first voltage is smaller than the second voltage is held in the holding circuit; and a switch that supplies the source current to a control electrode of the second switching element when the drive signal is the one logic level. ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ ​ A switching circuit includes a first switching element on a power supply side and a second switching element on a ground side connected in series, a first backflow diode connected in parallel with the first switching element, and a second backflow diode connected in parallel with the second switching element. ​ ​ ​ ​ a drive circuit which controls the magnitude of a current for charging a gate capacitance of the second switching element in accordance with the magnitude of the load current when the drive signal is one logic level, and which causes the second switching element to be turned off when the drive signal is another logic level, the detection circuit is a sample-and-hold circuit which samples and holds a voltage corresponding to the load current at a first timing, the drive circuit includes: a comparison circuit which compares the magnitude of a first voltage corresponding to the load current with a second voltage held by the sample-and-hold circuit; a holding circuit which holds a comparison result of the comparison circuit at a second timing; a current generation circuit which generates a source current which becomes larger when the comparison result indicating that the first voltage is larger than the second voltage is held in the holding circuit, and which becomes smaller when the comparison result indicating that the first voltage is smaller than the second voltage is held in the holding circuit; and a switch which supplies the source current to a control electrode of the second switching element when the drive signal is the one logic level.

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