Nonlinear robust control method and device for DC-DC converter

Abstract

The invention belongs to the technical field of power electronics, and provides a nonlinear robust control method and device for a DC-DC converter. The method comprises the following steps: acquiring an output voltage of the DC-DC converter and feeding back the output voltage to a terminal sliding mode controller, calculating a voltage error between a reference voltage and the output voltage of the DC-DC converter by the terminal sliding mode controller, and calculating the voltage error of the DC-DC converter according to a relation among a preset fractional order nonlinear sliding mode surface, a linear sliding mode surface and the voltage error, generating a first control signal as a reference current of the linear controller to form a voltage loop; and the output current of the DC-DC converter is obtained and fed back to the linear controller, the linear controller generates a second control signal in combination with the reference current and acts on a set power switch tube of the DC-DC converter to form a current loop, and finally DC-DC converter double-loop nonlinear robust control based on cooperation of the reference voltage, the output voltage and the output current is achieved. And good transient response is ensured.

Description

Technical Field

[0001] The present invention belongs to the technical field of power electronics, and particularly relates to a non-linear robust control method and device for a DC-DC converter. Background Art

[0002] The statements in this part merely provide background technical information related to the present invention and do not necessarily constitute prior art.

[0003] In order to improve the voltage transfer efficiency and transient response of the DC-DC converter, it is usually achieved by changing the control strategy and modulation strategy. Among them, in terms of the control strategy, the traditional control strategies include linear control strategies and traditional terminal sliding mode control strategies. Among them, for the traditional linear control strategy, when the load changes suddenly or the input voltage fluctuates, the dynamic response of the converter is slow and overshoot is likely to occur; although the prior art combines the linear control strategy with the traditional terminal sliding mode control strategy, these two can only handle the uncertainties of phase and control law matching, and there are still relatively slow convergence speed and chattering phenomenon (i.e., high-frequency oscillation), resulting in poor output voltage stability of the DC-DC converter. Summary of the Invention

[0004] In order to solve the technical problems existing in the above background art, the present invention provides a non-linear robust control method and device for a DC-DC converter, which can ensure the output voltage stability of the converter.

[0005] In order to achieve the above object, the present invention adopts the following technical solutions:

[0006] The first aspect of the present invention provides a non-linear robust control method for a DC-DC converter.

[0007] In one or more embodiments, a non-linear robust control method for a DC-DC converter includes:

[0008] Obtain the output voltage of the DC-DC converter and feedback it to the terminal sliding mode controller. The terminal sliding mode controller calculates the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generates a first control signal according to the relationship between the preset fractional-order non-linear sliding mode surface, linear sliding mode surface and voltage error, and uses it as the reference current of the linear controller to form a voltage loop;

[0009] Obtain the output current of the DC-DC converter and feedback it to the linear controller. The linear controller combines the reference current to generate a second control signal and acts on the set power switch tube of the DC-DC converter to form a current loop, and finally realizes the dual-loop non-linear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage and the output current of the DC-DC converter.

[0010] In some other embodiments, a control method for a DC-DC converter, which is applicable to a dual-bridge DC-DC converter, includes:

[0011] Obtain the output voltage of the DC-DC converter and feedback it to the terminal sliding mode controller. The terminal sliding mode controller calculates the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generates a first control signal based on the relationship between the preset fractional-order nonlinear sliding mode surface, the linear sliding mode surface, and the voltage error, and uses it as the reference current of the linear controller to form a voltage loop;

[0012] Obtain the output current of the DC-DC converter and feedback it to the linear controller. The linear controller generates and uses it as the external phase shift ratio signal of the converter in combination with the reference current to form a current loop;

[0013] Utilize the relationship between the preset internal phase shift ratio and the external phase shift ratio in the phase shift controller to obtain the internal phase shift ratio signal of the DC-DC converter, thereby realizing the dual-loop nonlinear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage, and the output current of the DC-DC converter.

[0014] The second aspect of the present invention provides a DC-DC converter.

[0015] A DC-DC converter includes: power switching tubes, a terminal sliding mode controller for the outer loop, and a linear controller for the inner loop;

[0016] The terminal sliding mode controller is used to calculate the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generate a first control signal based on the relationship between the preset fractional-order nonlinear sliding mode surface, the linear sliding mode surface, and the voltage error, and use it as the reference current of the linear controller to form a voltage loop;

[0017] The linear controller is used to generate a second control signal based on the output current of the DC-DC converter and the reference current and act on the set power switching tubes of the DC-DC converter to form a current loop, and finally realize the dual-loop nonlinear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage, and the output current of the DC-DC converter.

[0018] A DC-DC converter includes power switching tubes, a phase shift controller, a terminal sliding mode controller for the outer loop, and a linear controller for the inner loop;

[0019] The terminal sliding mode controller is used to calculate the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generate a first control signal as the reference current of the linear controller according to the relationship between the preset fractional-order nonlinear sliding mode surface, the linear sliding mode surface and the voltage error, so as to form a voltage loop;

[0020] The linear controller is used to generate a second control signal as the external phase shift ratio signal of the DC-DC converter based on the output current of the DC-DC converter and the reference current, so as to form a current loop;

[0021] The phase shift controller is used to obtain the internal phase shift ratio signal of the DC-DC converter by using the relationship between the preset internal phase shift ratio and the external phase shift ratio in the phase shift controller, so as to realize the dual-loop nonlinear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage and the output current of the DC-DC converter.

[0022] In one or more embodiments, a DC-DC converter as described above is coupled to a power generation system, an energy storage system or an electric vehicle power supply.

[0023] Compared with the prior art, the beneficial effects of the present invention are:

[0024] (1) The present invention uses the terminal sliding mode controller to calculate the voltage error between the reference voltage and the output voltage of the DC-DC converter, and then uses the relationship between the preset fractional-order nonlinear sliding mode surface, the linear sliding mode surface and the voltage error as the terminal sliding mode control strategy to obtain a first control signal as the reference current of the linear controller, forming a voltage loop; then uses the linear controller as the inner loop of the DC-DC converter to control the output current, realizing the chattering-free robust control and dual-loop control that converge in a finite time, and ensuring the stability of the output voltage under the conditions of changing the input and output voltages and varying the load of the DC-DC converter.

[0025] (2) For the dual-bridge DC-DC converter, it also determines the relationship between the internal phase shift ratio and the external phase shift ratio according to the phase shift controller and the topology of the DC-DC converter, and then combines the terminal sliding mode controller, the linear controller and the phase shift controller to calculate the internal phase shift ratio and the external phase shift ratio of the converter, improving the voltage transfer efficiency of the DC-DC converter while also accelerating the response speed of the DC-DC converter, so that the DC-DC converter quickly enters a stable state.

[0026] (3) The control strategy of the DC-DC converter of the present invention is applicable to the fast response charging and discharging requirements on the battery side in battery testing, and can be widely used in the fields of power generation systems, energy storage systems and electric vehicle charging and discharging.

[0027] Advantages of additional aspects of the present invention will be given in part in the following description, become apparent in part from the following description, or be learned through the practice of the present invention. BRIEF DESCRIPTION OF THE DRAWINGS

[0028] The accompanying drawings forming a part of this specification are used to provide a further understanding of the present invention. The schematic embodiments of the present invention and their descriptions are used to explain the present invention and do not constitute an improper limitation of the present invention.

[0029] Figure 1 is the main circuit topology diagram of a dual active bridge (DAB) converter;

[0030] Figure 2 is the flowchart of the control method of the converter in the first embodiment of the present invention;

[0031] Figure 3 is the control schematic diagram of the converter in the first embodiment of the present invention;

[0032] Figure 4 is the flowchart of the control method of the converter in the second embodiment of the present invention;

[0033] Figure 5 is the control schematic diagram of the converter in the second embodiment of the present invention;

[0034] Figure 6 is the traditional single-phase-shift double-loop PI linear control block diagram of a DAB converter;

[0035] Figure 7 is the TSM-PI double-loop control block diagram of single-phase-shift of a DAB converter;

[0036] Figure 8 is the transient response of three controllers, namely PI-PI, SM-PI, and TSM-PI, when the input voltage increases;

[0037] Figure 9 is the transient response of three controllers, namely PI-PI, SM-PI, and TSM-PI, when the output voltage increases;

[0038] Figure 10 is the transient response of three controllers, namely PI-PI, SM-PI, and TSM-PI, when the load changes;

[0039] Figure 11 is the voltage and current waveform diagram of double-phase-shift of a DAB converter. DETAILED DESCRIPTION OF THE EMBODIMENTS

[0040] The present invention will be further described below in conjunction with the accompanying drawings and embodiments.

[0041] It should be noted that the following detailed description is illustrative and is intended to provide further explanation of the present invention. Unless otherwise specified, all technical and scientific terms used herein have the same meaning as commonly understood by those of ordinary skill in the technical field to which the present invention pertains.

[0042] It should be noted that the terms used herein are only for describing specific embodiments and are not intended to limit the exemplary embodiments according to the present invention. As used herein, unless the context clearly indicates otherwise, the singular forms are also intended to include the plural forms. In addition, it should be understood that when the terms "comprising" and / or "including" are used in this specification, they specify the presence of features, steps, operations, devices, components, and / or combinations thereof.

[0043] Term Explanation:

[0044] DAB: Dual Active Bridge, a dual-active-bridge converter;

[0045] PI: Proportional-Integral, proportional integral;

[0046] PID: Proportional-Integral-Derivative, proportional integral derivative;

[0047] TSM: Terminal Sliding Mode, terminal sliding mode control;

[0048] EPS: Extended Phase-Shift, extended phase shift.

[0049] The DC-DC converter topology of the present invention includes, but is not limited to, buck, boost, buck-boost, flyback, and dual-active-bridge converters, etc.

[0050] Taking the dual-active-bridge converter as an example below, when calculating the duty cycle of the dual-active-bridge converter by combining a terminal sliding mode controller and a linear controller, the obtained duty cycle is used to implement the single-phase-shift control (SPWM) of the dual-active-bridge converter to adjust the phase-shift angle of the output voltage of the primary-side or secondary-side H-bridge.

[0051] Figure 1 The main circuit topology diagram of the dual-active-bridge converter (DAB) is given. The DAB converter is composed of two H-bridge converters (respectively set as the H1-bridge converter and the H2-bridge converter) connected by a transformer. Figure 1 In, the turns ratio of the primary and secondary sides of the high-frequency isolation transformer is set to k, k = 1:n; U in, i1 are respectively the input voltage (DC power supply) and input current of the DAB converter; The H1-bridge converter consists of an H-bridge formed by power switching transistors S1 to S4 and a support capacitor C1 on the input side connected in parallel therewith; The H2-bridge converter consists of an H-bridge formed by power switching transistors S5 to S8 and a filter capacitor C2 on the output side connected in parallel therewith; Among them, R represents the load, U o , i o are respectively the output voltage and current of the DAB converter; L s is the equivalent inductance of the transformer; U ab is the primary side of the DAB converter; U cd is the secondary side voltage of the DAB converter; i2 is the secondary side current of the DAB converter; i c is the current flowing into the filter capacitor C2.

[0052] Figure 2 discloses a non-linear robust control method for a DC-DC converter according to an embodiment of the present invention, and this method is generally applicable to various different forms of topological structures. Figure 3 takes the DAB converter as an example, and adopts the Figure 2 shown principle block diagram of the non-linear robust control method of the DC-DC converter.

[0053] Next, in conjunction with Figure 2 and Figure 3 , the following specific implementation process of the non-linear robust control method of the DC-DC converter is given, which includes:

[0054] S11: Obtain the output voltage of the DC-DC converter and feedback it to the terminal sliding mode controller. The terminal sliding mode controller calculates the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generates a first control signal as the reference current of the linear controller according to the relationship between the preset fractional-order non-linear sliding mode surface, linear sliding mode surface and voltage error, so as to form a voltage loop.

[0055] Figure 3 Among them, the reference voltage is U ref , and its value is the expected voltage matched and designed according to the specific reference of the DC-DC converter. The voltage error is set as e; The expression of the voltage error is:

[0056] e = U ref - U o (1)

[0057] The relationship between the fractional-order non-linear sliding mode surface, linear sliding mode surface and voltage error here is:

[0058] The fractional-order non-linear sliding mode surface is the fractional-order integral of the linear sliding mode surface and the weighted sum of the linear sliding mode surfaces; the linear sliding mode surface is the integral of the voltage error and the weighted sum of the voltage error.

[0059] Let the fractional-order non-linear sliding mode surface be σ, the linear sliding mode surface be s, and the relational expressions of the fractional-order non-linear sliding mode surface, the linear sliding mode surface, and the voltage error be:

[0060]

[0061] wherein, c1, c2, λ1, and λ2 are all numbers greater than zero and are all weighted values, i.e., gains; p and q are both odd numbers and satisfy

[0062] By differentiating the fractional-order non-linear sliding mode surface and the linear sliding mode surface, such that and both become 0, enabling the system to converge to the sliding mode surface within a finite time:

[0063]

[0064] Then, in combination with the energy equation of the DC-DC converter, the output quantity of the terminal sliding mode controller is obtained and used as the input reference current I of the linear controller SM :

[0065]

[0066] wherein, α is a constant coefficient greater than zero.

[0067] In Figure 3 the linear controller is a PI controller.

[0068] It should be noted here that the linear controller can also be a PID controller, and those skilled in the art can specifically set it according to the actual situation, which will not be elaborated here.

[0069] S12: Obtain the output current of the DC-DC converter and feedback it to the linear controller. The linear controller generates a second control signal in combination with the reference current and acts on the set power switch tube of the DC-DC converter to form a current loop, ultimately realizing the dual-loop non-linear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage, and the output current of the DC-DC converter.

[0070] For example, Figure 3 the duty cycle obtained in

[0071] is used to implement the single-phase phase-shift control of the dual-active bridge converter to adjust the phase-shift angle of the output voltage of the secondary H-bridge. Figure 3The duty cycle obtained can also be used to adjust the phase shift angle of the output voltage of the primary H-bridge.

[0072] In addition, according to the actual situation, the DAB converter can be converted into other converters to calculate the corresponding duty cycle, which will not be elaborated here.

[0073] The traditional integral sliding mode formula is s = c1e + c2∫(e)dt; although this traditional integral sliding mode control can eliminate the phase reach and the smoothness of the control law, it can only handle matched uncertainties, has a relatively slow convergence speed, and there is a chattering phenomenon, that is, high-frequency oscillation, resulting in inaccurate voltage in the DAB voltage control, increasing the positioning error and reducing the control performance.

[0074] The terminal sliding mode controller of this embodiment adopts a non-linear sliding mode surface based on fractional order which provides a faster convergence speed of the system state around the equilibrium point, that is, finite-time convergence. Essentially, it makes the sliding mode surface σ and its n - 1th derivative disappear, and the chattering induced alternately in σ by the discontinuous control essentially eliminates the chattering effect. Therefore, the terminal sliding mode controller of this embodiment can obtain better sliding mode accuracy and can achieve chattering-free robust control with finite-time convergence. n This embodiment uses the terminal sliding mode controller as the outer loop of the converter to control the output voltage, and uses the linear controller as the inner loop of the DC-DC converter to control the output current. Under the conditions of changing the input and output voltages and variable load of the DC-DC converter, the stability of the output voltage can still be guaranteed.

[0075] In another embodiment, the duty cycle of the converter is calculated by combining the terminal sliding mode control strategy, the linear control strategy and the phase shift control strategy. The phase shift control strategy here can be: the extended phase shift control (EPS) strategy or the dual phase shift control (DPS) strategy.

[0076] The corresponding flowchart of the non-linear robust control method for the DC-DC converter is given. Figure 4 The corresponding principle block diagram of the converter is given with the DAB converter as an example. Figure 5 The corresponding principle block diagram of the converter is given with the DAB converter as an example. Figure 4 The corresponding principle block diagram.

[0077] It should be noted here that Figure 4 the non-linear robust control method for the DC-DC converter given is applicable to the dual-bridge DC-DC converter, such as the dual active bridge converter (DAB), the dual-bridge series resonant converter (DBSRC), etc.

[0078] Next, in combination with Figure 4 and Figure 5 the specific implementation process of the terminal sliding mode dual-loop control method of this embodiment is given, which includes:

[0079] S21: Obtain the output voltage of the DC-DC converter and feedback it to the terminal sliding mode controller. The terminal sliding mode controller calculates the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generates a first control signal as the reference current of the linear controller according to the relationship between the preset fractional-order nonlinear sliding mode surface, the linear sliding mode surface and the voltage error, so as to form a voltage loop;

[0080] Figure 4 Among them, the reference voltage is U ref , and its value is the desired voltage matched and designed according to the specific reference of the converter. The voltage error is set as e; the expression of the voltage error is the same as that in formula (1).

[0081] The relationship between the fractional-order nonlinear sliding mode surface, the linear sliding mode surface and the voltage error here is:

[0082] The fractional-order nonlinear sliding mode surface is the fractional-order integral of the linear sliding mode surface and the weighted sum of the linear sliding mode surface; the linear sliding mode surface is the integral of the voltage error and the weighted sum of the voltage error.

[0083] Let the fractional-order nonlinear sliding mode surface be σ, the linear sliding mode surface be s, and the relationship expression between the fractional-order nonlinear sliding mode surface, the linear sliding mode surface and the voltage error is as shown in formula (2).

[0084] By taking the derivatives of the fractional-order nonlinear sliding mode surface and the linear sliding mode surface, make and both be 0, so that the system can converge to the sliding mode surface within a finite time. The derivatives of the corresponding fractional-order nonlinear sliding mode surface and linear sliding mode surface are as shown in formula (3).

[0085] Then, combined with the energy equation of the converter:

[0086]

[0087] Among them, P o and P are the average power and the output power respectively; E C is the energy of the filter capacitor on the output side.

[0088] The reduced-order equation of the DAB converter is obtained as:

[0089]

[0090] Among them, I and U are the instantaneous current value on the secondary side of the DAB converter and the instantaneous voltage value across the load respectively.

[0091] Combined with the reduced-order equation of the DAB converter, the output quantity of the terminal sliding mode control strategy is obtained and used as the input reference current I SM of the linear control strategy, as shown in formula (4).

[0092] In Figure 5 , the linear controller is a PI controller.

[0093] It should be noted here that the linear controller can also adopt a PID controller, and those skilled in the art can set it specifically according to the actual situation, which will not be elaborated here.

[0094] S22: Obtain the output current of the DC-DC converter and feedback it to the linear controller. The linear controller combines the reference current to generate and use it as the external phase-shift ratio signal of the converter to form a current loop.

[0095] S23: Utilize the relationship between the preset internal phase-shift ratio and the external phase-shift ratio in the phase-shift controller to obtain the internal phase-shift ratio signal of the DC-DC converter, thereby realizing the dual-loop nonlinear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage, and the output current of the DC-DC converter.

[0096] Among them, the relationship between the internal phase-shift ratio and the external phase-shift ratio is characterized by the input voltage of the converter, the output voltage, and the turns ratio of the transformer in the DC-DC converter.

[0097] In the process of determining the relationship between the internal phase-shift ratio and the external phase-shift ratio, different phase-shift control strategies control different converter topologies, and the specific expressions of the corresponding internal phase-shift ratio and the external phase-shift ratio are different. Here, taking the extended phase-shift control (EPS) strategy as an example to control the DAB converter will be elaborated in detail. Among them, the extended phase-shift control (EPS) strategy / dual phase-shift control (DPS) strategy can determine the specific expressions of the internal phase-shift ratio and the external phase-shift ratio corresponding to other dual-bridge converter topologies according to their strategy control characteristics.

[0098] Figure 11 The voltage and current waveforms of the double phase-shift of the DAB converter are given. From Figure 11 it can be seen that within one switching period, it can be divided into six switching modes. In the first switching mode, only the power switches S1, S3, S6, and S7 are turned on, and other power switches are turned off; in the second switching mode, only the power switches S1, S2, S6, and S7 are turned on, and other power switches are turned off; in the third switching mode, the power switches S1, S2, S5, and S8 are turned on, and other power switches are turned off; in the fourth switching mode, the power switches S2, S4, S5, and S8 are turned on, and other power switches are turned off; in the fifth switching mode, the power switches S3, S4, S5, and S8 are turned on, and other power switches are turned off; in the sixth switching mode, the power switches S3, S4, S6, and S7 are turned on, and other power switches are turned off.

[0099] Under the control of the extended phase-shift control (EPS) strategy, the inductor current of the DAB converter can be expressed as:

[0100]

[0101] The average power (P o ) delivered using EPS modulation is expressed as:

[0102]

[0103] where n represents the turns ratio of the transformer, Ls represents the equivalent inductance, f represents the switching frequency, and D1 and D2 are the internal and external phase-shift ratios of the DAB converter, respectively.

[0104] It is known that the relationship between D1 and D2 is:

[0105]

[0106] where k is expressed by the input and output voltages of the DAB converter and n.

[0107] In this embodiment, the EPS control strategy is used to adjust the internal and external phase-shift ratios, reducing the back power and improving the conversion efficiency.

[0108] This embodiment combines the PI control strategy with the voltage-current double closed-loop control, ensuring the stability of the output voltage, improving the dynamic response speed, and still maintaining a robust dynamic performance under different input and output conditions, thus improving the stability of the system.

[0109] The two-level DAB under the control method of this embodiment is simulated. The DAB circuit parameters and simulation parameters are shown in Table 1 and Table 2, respectively.

[0110] Table 1 Simulation parameters

[0111]

[0112]

[0113] Table 2 Controller parameters

[0114] Parameter <![CDATA[c1]]> 5 <![CDATA[c2]]> 40.521 <![CDATA[λ1]]> 0.03 <![CDATA[λ2]]> 15 p 7 q 5 <![CDATA[K p > 0.1 <![CDATA[K i > 0.1 α <![CDATA[1×10 -7 >

[0115] Figure 6 is the traditional single-phase-shift double closed-loop PI linear control block diagram of the DAB converter; two PI controllers are used in Figure 6 to implement the traditional single-phase-shift double closed-loop PI linear control. Figure 7 is the single-phase-shift TSM-PI double closed-loop control block diagram of the DAB converter according to the embodiment of the present invention.

[0116] Figure 8 The transient responses of three controllers, namely PI-PI, SM-PI, and TSM-PI, are given when the input voltage increases. Figure 9 The transient responses of three controllers, namely PI-PI, SM-PI, and TSM-PI, are given when the output voltage increases. Figure 10 The transient responses of three controllers, namely PI-PI, SM-PI, and TSM-PI, are given when the load changes.

[0117] In Figure 8 (a) of in when the input voltage increases from 80V to 100V, during the process of the PI-PI controller making the output voltage U o re-stabilize, a voltage fluctuation with an amplitude of 1V will occur. In Figure 8 (b) of in when the input voltage increases from 80V to 100V, during the process of the SM-PI controller making the output voltage U o re-stabilize, a voltage fluctuation with an amplitude of 3V will occur. In Figure 8 (c) of in when the input voltage increases from 80V to 100V, during the process of the TSM-PI controller making the output voltage U o re-stabilize, the amplitude of the voltage fluctuation is almost zero.

[0118] In Figure 9 (a) of o when the output voltage increases from 90V to 120V, the response time for the PI-PI controller to make the output voltage U Figure 9 (b) of o re-stabilize is 4ms. In Figure 9 (c) of o when the output voltage increases from 90V to 120V, the response time for the TSM-PI controller to make the output voltage U

[0119] In Figure 10 (a) of in when the load decreases from 30Ω to 15Ω, during the process of the PI-PI controller making the output voltage U o re-stabilize, a voltage fluctuation with an amplitude of 7V will occur, and the response time is 80ms. In Figure 10 (b) of in when the load decreases from 30Ω to 15Ω, during the process of the SM-PI controller making the output voltage Uo During the process of re - stabilization, there will be a voltage fluctuation with an amplitude of 6.6V, and the response time is 2ms. In Figure 10 (c) of in when the load decreases from 30Ω to 15Ω, the TSM - PI controller makes the output voltage U o During the process of re - stabilization, there will be a voltage fluctuation with an amplitude of 5V, and the response time is 2ms.

[0120] From the comparison in terms of response speed and output voltage stability, it can be seen that the TSM - PI controller of the present invention is faster and has no overshoot.

[0121] In one or more embodiments, a DC - DC converter is further provided, which includes: a power switch tube, a terminal sliding - mode controller for the outer loop, and a linear controller for the inner loop;

[0122] The terminal sliding - mode controller is used to calculate the voltage error between the reference voltage and the output voltage of the DC - DC converter, and generate a first control signal as the reference current of the linear controller according to the relationship among a preset fractional - order nonlinear sliding - mode surface, a linear sliding - mode surface, and the voltage error, so as to form a voltage loop;

[0123] The linear controller is used to generate a second control signal based on the output current of the DC - DC converter and the reference current and act on the set power switch tube of the DC - DC converter, so as to form a current loop, and finally realize the dual - loop nonlinear robust control of the DC - DC converter based on the coordination of the reference voltage, the output voltage, and the output current of the DC - DC converter.

[0124] Wherein, the relationship among the fractional - order nonlinear sliding - mode surface, the linear sliding - mode surface, and the voltage error is:

[0125] The fractional - order nonlinear sliding - mode surface is the fractional - order integral of the linear sliding - mode surface and the weighted sum of the linear sliding - mode surface; the linear sliding - mode surface is the integral of the voltage error and the weighted sum of the voltage error.

[0126] It should be noted here that the linear controller is a PI controller or a PID controller.

[0127] In some other embodiments, a DC - DC converter is further provided, which includes a power switch tube, a phase - shift controller, a terminal sliding - mode controller for the outer loop, and a linear controller for the inner loop;

[0128] The terminal sliding - mode controller is used to calculate the voltage error between the reference voltage and the output voltage of the DC - DC converter, and generate a first control signal as the reference current of the linear controller according to the relationship among a preset fractional - order nonlinear sliding - mode surface, a linear sliding - mode surface, and the voltage error, so as to form a voltage loop;

[0129] The linear controller is configured to generate a second control signal based on the output current of the DC-DC converter and the reference current, and use it as the external phase shift ratio signal of the DC-DC converter to form a current loop.

[0130] The phase shift controller is configured to utilize the relationship between the preset internal phase shift ratio and the external phase shift ratio within the phase shift controller to obtain the internal phase shift ratio signal of the DC-DC converter, thereby implementing the dual-loop non-linear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage, and the output current of the DC-DC converter.

[0131] The topology of the DC-DC converter in this embodiment is a full-bridge converter, such as a dual active bridge (DAB) converter, a dual-bridge series resonant converter (DBSRC), etc.

[0132] The above-mentioned DC-DC converter in the embodiment can be applied to an electric vehicle charging station. For example, an electric vehicle charging station includes a bidirectional isolated AC-DC input stage, the DC-DC converter of the present invention, and an intermediate battery storage system. Among them, the intermediate battery storage system can provide the energy required for ultra-fast charging, avoid power pulsation of the power grid, and can also be used in smart grid applications. The energy capacity of the intermediate storage battery is higher. During the ultra-fast charging process of the vehicle battery, the maximum discharge current of each battery in the intermediate storage battery does not exceed 3C-4C, which can achieve the long life of the fixed battery. The AC-DC input stage can be composed of a T-type inverter and a DC-DC converter system to provide isolation and balance the voltage of the fixed battery.

[0133] In other embodiments, when the DC-DC converter is also a bidirectional DC-DC converter, it can also be applied to a power generation system. Taking a solar photovoltaic power generation system as an example, when there is sufficient sunlight, the photovoltaic panels will generate a large amount of electrical energy. However, due to the changes in light intensity and sunshine time, the output power of the photovoltaic panels will also fluctuate. To solve this problem, a bidirectional DC-DC converter can be combined with an energy storage device (such as a storage battery) to achieve the storage and regulation of electrical energy. When the output power of the photovoltaic panels is greater than the load demand, the excess electrical energy will be stored in the storage battery; when the output power of the photovoltaic panels is insufficient, the storage battery will supply power to the load through the bidirectional DC-DC converter. This process of energy storage and release not only ensures the stable operation of the power system but also improves the energy utilization efficiency.

[0134] In other embodiments, when the DC-DC converter is also a bidirectional DC-DC converter, it can also be applied to energy storage systems. The bidirectional DC-DC converter controls the magnitudes of the output voltage and current by adjusting the duty cycle of the switching device (i.e., the ratio of the conduction time of the switching device to the cycle time). In the charging mode, the converter converts the high-voltage direct current of the power grid into the low-voltage direct current suitable for charging the energy storage device; in the discharging mode, the converter converts the low-voltage direct current of the energy storage device into the high-voltage direct current required by the power grid. This flexible energy conversion method enables the bidirectional DC-DC converter to have broad application prospects in energy storage systems. In distributed energy systems, the bidirectional DC-DC converter can achieve the complementarity and optimized utilization between different energy forms. For example, in a microgrid, the converter can achieve the integration and efficient utilization of various renewable energy sources such as solar energy and wind energy; in an intelligent building, the converter can achieve the energy exchange and optimized management between solar power generation, energy storage devices and the power grid. In an electric vehicle, the bidirectional DC-DC converter can achieve the energy exchange between the power battery and the power grid, providing stable power support for the electric vehicle. At the same time, during the regenerative braking process, the converter can also recover the braking energy and improve the energy utilization efficiency.

[0135] It should be noted here that in other embodiments, the converter of the present invention can also be applied to other electronic devices for electric energy conversion according to the needs of those skilled in the art.

[0136] The above are only the preferred embodiments of the present invention and are not intended to limit the present invention. For those skilled in the art, the present invention can have various changes and modifications. Any modification, equivalent replacement, improvement, etc. made within the spirit and principle of the present invention shall be included within the protection scope of the present invention.

Claims

1. A nonlinear robust control method for a DC-DC converter, characterized in that: include: The output voltage of the DC-DC converter is obtained and fed back to the terminal sliding mode controller, and the terminal sliding mode controller calculates the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generates a first control signal as a reference current of the linear controller according to the relationship between the preset fractional-order nonlinear sliding surface, the linear sliding surface and the voltage error, so as to form a voltage loop; The output current of the DC-DC converter is obtained and fed back to the linear controller, and the linear controller generates a second control signal in combination with the reference current and acts on the set power switch tube of the DC-DC converter to form a current loop, thereby finally realizing the dual-loop nonlinear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage of the DC-DC converter and the output current.

2. The control method of the DC-DC converter according to claim 1, characterized in that: The relationship between the fractional-order nonlinear sliding surface, the linear sliding surface and the voltage error is: The fractional-order nonlinear sliding surface is a weighted sum of the fractional-order integral of the linear sliding surface and the linear sliding surface; the linear sliding surface is a weighted sum of the integral of the voltage error and the voltage error.

3. A DC-DC converter, characterized in that: include: Power switch tube, terminal sliding mode controller of the outer loop and linear controller of the inner loop; The terminal sliding mode controller is used to calculate the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generate a first control signal as a reference current of the linear controller according to the relationship between the preset fractional-order nonlinear sliding mode surface, the linear sliding mode surface and the voltage error, so as to form a voltage loop; The linear controller is used to generate a second control signal based on the output current of the DC-DC converter and the reference current and act on the set power switch tube of the DC-DC converter to form a current loop, and finally realize the DC-DC converter dual-loop nonlinear robust control based on the coordination of the reference voltage, the output voltage of the DC-DC converter and the output current.

4. The DC-DC converter according to claim 3, characterized in that: The relationship between the fractional-order nonlinear sliding surface, the linear sliding surface and the voltage error is: The fractional-order nonlinear sliding surface is a weighted sum of the fractional-order integral of the linear sliding surface and the linear sliding surface; the linear sliding surface is a weighted sum of the integral of the voltage error and the voltage error.

5. A control method for a DC-DC converter, characterized in that: Applicable to a dual-bridge DC-DC converter, the control method of the DC-DC converter comprises: The output voltage of the DC-DC converter is obtained and fed back to the terminal sliding mode controller, and the terminal sliding mode controller calculates the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generates a first control signal as a reference current of the linear controller according to the relationship between the preset fractional-order nonlinear sliding surface, the linear sliding surface and the voltage error, so as to form a voltage loop; The output current of the DC-DC converter is obtained and fed back to the linear controller, and the linear controller generates a reference current and serves as an outward shift comparison signal of the converter to form a current loop; By utilizing the relationship between the inner phase shift ratio and the outer phase shift ratio preset in the phase shift controller, the inner phase shift ratio signal of the DC-DC converter is obtained, thereby realizing the dual-loop nonlinear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage of the DC-DC converter and the output current.

6. The control method of the DC-DC converter according to claim 5, characterized in that: The relationship between the fractional-order nonlinear sliding surface, the linear sliding surface and the voltage error is: The fractional-order nonlinear sliding surface is a weighted sum of the fractional-order integral of the linear sliding surface and the linear sliding surface; the linear sliding surface is a weighted sum of the integral of the voltage error and the voltage error.

7. The control method of the DC-DC converter according to claim 5, characterized in that: The phase-shift controller is an extended phase-shift controller or a dual phase-shift controller.

8. A DC-DC converter, characterized in that: It includes a power switch tube, a phase shift controller, an outer loop terminal sliding mode controller and an inner loop linear controller; The terminal sliding mode controller is used to calculate the voltage error between the reference voltage and the output voltage of the DC-DC converter, and generate a first control signal as a reference current of the linear controller according to the relationship between the preset fractional-order nonlinear sliding mode surface, the linear sliding mode surface and the voltage error, so as to form a voltage loop; The linear controller is used to generate a second control signal based on the output current of the DC-DC converter and the reference current and use the second control signal as an outward shift comparison signal of the DC-DC converter to form a current loop; The phase-shift controller is used to obtain the inner phase shift signal of the DC-DC converter by utilizing the relationship between the inner phase shift ratio and the outer phase shift ratio preset in the phase-shift controller, thereby realizing the dual-loop nonlinear robust control of the DC-DC converter based on the coordination of the reference voltage, the output voltage of the DC-DC converter and the output current.

9. The DC-DC converter according to claim 8, characterized in that: The relationship between the fractional-order nonlinear sliding surface, the linear sliding surface and the voltage error is: The fractional-order nonlinear sliding surface is a weighted sum of the fractional-order integral of the linear sliding surface and the linear sliding surface; the linear sliding surface is a weighted sum of the integral of the voltage error and the voltage error.

10. A DC-DC converter according to claim 3 or 8, characterized in that: The DC-DC converter is coupled to a power generation system, an energy storage system or an electric vehicle power supply.

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