Hybrid frequency conversion phase shift modulation method and device for integrated power decoupling of MBSRC

Abstract

The invention discloses a hybrid frequency conversion phase shift modulation method and device for integrated power decoupling of MBSRC, and the method comprises the steps: constructing an equivalent two-port converter, and determining the resonant cavity characteristic impedance of a resonant cavity between two ports; determining a current transient value, transmission power and circulation power of the resonant cavity between the two ports according to the resonant cavity characteristic impedance so as to determine a total resonant cavity current transient value, total transmission power and total circulation power at the port switching moment; determining a soft switching condition according to the total resonant cavity current transient value at the port switching moment; determining the optimal frequency ratio of the MBSRC by taking the minimum port total circulation power as a target; constructing a multi-dimensional nonlinear power equation set based on the total transmission power of each port; and fixing the optimal frequency ratio, and iteratively solving the multi-dimensional nonlinear power equation set to obtain the phase shift ratio of each port so as to realize the target power distribution of each port. According to the invention, wide-range ZVS (Zero Voltage Suppressor) and circulating current power minimization are realized, and meanwhile, power decoupling in MBSRC is ensured.

Description

Technical Field

[0001] This invention relates to the field of multi-bridge series resonant converter technology, and in particular to a hybrid frequency conversion phase-shift modulation method and apparatus for integrated power decoupling in MBSRC. Background Technology

[0002] Electric vehicle (EV) charging systems based on two-stage power electronic transformers (PETs) offer significant advantages over traditional systems in terms of efficiency, control flexibility, and cost-effectiveness. The isolation converter is a key component in two-stage PETs for achieving high efficiency, high power density, and high electrical insulation. Among various converters, the multi-port converter (MPC) has become the preferred choice for isolation converters in two-stage PETs due to its high efficiency, high power density, and flexible power distribution capabilities.

[0003] However, the wide output voltage range and dynamic load characteristics of electric vehicle batteries pose substantial challenges to the operation of MPCs, such as wide-range zero-voltage switching (ZVS), minimum circulating current power, and power decoupling. For MPCs used in PET-based two-stage multi-port electric vehicle charging stations, wide-range ZVS, minimum circulating current power, and power decoupling are required. However, current research has not yet simultaneously met all three requirements. Summary of the Invention

[0004] This invention aims to at least partially address the technical problems in related technologies. Therefore, a first objective of this invention is to provide a hybrid frequency conversion phase-shift modulation method for integrated power decoupling in MBSRC, which achieves wide-range ZVS operation and minimizes circulating current power while ensuring power decoupling in MBSRC and improving flexible power allocation capabilities.

[0005] A second objective of the present invention is to provide a control device.

[0006] To achieve the above objectives, the present invention is implemented through the following technical solution:

[0007] A hybrid frequency conversion phase-shift modulation method for integrated power decoupling in MBSRC includes:

[0008] An equivalent two-port converter is constructed based on two ports in the MBSRC, and the equivalent inductance and equivalent capacitance of the resonant cavity between the two ports and the characteristic impedance of the resonant cavity are determined based on the equivalent two-port converter. The MBSRC is a multi-port active bridge series resonant converter.

[0009] The transient current, transmission power, and circulating power of the resonant cavity between the two ports are determined based on the characteristic impedance of the resonant cavity, so as to determine the total transient current of the resonant cavity at the phase-leading and phase-lagging switching times of each port, as well as the total transmission power and total circulating power of each port.

[0010] The soft-switching conditions of MBSRC are determined based on the transient values ​​of the total resonant cavity current at two different times for each port.

[0011] The optimal frequency ratio of MBSRC is determined with the goal of minimizing the total circulating current power at each port;

[0012] A multidimensional nonlinear power equation system is constructed based on the total transmission power of each port, including multiple shift ratios corresponding to each port.

[0013] By fixing the optimal frequency ratio, the multidimensional nonlinear power equations are iteratively solved to obtain the shift ratio of each port, so as to achieve the target power allocation of each port according to the shift ratio of each port.

[0014] In one possible implementation, determining the equivalent inductance and equivalent capacitance of the resonant cavity between the two ports includes:

[0015] Determine the turns ratio of the multi-winding transformer between the two ports;

[0016] Construct the star-equivalent circuit and delta-equivalent circuit of MBSRC. In the star-equivalent circuit, calculate the inductance and capacitance values ​​from one port to the other port based on the turns ratio of the multi-winding transformer between the two ports. In the delta-equivalent circuit, calculate the equivalent inductance and equivalent capacitance of the resonant cavity between the two ports based on the inductance and capacitance values ​​calculated in the star-equivalent circuit.

[0017] In one possible implementation, the transient current value of the resonant cavity between the two ports is determined based on the characteristic impedance of the resonant cavity, in order to determine the total transient current value of the resonant cavity at the phase-leading switching moment and the phase-lag switching moment of each port, including:

[0018] The transient values ​​of the resonant cavity current between the two ports are determined based on the transient values ​​of the resonant cavity current at the time of phase-leading switching and phase-lag switching, respectively.

[0019] Calculate the sum of the transient values ​​of the resonant cavity current at the phase-leading switching moment between the port that is the phase-leading port and the other ports in the MBSRC, and the sum of the transient values ​​of the resonant cavity current at the phase-lag switching moment, to obtain the total transient value of the resonant cavity current at the phase-leading switching moment and the phase-lag switching moment of the port.

[0020] In MBSRC, the total transient value of the resonant cavity current at each port is calculated according to the calculation method of the port. The phase lead switching time and phase lag switching time of each port are calculated.

[0021] In one possible implementation, determining the total transmission power and total circulating current power of each port includes:

[0022] The transmission power and circulating power between the two ports are obtained by multiplying and integrating the transient current value of the resonant cavity between the two ports and the AC voltage of the two ports that are phase leading ports.

[0023] Calculate the transmission power and circulating power between the port that is the phase-leading port and the other ports in the MBSRC, and sum them to obtain the total transmission power and total circulating power of the port.

[0024] In MBSRC, the total transmission power and total circulating power of each port are calculated according to the calculation method described above.

[0025] In one possible implementation, the soft-switching condition for MBSRC is:

[0026] The phase of the port leads the transient value of the total resonant cavity current at the switching moment to less than 0;

[0027] The transient value of the total resonant cavity current at the port phase lags the switching moment and is greater than 0.

[0028] In one possible implementation, the method further includes: while constraining the soft-switching conditions of the MBSRC, controlling the characteristic impedance of the resonant cavity to be less than or equal to the minimum characteristic impedance required by the system, wherein the minimum characteristic impedance required by the system is the characteristic impedance of the resonant cavity calculated under the condition that the maximum power at the port is equal to the rated power.

[0029] In one possible implementation, the frequency ratio is the ratio of the switching frequency of the MBSRC to the resonant frequency of the resonant cavity.

[0030] In one possible implementation, the optimal frequency ratio is fixed, and the multidimensional nonlinear power equations are solved iteratively, including:

[0031] Determine the Jacobian matrix of the power vector, which includes the total transmitted power at each port, in the multidimensional nonlinear power equation set;

[0032] The multidimensional nonlinear power equations are solved iteratively using the following formulas until convergence:

[0033] D k+1 =D k -[F'(D k )] -1 [F(D k )-P rated ]

[0034] Among them, D k D k+1 F(D) is the shift ratio vector that includes all port shift ratios at the k-th and k+1-th iterations. k ) is the shift ratio vector Dk The corresponding power vector includes the total transmission power of each port; F'(D k ) is F(D k The Jacobian matrix of F'(D) k )] -1 Let F'(D) be the Jacobian matrix. k The inverse matrix of P) rated It is a vector that includes the target rated power values ​​for each port.

[0035] In one possible implementation, the method further includes:

[0036] The shift ratio of each port obtained by iterative solution is mapped to the phase offset of the H-bridge at each port. At the same time, the optimal frequency ratio is converted into the actual switching frequency so that the modulation signal can be obtained according to the mapped phase offset and the actual switching frequency to drive the switching devices in MBSRC.

[0037] To achieve the above objectives, a second aspect of the present invention provides a control device, including a memory, a processor, and a microcontroller program stored in the memory and executable on the processor. When the processor executes the microcontroller program, it implements the hybrid frequency conversion phase shift modulation method for integrated power decoupling for MBSRC described above.

[0038] This invention has at least the following technical effects:

[0039] This invention provides a hybrid frequency conversion phase-shift modulation method and apparatus for integrated power decoupling of MBSRC. Specifically, the method first determines the transient current value of the resonant cavity between ports, then determines the total transient current value of the resonant cavity at the port's phase-leading and phase-lag switching moments, thereby determining the soft-switching conditions of the MBSRC and achieving wide-range zero-voltage switching (ZVS) of the switching device. Simultaneously, the optimal frequency ratio of the MBSRC is determined with the goal of minimizing the total circulating current power at each port. Then, based on the total transmission power of each port, a multidimensional nonlinear power equation system is constructed, including multiple shift ratios corresponding to each port. The optimal frequency ratio is fixed, and the multidimensional nonlinear power equation system is iteratively solved to obtain the shift ratio of each port, thus minimizing the circulating current power and achieving power decoupling. Therefore, this invention, by combining frequency conversion control with phase-shift modulation, effectively solves the challenges posed by the wide output voltage range and dynamic load conditions of electric vehicle batteries, achieving wide-range zero-voltage switching, minimizing circulating current power, achieving power decoupling, and improving flexible power allocation capabilities.

[0040] Additional aspects and advantages of the invention will be set forth in part in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. Attached Figure Description

[0041] Figure 1 This is a schematic diagram of a six-port MBSRC structure based on a multi-winding transformer according to an embodiment of the present invention.

[0042] Figure 2 This is a flowchart of a hybrid frequency conversion phase-shift modulation method for integrated power decoupling in MBSRC, according to an embodiment of the present invention.

[0043] Figures 3(a) to 3(b) These are the Y-shaped equivalent circuit diagram and the triangular equivalent circuit diagram of the MBSRC according to an embodiment of the present invention.

[0044] Figure 4 This is a schematic diagram of an equivalent two-port converter according to an embodiment of the present invention.

[0045] Figure 5 This is a key waveform diagram of the power transmission between port i and port j in an embodiment of the present invention.

[0046] Figure 6 This is a schematic diagram of the ZVS boundary under different working conditions according to an embodiment of the present invention.

[0047] Figure 7 This is a schematic diagram showing the minimum effective value of the resonant cavity current under different operating conditions with symmetrical load according to an embodiment of the present invention.

[0048] Figure 8 This is a schematic diagram of the modulation strategy for MBSRC according to an embodiment of the present invention.

[0049] Figure 9 This is a schematic diagram of the efficiency of MBSRC at 16% to 100% of rated power measured under different output voltages according to an embodiment of the present invention.

[0050] Figure 10 This is a schematic diagram of the measurement results of the power meter under peak efficiency and full-load efficiency according to an embodiment of the present invention.

[0051] Figure 11 This is a ZVS waveform diagram of MBSRC according to an embodiment of the present invention.

[0052] Figure 12 This is a schematic diagram of the measured efficiency of MBSRC under constant frequency and under asymmetrical load conditions according to an embodiment of the present invention.

[0053] Figure 13 This is a schematic diagram of the transient performance of port voltage and current under load changes at port 6 according to an embodiment of the present invention.

[0054] Figure 14 This is a schematic diagram of MBSRC efficiency measured under asymmetrical load conditions from 16% to 75% according to an embodiment of the present invention. Detailed Implementation

[0055] The following describes this embodiment in detail. Examples of the embodiments are shown in the accompanying drawings, wherein the same or similar reference numerals denote the same or similar elements or elements having the same or similar functions throughout. The embodiments described below with reference to the accompanying drawings are exemplary and intended to explain the invention, and should not be construed as limiting the invention.

[0056] Before describing the hybrid frequency conversion phase-shift modulation method for integrated power decoupling in MBSRC (Multiple Active Bridge Series Resonant Converter) of this embodiment, the MBSRC structure of this embodiment will be briefly described.

[0057] It should be noted that the hybrid frequency conversion phase-shift modulation method and apparatus for integrated power decoupling in MBSRC of this embodiment can be applied to an n-port MBSRC structure. The following description only uses a six-port MBSRC structure. It can be understood that n can be greater than 6, that is, the number of ports can be expanded to multiple ports greater than 6.

[0058] Figure 1 This is a schematic diagram of a six-port MBSRC structure based on a multi-winding transformer, according to an embodiment of the present invention. Figure 1 As shown, this six-port MBSRC consists of six H-bridges, one multi-winding transformer (MWT), six resonant inductors, and six resonant capacitors. Each H-bridge is composed of four SiC MOSFETs (silicon carbide metal-oxide-semiconductor transistors). The DC ports of the six ports are named Port#1, ..., Port#6. Figure 1 The DC voltage and current at port i are defined as V. i and I i The resonant inductance and resonant capacitance are defined as L, ... ri and C ri The port AC voltage and AC current of a multi-winding transformer are defined as v0 and v1, respectively. i (t) and i i (t). The turns ratio of a multi-winding transformer at port j to port i is defined as k. ij AC voltage v between ports i and j of a multi-winding transformer i (t) and v j The shift ratio of (t) is defined as D ij The DC voltage gain at port i and port j is defined as M. ij =k ij ·v j (t) / v i (t); The resonant frequency of the MBSRC cavity is defined as f r The switching frequency is defined as f s The ratio, i.e., the frequency ratio F, is F = f s / f r .

[0059] The hybrid frequency conversion phase-shift modulation method and apparatus for integrated power decoupling of MBSRC according to this embodiment are described below with reference to the accompanying drawings.

[0060] Figure 2 This is a flowchart of a hybrid frequency conversion phase-shift modulation method for integrated power decoupling in MBSRC, according to an embodiment of the present invention. Figure 2 As shown, the method includes:

[0061] Step S101: Construct an equivalent two-port converter based on the two ports in the MBSRC, and determine the equivalent inductance and equivalent capacitance of the resonant cavity between the two ports and the characteristic impedance of the resonant cavity based on the equivalent two-port converter, wherein the MBSRC is a multi-port active bridge series resonant converter.

[0062] In one embodiment of the present invention, determining the equivalent inductance and equivalent capacitance of the resonant cavity between two ports includes: determining the turns ratio of the multi-winding transformer between the two ports; constructing a star equivalent circuit and a delta equivalent circuit of the MBSRC; calculating the inductance and capacitance reduction values ​​from one port to the other port based on the turns ratio of the multi-winding transformer between the two ports in the star equivalent circuit, so that the equivalent inductance and equivalent capacitance of the resonant cavity between the two ports can be calculated in the delta equivalent circuit based on the inductance and capacitance reduction values ​​calculated in the star equivalent circuit.

[0063] In a six-port MBSRC, the power characteristic analysis of the ports is relatively complex due to the coupling between them. However, by using virtual branches to decompose a six-port MBSRC into multiple DBSRCs (Dual Active Bridge Series Resonant Converters), the complexity of the port power characteristic analysis can be significantly reduced. Therefore, a virtual branch needs to be constructed between any two ports i and j. The construction of the virtual branch requires the transformation of its impedance parameters to obtain the equivalent inductance and equivalent capacitance between the two ports, and then the characteristic impedance of its resonant cavity can be calculated.

[0064] First, we need to obtain the turns ratio of the multi-winding transformer, and denote the turns ratio between port i and port j as k. ij Therefore, the transformation matrix K of the six-port MBSRC can be obtained:

[0065]

[0066] Multi-winding transformers are typically modeled using Y-type equivalent circuits (i.e., star-type or delta-type equivalent circuits) as shown in Figures 3(a) and 3(b). For Y-type equivalent circuits, the equivalent parameters are readily available. However, this model is not suitable for analyzing power transfer characteristics between ports. For delta-type equivalent circuits, the power transfer characteristics between ports can be clearly analyzed using virtual branches. However, it is difficult to directly extract the equivalent impedance from the delta-type equivalent model of a multi-winding transformer. In this embodiment, the equivalent impedance of the delta-type equivalent circuit can be obtained from the Y-type equivalent circuit.

[0067] In Figure 3(a), the reduced impedance of each port in the Y-type equivalent circuit is denoted as Z. i In the Δ-type equivalent circuit, the virtual impedance between port i and port j is denoted as Z. ij .

[0068] In the Y-type equivalent circuit, the resonant inductance L at port i is... ri Resonant capacitor C ri The inductance and capacitance values ​​after being reduced to port j are denoted as L. ij C ij The reduced parameters are expressed as the inductance reduction matrix L. Y Capacitance reduction matrix C Y , where L ij =L ri ·k ij 2 C ij =C ri / k ij 2 .

[0069]

[0070] In the delta-type equivalent circuit, the equivalent inductance, equivalent capacitance, and equivalent impedance (the characteristic impedance of the resonant cavity, i.e., the aforementioned virtual impedance) between the two ports are denoted as: L ij ′、C ij ′、Z ij And Z ij The equivalent inductance L between the two ports in the delta-type equivalent circuit can be used to... ij ′、Equivalent capacitance C ij ′ is represented as:

[0071]

[0072] Among them, L ii C ii L represents the equivalent inductance and equivalent capacitance values ​​referred from port i to port i in the Y-type equivalent circuit, respectively. in ′、C in′ represent the equivalent inductance and equivalent capacitance values ​​between port i and port n in the Δ-type equivalent circuit, respectively.

[0073] Step S102: Determine the transient current value, transmission power and circulating current power of the resonant cavity between the two ports based on the characteristic impedance of the resonant cavity, so as to determine the total transient current value of the resonant cavity at the phase-leading switching time and the phase-lagging switching time of each port, as well as the total transmission power and the total circulating current power of each port.

[0074] In one embodiment of the present invention, determining the transient current value of the resonant cavity between two ports based on the characteristic impedance of the resonant cavity, in order to determine the total transient current value of the resonant cavity between the phase-leading switching time and the phase-lag switching time of each port, includes:

[0075] The transient values ​​of the resonant cavity current at the phase-leading and phase-lag switching times between the two ports are determined based on the transient values ​​of the resonant cavity current at the phase-leading switching times between the port (which is the phase-leading port) and the other ports in the MBSRC. The sum of the transient values ​​of the resonant cavity current at the phase-leading switching times and the sum of the transient values ​​of the resonant cavity current at the phase-lag switching times are calculated to obtain the total transient values ​​of the resonant cavity current at the phase-leading and phase-lag switching times of the port. The total transient values ​​of the resonant cavity current at the phase-leading and phase-lag switching times of each port in the MBSRC are calculated according to the calculation method of the port.

[0076] In one embodiment of the present invention, determining the total transmission power and total circulating power of each port includes: multiplying and integrating the transient current value of the resonant cavity between the two ports and the AC voltage of the port that is the phase-leading port to obtain the transmission power and circulating power between the two ports; calculating the transmission power and circulating power between the port that is the phase-leading port and the other ports in the MBSRC respectively, and summing them respectively to obtain the total transmission power and total circulating power of the port; and calculating the total transmission power and total circulating power of each port in the MBSRC according to the calculation method of the port.

[0077] Specifically, to simplify the analysis process, virtual branches were constructed between multi-port converters. The equivalent two-port converters connecting the two ports via these virtual branches are shown below. Figure 4 As shown.

[0078] ω0 is the operating angular frequency of the resonant cavity. Among them, L ij 'for Figure 4 L in rij C ij 'for Figure 4 C in rij Since ports i and j are arbitrary, we assume Dij If the phase of the AC voltage at port i is greater than or equal to the phase of the AC voltage at port j, then power will flow from port i to port j. Port i is defined as the leading port, and port j as the lagging port. The switching period of the equivalent two-port converter is T. s (T s =1 / f s Due to the actual circuit dead time t DT Generally much smaller than the phase shift angle, i.e., the phase shift ratio, therefore the working state is simplified to a single state, namely T. s / 2>D ij >t DT .

[0079] The mathematical model of an equivalent two-port converter is mainly established using the fundamental wave method and the time-domain method. Compared with the frequency domain analysis method, the time-domain analysis method can establish a more accurate mathematical model and more precisely analyze the power transfer characteristics and soft-switching characteristics of the equivalent two-port converter. Figure 5 The equivalent two-port converter waveform shown is used to model the converter. In the figure, t0 to t8 is a complete cycle denoted as T. s Where t0 is the start time of the equivalent two-port converter's operating cycle, and t8 is the end time of the equivalent two-port converter's operating cycle. s The first half of the cycle is symmetrical to the second half, so only t0 to t4 are analyzed. Before time t0, the switching transistor S at port i... i1 and S i2 Activate, port j's S j2 S j3 Activated, at this time v i (t)=-V i and v j (t)=-V j During the time period t0 to t1, the state of port j remains unchanged, while v... j (t)=-V j S of port i i2 S i3 When the resonant cavity current is turned off, the polarity of the current remains unchanged. i1 (Switching transistor S) i1 (anti-parallel diode) and D i2 (Switching transistor S) i2 The anti-parallel diode (freewheeling) at time t1, S i1 and S i4 Activate, v i (t) is reversed to V i t1 is the moment when the voltage at port i reverses; during the period from t1 to t2, the states of the switches at ports i and j remain unchanged, v i (t) and v j(t) With opposite voltage polarities, the resonant cavity current begins to gradually increase and crosses zero at time t1'; during the time interval t2 to t3, at time t2, S j2 and S j3 When the resonant cavity current is turned off, the polarity of the current remains unchanged. j1 (Switching transistor S) j1 (anti-parallel diode) and D j4 (Switching transistor S) j4 The anti-parallel diode continues to freewheel until time S at time t3. j1 and S j4 Activate, v j (t) is reversed to V j t2 is the time when switch j is turned off, and t3 is the time when the voltage at port j reverses; during the period from t3 to t4, the states of port i and port j remain unchanged, v i (t) and v j (t) Voltages with the same polarity are V i and V j t4 is the end time of the first half of the working cycle of the equivalent two-port converter.

[0080] Based on the above analysis, we can base our analysis on the characteristic impedance Z of the resonant cavity. ij The transient current values ​​i of the resonant cavity between port i and port j are obtained during the time periods t0 to t3 and t3 to t4. L :

[0081]

[0082] Among them, i t0 i t2 The transient values ​​of the resonant cavity current at times t0 and t2 are u and u, respectively. C0_ij V is the capacitor voltage value at the switching state transition time t0. i M is the DC voltage at port i. ij Let i be the DC voltage gain at port i and port j. t0 Used to determine the zero-voltage switching (ZVS) status of the phase-leading port, i t2 Used to determine the ZVS status of the phase lag port. Therefore, i t0 It can also be characterized as the transient value of the resonant cavity current at the moment of phase-leading switching between the two ports, i t2 It can also be characterized as the transient value of the resonant cavity current at the moment of phase lag switching between the two ports.

[0083] According to the symmetry of current, I t0 =-I t4 , among which, I t0 I t4 Let i represent the transient values ​​of the resonant cavity current at times t0 and t4, respectively. The i-value between port i and port j can be obtained by solving formula (5).t0 and i t2 :

[0084]

[0085] The transmission power P between ports i and j can be obtained by integrating the product of the AC voltage and current at the virtual branch ports. ij and circulating power Q ij .

[0086]

[0087] Where Q1 and Q2 represent the integral values ​​of voltage and current when the polarities of the leading port voltage and current are opposite, Q1 indicates that the voltage is positive and the current is negative during the corresponding time period, and Q2 indicates that the polarities of the voltage and current are opposite to those of Q1 during the corresponding time period. i (t) represents the AC terminal voltage of port i at time t, where i L1 (t) represents the transient value of the resonant cavity current during the period from t0 to t3, t'1 represents the negative zero-crossing moment of the resonant cavity current, and t'5 represents the positive zero-crossing moment of the resonant cavity current. Q ji Let t be the circulating current power between port j and port i. Q3 and Q4 represent the integral values ​​of voltage and current when the polarities of the voltage and current at the over-lagging ports are opposite, respectively. Q3 indicates that the voltage is positive and the current is negative during the corresponding time period, and Q4 indicates that the voltage and current polarities are opposite to those of Q3 during the corresponding time period. t6 is the moment when the voltage polarity at port j reverses.

[0088] Since the converter operates in inductive mode, time t'1 can be determined as: t'1 > t0. The circulating current power at the phase-lagging port needs to be expressed as:

[0089]

[0090] Because there is a phase difference between the AC voltages at the ports, the transient value of the total resonant cavity current i at the phase-leading switching moment of port i, which is a phase-leading port, can be determined according to the timing sequence. t0_i The transient value of the total resonant cavity current i at the phase lag switching moment t2_i as follows:

[0091]

[0092] Among them, i t0_i,j Let i be the transient value of the resonant cavity current at the moment of phase lead switching between port i and the other ports j in the MBSRC. t2_i,j The transient value of the resonant cavity current at the phase lag switching moment between port i and the other ports j in the MBSRC.

[0093] For MBSRC, the total transmission power and total circulating current power of port i can be obtained by superimposing multiple ports:

[0094]

[0095]

[0096] Among them, P i Q i P represents the total transmission power and total circulating current power of port i, respectively. ij Q ij These represent the transmission power and circulating power between port i and the remaining ports j in the MBSRC, respectively.

[0097] It should be noted that formula (6) describes the current value at the switching moment of MBSRC, which will be used for the analysis of the soft switching state of the converter below; formula (11) describes the transmission power of port i in MBSRC, which will be used for the iterative power decoupling below; formula (12) describes the circulating current power of port i in MBSRC, which is the prerequisite for the dynamic frequency adjustment based on minimizing the circulating current power below.

[0098] Step S103: Determine the soft-switching conditions of MBSRC based on the transient values ​​of the total resonant cavity current at the two times for each port.

[0099] The soft-switching conditions for MBSRC are: the transient value of the total resonant cavity current at the phase-leading switching moment of the port is less than 0; and the transient value of the total resonant cavity current at the phase-lag switching moment of the port is greater than 0.

[0100] In MBSRC, the range of ZVS using only phase-shift modulation is limited. As can be seen from formulas (6) and (10), the realization of soft switching of power devices is directly related to the resonant cavity parameters, frequency ratio, and phase shift ratio. Therefore, the soft switching range of the converter can be expanded by introducing frequency conversion control and designing appropriate resonant cavity parameters.

[0101] The transient current value at the switching moment is expressed as shown in formula (6). Both current values ​​are related to the characteristic impedance Z of the resonant cavity. ij Frequency ratio F, shift ratio D ij The function. According to formula (10), the soft-switching condition for MBSRC is expressed as follows:

[0102]

[0103] Among them, i leading i lagging These are the transient current values ​​at the phase-leading and phase-lagging switching moments of port i, respectively, f(Z) ij (F, D1, ..., D6) represents the transient current function, where D1 to D6 are the shift ratios of ports 1 to 6, respectively. Where i leadingThis determines the ZVS status of the switching transistor in the phase-leading port, i lagging This determines the ZVS status of the switching transistor in the load port.

[0104] To simplify the analysis, we will take virtual ports i and j as examples, and ignore all losses. The AC voltage shift ratio of the ports for both is D. The ZVS operating region can be calculated as follows: Figure 6 As shown.

[0105] exist Figure 6 When the output voltage changes from 320V to 800V, the characteristic impedance Z of the resonant cavity... rij That is, the aforementioned Z ij The range is from 20Ω to 40Ω. For example... Figure 6 As shown in (g)~6(i), when the voltage gain M ij Below the transformer turns ratio k ij (M ij =0.51), ZVS is lost in the power device at port j. Conversely, as Figure 6 As shown in (a)–6(c), when M ij More than k ij At that time, port i experienced a ZVS loss. Furthermore, Figure 6 (a) Figure 6 (d) and 6(g) show that the ZVS region expands as the characteristic impedance of the resonant cavity increases. This trend is evident in... Figure 6 Similar conclusions were also reached in (b), 6(e), 6(h), and 6(c), 6(f), and 6(i). Furthermore, Figure 6 (a)–6(c) show that increasing the characteristic impedance of the resonant cavity leads to an expansion of the ZVS region. The characteristic impedance of the resonant cavity is optimized to 40Ω to ensure maximum power transfer capability at extreme output voltages (V). j Under light load conditions (approximately 25% of rated power) of 320V or 800V, the ZVS loss problem becomes prominent.

[0106] For MBSRCs used in PET-based two-stage multi-port electric vehicle charging stations, their output ports are typically connected to electric vehicle batteries with different voltages and charging powers. Output ports with larger voltage gain deviations from the transformer turns ratio (turns ratio) and lower output power ports have narrower ZVS ranges. To achieve ZVS operation over a wide voltage and load range, the design of the resonant cavity characteristic impedance is crucial. Considering maximum power transfer capability, the standard for resonant cavity characteristic impedance design is expressed as:

[0107]

[0108] Among them, P imax P represents the maximum output power of port i.rating This indicates the rated output power of port i.

[0109] Therefore, while constraining the soft-switching conditions of the MBSRC, the characteristic impedance of the resonant cavity is also controlled to be less than or equal to the minimum characteristic impedance required by the system. The minimum characteristic impedance required by the system is the characteristic impedance of the resonant cavity calculated under the condition that the maximum output power at the port equals the rated power. The converter needs to provide rated output power within a specified output voltage range. For example... Figure 6 As shown in (c), 6(f), and 6(i), when the frequency ratio F is in the range of 1.2 to 2, the characteristic impedance of the resonant cavity needs to be maintained at around 40Ω.

[0110] When the characteristic impedance of the resonant cavity is small, the amplitude of the resonant cavity current is large, which can provide enough energy to discharge the junction capacitance of the switching transistor, thereby widening the operating range of ZVS.

[0111] Step S104: Determine the optimal frequency ratio of MBSRC with the goal of minimizing the total circulating current power at each port.

[0112] Step S105: Construct a multidimensional nonlinear power equation system based on the total transmission power of each port, including multiple shift ratios corresponding to each port.

[0113] Step S106: Fix the optimal frequency ratio, iteratively solve the multidimensional nonlinear power equation system, and obtain the shift ratio of each port so as to achieve the target power allocation of each port according to the shift ratio of each port.

[0114] Specifically, for MBSRC applied to multi-port electric vehicle charging stations based on two-stage power electronic transformers, its output ports also need to be used as chargers for electric vehicles; therefore, power decoupling is required between its output ports. Due to the presence of frequency conversion modulation, the frequency ratio, i.e., variable F, needs to be increased. This embodiment proposes an improved method based on minimizing circulating current power. The value of F is first determined directly by minimizing the circulating current power, thereby simplifying the system equations to variables containing only the shift ratio variable D = [D1, D2, ..., D]. n Then iterative calculations are performed.

[0115] A six-port MBSRC has six shift ratios. Defining each shift ratio requires a parameter phase. Taking port 1 as the reference phase (D1 = 0), the shift ratios of the remaining ports are defined as D... i (i = 2, 3, 4, 5, 6), ignoring all losses, the transmission power of port 1 can be obtained based on the power balance of all port inputs and outputs:

[0116]

[0117] For port i (i = 1, 2, 3, 4, 5, 6), its transmission power can be expressed as a system of equations with 7 variables: P i =P i (F,D i1 D i2 ,…,D i6 ), where D i1 =D i -D1;D i2 D i6 Similarly. Since D1 = 0, there are 6 independent variables in the system of equations, so this 6-dimensional system of equations can be described as:

[0118]

[0119] Among them, P 2i (F,D2,D3,D4,D5,D6) represent the transmission power between port 2 and other ports i, P 6i (F,D2,D3,D4,D5,D6) represents the transmission power between port 6 and other ports i, and P2 and P6 represent the total transmission power of ports 2 and 6, respectively.

[0120] The power transfer characteristics of the converter can be fully described by a six-variable nonlinear equation system consisting of five equations (P2 to P6), in which the circulating power also exhibits similar characteristics. Compared to the decoupling control strategy of the MAB (Multiple Active Bridge) converter, the introduction of the frequency ratio F causes this equation system to exhibit multiple solutions, making it impossible to solve directly using traditional numerical methods. It can be understood that the circulating power is related to the frequency ratio F and the shift ratio D. i The function (i = 2, 3, ..., 6) and the dynamic frequency adjustment method of MBSRC can be described as: under the premise of satisfying the output power, finding the frequency ratio that minimizes the circulating current power.

[0121] To simplify the analysis, we will now base our approach on the following: Figure 7 The two-port virtual branch shown is used to analyze the circulating current power of the converter, with an input voltage Vin = 550V, an output voltage Vout = 600V, a rated output power Praating = 20kW, and a resonant cavity characteristic impedance Z. ij Taking 35 as an example, we analyze the circulating power characteristics. The thermographic icon represents the magnitude of the circulating power. It can be seen that under the condition of ZVS, there is a minimum circulating operating point for each output power, which is the optimal operating point.

[0122] like Figure 7As shown, under the rated power output condition of the converter, there exists an operating point in the combination of shift ratio variable and frequency parameter that minimizes the circulating current power. When ports 4, 5, and 6 are in equal power condition, the output voltage and power of each port have consistent characteristics, and the value of the frequency ratio F can be obtained through the analysis method of the two-port virtual branch mentioned above. However, in the asymmetric power distribution mode, there are extremely complex power flows between the ports. At this time, the frequency converter control can be described as a problem of solving for the minimum circulating current power while satisfying the ZVS condition and achieving the rated output power.

[0123] Based on the above analysis, determining the frequency ratio F transforms the decoupling control into solving a five-dimensional nonlinear equation system. The variables to be solved for this six-port converter are the shift ratios D2, ..., D6. Its mathematical representation can be formally expressed as a five-element nonlinear equation system F(D2, ..., D6), specifically containing the following five independent equations:

[0124]

[0125] Wherein, P2 to P6 are the total transmission power of ports 2 to 6 respectively, and f2(D2,…,D6) to f6(D2,…,D6) are the transmission power functions of P2 to P6 respectively. The five independent equations constitute the multidimensional nonlinear power equation system.

[0126] Such a nonlinear multivariate system can be represented as:

[0127] F(D2,…,D6)=f2(D2,…,D6),…,f6(D2,…,D6) T (18)

[0128] Where F(D2,…,D6) is a power vector that includes the total transmission power of each port.

[0129] For the system of equations (16), the mathematical construction of its Jacobian matrix is ​​as follows:

[0130]

[0131] Where F′(D) is the power vector corresponding to the shift ratio vector D, which includes the total transmission power of each port. Let be the partial derivatives of the total transmission power of port i with respect to ports 2 through 6, i = 2, 3, 4, 5, 6.

[0132] Its solution can be described as follows:

[0133] D k+1 =D k -[F'(D k )] -1 [F(Dk )P rated (20)

[0134] Among them, D k D k+1 F(D) is the shift ratio vector that includes all port shift ratios at the k-th and k+1-th iterations. k ) is the shift ratio vector D k The corresponding power vector includes the total transmission power of each port; F'(D k ) is F(D k The Jacobian matrix of F'(D) k )] -1 Let F'(D) be the Jacobian matrix. k The inverse matrix of P) rated It is a vector that includes the target rated power values ​​for each port.

[0135] Therefore, the optimal frequency ratio can be fixed, and then the multidimensional nonlinear power equations can be solved iteratively according to formula (20) to obtain the shift ratio of each port.

[0136] Based on the above analysis, a proposed modulation scheme for MBSRC can be established, such as... Figure 8 As shown, firstly, the characteristic impedance of the converter's resonant cavity is designed according to predefined parameters to extend the operating range of the ZVS. Then, the frequency ratio F is determined by minimizing the circulating current power. Five PI (proportional-integral) controllers generate reference power values ​​based on output voltage reference values, which are used as inputs for decoupling control. Finally, the shift ratios of all ports are obtained through an iterative algorithm. These shift ratios are then mapped to the phase offsets of the H-bridges at each port, while the optimal frequency ratio is converted to the actual switching frequency. A modulation signal is then generated based on the mapped phase offsets and the actual switching frequency. This modulation signal drives the switching devices in the MBSRC, achieving power decoupling and thus flexible power allocation.

[0137] To verify the effectiveness of the proposed modulation scheme, the efficiency of the MBSRC prototype was tested under the condition that ports 4, 5, and 6 operate at the same output voltage and power. Figure 9 The measured efficiency of the MBSRC is explained at different output voltages and output power ranges from 16% to 100% of the rated power. It can be seen that a peak efficiency of 99.225% can be achieved at an output voltage of 550V and a total output power of 15kW. Furthermore, Figure 10 Screenshots of the power analyzer are provided for three different load port voltages: 320V, 550V, and 800V. These screenshots show the full-load and peak efficiency under each operating condition. Figure 10(a) to 10(c) represent the peak efficiencies corresponding to load port voltages of 320V, 550V, and 800V. Figure 10 (d) to 10(f) represent the full-load efficiency corresponding to load port voltages of 320V, 550V, and 800V. Furthermore, at voltage gain M... ij Within the range of 0.51 to 1.276, the peak efficiency and full-load efficiency of the converter exceed 98% and 97%, respectively.

[0138] It is important to note that, such as Figure 9 As shown in the operating points A, B, and C, the efficiency is low when the MBSRC output voltage is high but the power is low because the power devices on the lead port lose ZVS, and the switching losses also increase significantly. This is because the converter voltage gain M... ij The range of variation is relatively wide, while MBSRC is in M ij Much lower than the ratio k ij At this time, the ZVS of the power device at the hysteresis port will be lost, in M ij Much higher than k ij At this time, power devices with the leading port will lose ZVS. For example... Figure 6 (c) Figure 6 (f) Figure 6 As shown in (i), under the premise of satisfying the maximum transmission power, the characteristic impedance Z of the resonant cavity is... ij Designed to be 40Ω, this is under extreme operating conditions (V j =320V and V j When the voltage is 800V, the ZVS may be lost under light load (approximately 25% or less of the rated power).

[0139] To verify the ZVS status of the converter under the control algorithm, tests were conducted at low voltage and light load (U out1 =U out2 =U out3 =350V, P out1 =P out2 =P out3 =3.3kW), high voltage full load (U out1 =U out2 =U out3 =800V, P out1 =P out2 =P out3 The ZVS condition of the primary and secondary switching devices of the converter was tested at a power consumption of 20kW, where U out1 ~U out3 These are the output voltages of ports 1 through 3, P out1 ~P out3 These represent the output power of ports 1 through 3, respectively. Figure 11 The experimental waveforms of the converter are shown, including AC port current, drive voltage, and transistor voltage drop. Figure 11 The upper left and upper right figures respectively illustrate the soft-switching behavior of the primary-side switching devices under low-voltage light load and high-voltage full load conditions. Figure 11 The lower left and lower right figures illustrate the soft-switching behavior of the secondary-side switching devices under low-voltage light load and high-voltage full load conditions, respectively. Experimental results show that the converter can maintain ZVS (Zero-Voltage Switching) over a wide load and voltage range using the proposed control method.

[0140] In the proposed modulation scheme, a frequency conversion control method is specifically designed to minimize circulating current power. To verify the effectiveness of the proposed modulation scheme in minimizing circulating current power, two typical operating cases are demonstrated using fixed frequency and frequency conversion control. One case is when the output voltage is lower than the input voltage (Vin = 550V, U...). out1 =U out2 =U out3 =400V), another is that the output voltage is higher than the input voltage (Vin=550V, U out1 =U out2 =U out3 =600V). For each case, the efficiency of the MBSRC was measured at two fixed frequency ratios: F = 1.4, and compared with the efficiency of the MBSRC under the proposed variable frequency control. The efficiency comparison between the two cases is as follows: Figure 12 As shown. From Figure 12 It can be seen that, at a fixed frequency, a higher frequency produces better efficiency than F. However, at a fixed frequency ( Figure 12 The efficiency of the (dashed line in the diagram) is always lower than that of the proposed modulation scheme. Figure 12 The efficiency (solid line). Compared with the fixed frequency method, experimental results confirm that the proposed variable frequency control method can minimize the circulating current power of the MBSRC and improve its efficiency across the entire load range.

[0141] To evaluate the power decoupling performance of MBSRC under the proposed modulation scheme, a load step change experiment was conducted under the following conditions: U out1 =U out2 =U out3 =550V, U out4 =U out5 =U out6 =550V, P out4 =P out5 =P out6 =10kW. For example... Figure 13As shown, a step load change is applied to port 6, with the load first decreasing from half load (22A) to light load (3.3V), and then increasing from light load (3.3A) back to half load (22A). During these transitions, the output voltage at port 6 remains almost constant, while the output currents at ports 4 and 5 are unaffected. These experimental results confirm that the proposed modulation scheme can achieve power decoupling control of MBSRC.

[0142] To evaluate the efficiency performance of the MBSRC under unbalanced load and voltage conditions, two typical operating cases were demonstrated. One case involved the same output voltage (U) under different loads. out1 =U out2 =U out3 =550V, U out4 =U out5 =U out6 =550V, P out4 =10kW, P out5 =15kW, P out6 =20kW), another situation is different output voltages (U) under the same load. out1 =U out2 =U out3 =500V, U out4 =400V, U out5 =600V, P out4 =P out5 =P out6 =15kW). The measurement efficiency in these two cases is as follows: Figure 14 As shown, even under conditions of unbalanced load and voltage, the efficiency of the MBSRC can still remain above 97%.

[0143] Furthermore, the present invention also provides a control device, including a memory, a processor, and a computer program stored in the memory and executable on the processor. When the processor executes the computer program, it implements the above-described hybrid frequency conversion phase-shift modulation method for integrated power decoupling of MBSRC.

[0144] In summary, this paper proposes a hybrid frequency conversion phase-shift modulation scheme for integrated power decoupling of MBSRC in a two-level PET multi-port electric vehicle charging station. By combining frequency conversion control with phase-shift modulation, this scheme effectively addresses the challenges posed by the wide output voltage range and dynamic load conditions of electric vehicle batteries. It achieves wide-range zero-voltage switching, minimizes circulating current power, and realizes power decoupling. Simulation results show that the MBSRC can achieve a peak efficiency of 99.225% with an output voltage ranging from 320V to 800V, exceeding 98% efficiency under peak load, and exceeding 97% efficiency under full load conditions, as well as wide-range ZVS and power decoupling.

[0145] The device embodiments described above are merely illustrative. The units described as separate components may or may not be physically separate. The components shown as units may or may not be physical units; that is, they may be located in one place or distributed across multiple network units. Some or all of the modules can be selected to achieve the purpose of this embodiment according to actual needs. Those skilled in the art can understand and implement this without any creative effort.

[0146] Through the above description of the embodiments, those skilled in the art can clearly understand that each embodiment can be implemented by means of software plus necessary general-purpose hardware platforms, and of course, it can also be implemented by hardware. Based on this understanding, the above technical solutions, in essence or the part that contributes to the prior art, can be embodied in the form of a software product. This computer software product can be stored in a computer-readable storage medium, such as ROM / RAM, magnetic disk, optical disk, etc., and includes several instructions to cause a computer device (which may be a personal computer, server, or network device, etc.) to execute the methods described in the various embodiments or some parts of the embodiments.

[0147] Finally, it should be noted that the above embodiments are preferred embodiments of the present invention, but the embodiments of the present invention are not limited to the above embodiments. Any changes, modifications, substitutions, combinations, or simplifications made without departing from the spirit and principle of the present invention should be considered equivalent substitutions and are included within the protection scope of the present invention.

Claims

1. A hybrid frequency conversion phase-shift modulation method for integrated power decoupling in MBSRC, characterized in that, include: An equivalent two-port converter is constructed based on two ports in the MBSRC, and the equivalent inductance and equivalent capacitance of the resonant cavity between the two ports and the characteristic impedance of the resonant cavity are determined based on the equivalent two-port converter. The MBSRC is a multi-port active bridge series resonant converter. The transient current, transmission power, and circulating power of the resonant cavity between the two ports are determined based on the characteristic impedance of the resonant cavity, so as to determine the total transient current of the resonant cavity at the phase-leading and phase-lagging switching times of each port, as well as the total transmission power and total circulating power of each port. The soft-switching conditions of MBSRC are determined based on the transient values ​​of the total resonant cavity current at two different times for each port. The optimal frequency ratio of MBSRC is determined with the goal of minimizing the total circulating current power at each port; A multidimensional nonlinear power equation system is constructed based on the total transmission power of each port, including multiple shift ratios corresponding to each port. By fixing the optimal frequency ratio, the multidimensional nonlinear power equations are iteratively solved to obtain the shift ratio of each port, so as to achieve the target power allocation of each port according to the shift ratio of each port.

2. The method as described in claim 1, characterized in that, Determining the equivalent inductance and equivalent capacitance of the resonant cavity between the two ports includes: Determine the turns ratio of the multi-winding transformer between the two ports; Construct the star-equivalent circuit and delta-equivalent circuit of MBSRC. In the star-equivalent circuit, calculate the inductance and capacitance values ​​from one port to the other port based on the turns ratio of the multi-winding transformer between the two ports. In the delta-equivalent circuit, calculate the equivalent inductance and equivalent capacitance of the resonant cavity between the two ports based on the inductance and capacitance values ​​calculated in the star-equivalent circuit.

3. The method as described in claim 1, characterized in that, The transient current value of the resonant cavity between the two ports is determined based on the characteristic impedance of the resonant cavity, in order to determine the total transient current value of the resonant cavity at the phase-leading and phase-lag switching times of each port, including: The transient values ​​of the resonant cavity current between the two ports are determined based on the transient values ​​of the resonant cavity current at the time of phase-leading switching and phase-lag switching, respectively. Calculate the sum of the transient values ​​of the resonant cavity current at the phase-leading switching moment between the port that is the phase-leading port and the other ports in the MBSRC, and the sum of the transient values ​​of the resonant cavity current at the phase-lag switching moment, to obtain the total transient value of the resonant cavity current at the phase-leading switching moment and the phase-lag switching moment of the port. In MBSRC, the total transient value of the resonant cavity current at each port is calculated according to the calculation method of the port. The phase lead switching time and phase lag switching time of each port are calculated.

4. The method as described in claim 1, characterized in that, Determine the total transmission power and total circulating current power of each port, including: The transmission power and circulating power between the two ports are obtained by multiplying and integrating the transient current value of the resonant cavity between the two ports and the AC voltage of the two ports that are phase leading ports. Calculate the transmission power and circulating power between the port that is the phase-leading port and the other ports in the MBSRC, and sum them to obtain the total transmission power and total circulating power of the port. In MBSRC, the total transmission power and total circulating power of each port are calculated according to the calculation method described above.

5. The method as described in claim 1, characterized in that, The soft-switching condition for MBSRC is: The phase of the port leads the transient value of the total resonant cavity current at the switching moment to less than 0; The transient value of the total resonant cavity current at the port phase lags the switching moment and is greater than 0.

6. The method as described in claim 5, characterized in that, The method further includes: while constraining the soft-switching conditions of the MBSRC, controlling the characteristic impedance of the resonant cavity to be less than or equal to the minimum characteristic impedance required by the system, wherein the minimum characteristic impedance required by the system is the characteristic impedance of the resonant cavity calculated under the condition that the maximum power at the port is equal to the rated power.

7. The method as described in claim 1, characterized in that, The frequency ratio is the ratio of the switching frequency of MBSRC to the resonant frequency of the resonant cavity.

8. The method as described in claim 1, characterized in that, By fixing the optimal frequency ratio, iteratively solving the multidimensional nonlinear power equations includes: Determine the Jacobian matrix of the power vector, which includes the total transmitted power at each port, in the multidimensional nonlinear power equation set; The multidimensional nonlinear power equations are solved iteratively using the following formulas until convergence: D k+1 D k -[F'(D k )] -1 [F(D k )-P rated ] Among them, D k D k+1 F(D) is the shift ratio vector that includes all port shift ratios at the k-th and k+1-th iterations. k ) is the shift ratio vector D k The corresponding power vector includes the total transmission power of each port; F'(D k ) is F(D k The Jacobian matrix of F'(D) k )] - 1 is the Jacobian matrix F'(D) k The inverse matrix of P) rated It is a vector that includes the target rated power values ​​for each port.

9. The method according to any one of claims 1 to 8, characterized in that, The method further includes: The shift ratio of each port obtained by iterative solution is mapped to the phase offset of the H-bridge at each port. At the same time, the optimal frequency ratio is converted into the actual switching frequency so that the modulation signal can be obtained according to the mapped phase offset and the actual switching frequency to drive the switching devices in MBSRC.

10. A control device, comprising a memory, a processor, and a microcontroller program stored in the memory and executable on the processor, characterized in that, When the processor executes the microcontroller program, it implements the method as described in any one of claims 1 to 9.

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