Flyback converter, constant current control method and lighting system

Abstract

The application discloses a flyback converter, a constant-current control method and a lighting system. The flyback converter comprises the following steps: sampling a loop current in a resonant loop to obtain a first sampling signal; obtaining a zero-crossing moment of an excitation current in the resonant loop; turning off a first switch tube when the first sampling signal reaches a reference value, and turning off a second switch tube after the zero-crossing moment of the excitation current in the resonant loop, so as to control the average value of the excitation current to be constant. The application can realize the constant of the average value of the excitation current in each switching cycle based on the loop current in the resonant loop of the primary side part of the converter, and then realize the constant-current control of the output current of the converter. The control method is simple, the cost is low, the circuit volume is small, and the constant-current effect is better.

Description

Technical Field

[0001] This invention relates to the field of switching power supply technology, specifically to a flyback converter, a constant current control method, and a lighting system. Background Technology

[0002] With the increasing trend of global warming and the continuous depletion of Earth's resources in recent years, energy issues have gradually become one of the most pressing concerns. The amount of electricity consumed by buildings each year is enormous, and building lighting is one of the main sources of this energy consumption. Therefore, improving lighting efficiency is a major challenge.

[0003] In recent years, light-emitting diodes (LEDs) have gradually become the mainstream light source due to their high luminous efficiency. Furthermore, due to their unique physical structure and optical and electrical characteristics, LEDs have significant advantages compared to other traditional light sources. The diode driver plays a crucial role in controlling energy efficiency, so system efficiency can be further improved through appropriate modifications and optimizations to the driver circuit. Traditional LED driving systems achieve LED control based on single-stage or multi-stage drivers. A single-stage LED driver is a DC / DC converter with a constant output current, and it can also achieve power factor correction (PFC). A multi-stage system, for example, consists of a PFC circuit and a DC / DC converter. The PFC circuit typically uses a BOOST topology as the pre-stage to provide a stable output voltage (e.g., 400V) as the input to the subsequent stage, while the DC / DC converter acts as the subsequent stage, converting the output voltage provided by the PFC circuit into the required DC voltage. Moreover, due to the unique characteristics of LEDs, the DC / DC converter always needs to include a current loop to achieve constant current control.

[0004] Currently, the main DC / DC converter topologies include LLC resonant converters and asymmetric half-bridge flyback (AHB-Flyback) topologies. For example... Figure 1 The diagram shows a schematic of a lighting system composed of PFC and LLC. In the LLC resonant converter, a resonant inductor is incorporated into the primary winding of the transformer. Compared to the AHB-Flyback system, which primarily utilizes the leakage inductance of the transformer's primary winding, the hardware cost of using an LLC resonant converter is relatively higher. Furthermore, the LLC resonant converter is a third-order system, making its compensation circuit design more difficult and its control more complex. To achieve better loop stability, charge control is required, further complicating the sampling system and increasing the overall system cost.

[0005] The asymmetric half-bridge flyback converter has lower voltage stress, and can utilize the energy of the leakage inductance to realize zero-voltage turn-on of the switch tube, so as to realize high efficiency and obtain popularity. The asymmetric half-bridge flyback converter can also realize constant output current through pulse width modulation (PWM) control. The pulse width modulation is a very effective technology capable of accurately controlling the current in an analog circuit. According to the change of the load, the pulse width modulation controls the difference between the current in the circuit and the reference current in a closed loop feedback, and then adjusts the bias of the gate or base of the switch tube through a series of compensation circuits and drive circuits, so as to change the conduction time of the switch tube in the switching power supply, and change the current in the circuit, that is, the output current can be kept constant under the condition of the change of the working condition through the PWM.

[0006] Most of the constant current closed loop control methods for the asymmetric half-bridge flyback converter directly sample the current in the secondary loop, which requires additional devices such as optocoupler and isolation operational amplifier. These devices have many shortcomings, such as: 1) increasing the cost and the size of the circuit; 2) when the optocoupler is directly used for isolated transmission of analog signals, the nonlinearity of the optocoupler needs to be considered, which greatly increases the difficulty of circuit design; 3) the transmission delay of the optocoupler is large, in order to ensure the accuracy of the turn-on and turn-off of the switch tube, the structure parameters of each path must be consistent, and the delay of each path must be consistent, which also increases the difficulty of circuit design.

[0007] Therefore, it is necessary to provide an improved technical solution to overcome the above technical problems in the prior art. SUMMARY

[0008] In order to solve the above technical problems, the present application provides a flyback converter, a constant current control method and a lighting system, which can realize the constant of the average value of the excitation current in each switch tube cycle based on the loop current in the resonant loop of the primary side of the converter, and further realize the constant current control of the output current of the converter. The control method is simple, the constant current effect is better, and no additional resonant inductor and devices such as optocoupler and isolation operational amplifier are needed when realizing voltage conversion, so that the cost and the size of the circuit can be reduced.

[0009] According to a first aspect of the present application, a constant current control method of a flyback converter is provided, the flyback converter comprising a first switch tube and a second switch tube constituting a half-bridge, a first inductor, a first capacitor and a primary winding in a transformer constituting a resonant loop with the second switch tube, wherein in each switching cycle, the constant current control method comprises:

[0010] sampling the loop current in the resonant loop to obtain a first sampling signal;

[0011] obtaining a zero-crossing time of the excitation current in the resonant loop;

[0012] turning off the first switch tube when the first sampling signal reaches a reference value, and turning off the second switch tube after the zero-crossing time of the excitation current, so as to control the average value of the excitation current to be constant.

[0013] Optionally, the reference value is a preset fixed value.

[0014] Optionally, the reference value is an error signal generated according to the loop current feedback.

[0015] Optionally, turning off the first switch tube when the first sampling signal reaches a reference value comprises:

[0016] charging the second capacitor by the first current source, the second current source and the third current source, so as to obtain the error signal across the second capacitor;

[0017] comparing the first sampling signal and the error signal, and turning off the first switch tube when the first sampling signal reaches the error signal,

[0018] wherein the first current source injects current into the second capacitor, the second current source and the third current source extract current from the second capacitor at different times, and the second current source is controlled by a first feedback signal and the third current source is controlled by a second feedback signal.

[0019] Optionally, the first feedback signal is the first sampling signal; and

[0020] The method for obtaining the second feedback signal comprises:

[0021] sampling and holding the peak value of the first sampling signal before the first switch tube is turned off, to obtain a second sampling signal;

[0022] performing gain processing on the second sampling signal by setting a gain coefficient, to obtain the second feedback signal.

[0023] Optionally, the gain coefficient is 0.5.

[0024] Optionally, the method for determining the zero-crossing time of the excitation current comprises:

[0025] integrating a predetermined parameter of the flyback converter with respect to time during the conduction of the first switch tube to obtain a first integral result;

[0026] integrating the predetermined parameter of the flyback converter with respect to time after the conduction of the second switch tube to obtain a second integral result, and obtaining the zero-crossing time of the excitation current when the second integral result reaches the first integral result.

[0027] wherein the transformer comprises a primary winding, a secondary winding and an auxiliary winding, and the predetermined parameter is a voltage across any winding of the transformer.

[0028] Optionally, the method of determining the zero-crossing time of the excitation current comprises:

[0029] charging a preset capacitor based on a current on the auxiliary winding of the transformer during the first switch tube is turned on;

[0030] discharging the preset capacitor based on the current on the auxiliary winding of the transformer after the second switch tube is turned on, and obtaining the zero-crossing time of the excitation current when a voltage across the preset capacitor is lower than a preset voltage value.

[0031] Optionally, turning off the second switch tube after the zero-crossing time of the excitation current in the resonant circuit comprises:

[0032] turning off the second switch tube after a preset time from the zero-crossing time of the excitation current; or

[0033] turning off the second switch tube when the excitation current reaches a preset negative value.

[0034] Optionally, the constant current control method further comprises:

[0035] turning on the second switch tube after a first dead time when the first switch tube is turned off;

[0036] turning on the first switch tube after a second dead time when the second switch tube is turned off or a drain-source voltage across the first switch tube is lower than a preset voltage value.

[0037] According to a second aspect of the present application, there is provided a flyback converter, comprising: a transformer having a primary winding and a secondary winding;

[0038] a first switch tube and a second switch tube connected in series between an input terminal and a reference ground;

[0039] a first inductor and a first capacitor connected to the primary winding and the second switch tube to form a resonant circuit;

[0040] a current sampling unit connected to the resonant circuit to sample a loop current in the resonant circuit to obtain a first sampling signal;

[0041] a zero-crossing detection unit to obtain a zero-crossing time of an excitation current in the resonant circuit;

[0042] a control circuit, connected with control ends of the first switch tube and the second switch tube, for controlling conduction states of the first switch tube and the second switch tube,

[0043] The control circuit is connected with the current sampling unit and the zero-crossing detection unit, for turning off the first switch tube when the first sampling signal reaches a reference value, and turning off the second switch tube after a zero-crossing moment of the excitation current in the resonant loop, so as to control the average value of the excitation current to be constant.

[0044] Optionally, the control circuit is further for turning on the second switch tube after a first dead time when the first switch tube is turned off; and

[0045] The first switch tube is turned on after a second dead time when the second switch tube is turned off or when a drain-source voltage across the first switch tube is lower than a preset voltage value.

[0046] Optionally, the control circuit comprises:

[0047] a first conduction trigger signal generation module, for generating a conduction trigger signal of the first switch tube based on a drain-source voltage across the first switch tube, or for generating the conduction trigger signal of the first switch tube based on an off trigger signal of the second switch tube and the second dead time;

[0048] a first off trigger signal generation module, for generating an off trigger signal of the first switch tube based on the first sampling signal and the reference value;

[0049] a second conduction trigger signal generation module, for generating a conduction trigger signal of the second switch tube based on the off trigger signal of the first switch tube and the first dead time;

[0050] a second off trigger signal generation module, for generating an off trigger signal of the second switch tube based on the zero-crossing moment of the excitation current.

[0051] Optionally, the control circuit further comprises:

[0052] a first RS flip-flop, a reset end of which is connected with an output end of the first off trigger signal generation module, and a set end of which is connected with an output end of the first conduction trigger signal generation module;

[0053] a first driver, connected between an output end of the first RS flip-flop and a control end of the first switch tube;

[0054] a second RS flip-flop, a reset end of which is connected with an output end of the second off trigger signal generation module, and a set end of which is connected with an output end of the second conduction trigger signal generation module;

[0055] a second driver connected between an output terminal of the second RS flip-flop and a control terminal of the second switch tube.

[0056] Optionally, the reference value is a preset fixed value.

[0057] Optionally, the reference value is an error signal generated according to the loop current feedback.

[0058] Optionally, the first turn-off trigger signal generation module comprises:

[0059] an error signal generation unit configured to generate an error signal according to the first feedback signal and the second feedback signal; and

[0060] a comparison unit having a non-inverting input terminal receiving the error signal, an inverting input terminal receiving the first sampling signal, and an output terminal outputting the turn-off trigger signal of the first switch tube.

[0061] Optionally, the error signal generation unit comprises:

[0062] a first current source and a second capacitor connected between a power supply terminal and a ground terminal, the first current source being configured to inject a current to a middle node of the first current source and the second capacitor;

[0063] a second current source and a first switch connected between the middle node and the ground terminal, the second current source being configured to extract a current from the middle node when the first switch is closed;

[0064] a third current source and a second switch connected between the middle node and the ground terminal, the third current source being configured to extract a current from the middle node when the second switch is closed;

[0065] a selection unit configured to generate first and second selection signals that are mutually inverted according to a control signal of the first switch tube and a zero-crossing moment of the excitation current, the first selection signal being effective to control the first switch to be closed, and the second selection signal being effective to control the second switch to be closed;

[0066] a feedback signal generation unit configured to generate the first and second feedback signals according to the first sampling signal;

[0067] wherein the second current source is a voltage-controlled current source controlled by the first feedback signal, the third current source is a voltage-controlled current source controlled by the second feedback signal, and the first, second and third current sources jointly charge the second capacitor to generate the error signal across the second capacitor.

[0068] Optionally, the feedback signal generation unit comprises:

[0069] a sample-and-hold unit configured to sample and hold a peak value of the first sampling signal before the first switch tube is turned off, to obtain a second sampling signal;

[0070] a voltage gain unit configured to perform gain processing on the second sampling signal according to a gain coefficient, to obtain the second feedback signal,

[0071] wherein the first feedback signal is a first sampling signal obtained by real-time sampling of the current sampling unit.

[0072] Optionally, the gain coefficient is 0.5.

[0073] According to a third aspect of the present application, there is provided a lighting system, comprising: a rectifier circuit configured to rectify an input alternating current power supply to output a first voltage signal;

[0074] a power factor correction circuit connected to the rectifier circuit, configured to perform power factor correction on the first voltage signal and output a second voltage signal; and

[0075] The flyback converter described above is used to implement constant current driving of an LED load.

[0076] The flyback converter, the constant current control method and the lighting system provided by the present application have the following advantages: in each switching period of the converter, the first switch tube is turned off when the loop current in the resonant loop of the primary side reaches a reference value, and the second switch tube is turned off when the excitation current reaches zero, so that the maximum and minimum values of the excitation current when it changes are constant, and the average value of the excitation current in each switching period of the switch tube is constant, i.e. constant current control of the output current of the converter is equivalently achieved. The control method is simple and the constant current effect is better. At the same time, when voltage conversion is implemented, no additional resonant inductor and devices such as optocoupler and isolation operational amplifier are needed, which can reduce the cost and the size of the circuit.

[0077] It should be noted that the general description above and the detailed description below are only exemplary and explanatory, and cannot limit the present application. BRIEF DESCRIPTION OF DRAWINGS

[0078] Figure 1 FIG. 1 shows a structure schematic diagram of a lighting system according to an embodiment of the present application;

[0079] Figure 2 FIG. 1 shows a structure schematic diagram of a lighting system according to an embodiment of the present application;

[0080] Figure 3 FIG. 1 shows a structure schematic diagram of a lighting system according to an embodiment of the present application;

[0081] Figure 4 Fig. 3 shows a structure diagram of a first off trigger signal generating unit according to an embodiment of the present application;

[0082] Figure 5 Fig. 4 shows a timing waveform diagram of each current in a steady state of a flyback converter according to an embodiment of the present application;

[0083] Figure 6 Fig. 5 shows a timing waveform diagram of partial signals in a closed loop current sampling circuit of a flyback converter according to an embodiment of the present application;

[0084] Figure 7 Fig. 6 shows a flow diagram of a constant current control method of a flyback converter according to an embodiment of the present application. DETAILED DESCRIPTION

[0085] In order to make the present application more comprehensively understood, the following will make a more comprehensive description of the present application with reference to the related drawings. The preferred embodiments of the present application are shown in the drawings. However, the present application can be realized in different forms and is not limited to the embodiments described herein. On the contrary, the purpose of providing these embodiments is to make the disclosure of the present application more thorough and comprehensive.

[0086] As shown in Fig. 1, the lighting system in the embodiment of the present application comprises an AC power supply 1, a rectifier circuit 2, a power factor correction circuit 3 and a flyback converter 5. Figure 2

[0087] The rectifier circuit 2 is used to rectify the AC power input by the AC power supply 1, so as to output a first voltage signal to the power factor correction circuit 3. Optionally, the rectifier circuit 20 can be a full-bridge rectifier circuit or a half-bridge rectifier circuit.

[0088] The power factor correction circuit 3 is connected with the rectifier circuit 2, and is used to perform power factor correction on the first voltage signal output by the rectifier circuit 2, and output a second voltage signal. The power factor correction circuit 3 generally adopts a BOOST topology structure and serves as a front stage, so as to provide a stable second voltage signal (such as 400V) based on the rectified first voltage signal as an input of a rear stage circuit such as the flyback converter 5.

[0089] The flyback converter 5 is connected with the LED load, and is used to provide a driving current for the LED load to drive the LED load. In the embodiment, the flyback converter 5 is used to realize constant current driving of the LED load.

[0090] ​In the present application, the flyback converter 5 is for example an asymmetric half-bridge flyback converter, which comprises a transformer 51 comprising a primary winding Np and a secondary winding Ns, a first switch Q1 and a second switch Q2 connected in series between an input terminal and a reference ground and constituting a half-bridge, a first inductor Lk, a first capacitor Cr, a zero-crossing detection unit 52, a current sampling unit 53 and a control circuit 54.

[0091] The drain of the first switch Q1 is connected to the input terminal to receive the second voltage signal, and the gate of the first switch Q1 is connected to the control circuit 54. The drain of the second switch Q2 is connected to the source of the first switch Q1, the source of the second switch Q2 is connected to the reference ground, and the gate of the second switch Q2 is connected to the control circuit 54. The capacitors C11 and C12 are respectively the junction capacitances of the first switch Q1 and the second switch Q2. In the same switching cycle, the first switch Q1 and the second switch Q2 are turned on in time to transfer the second voltage signal from the primary side of the transformer 51 to the secondary side. In a possible embodiment, the first switch Q1 and the second switch Q2 are both NMOS field effect transistors.

[0092] The first inductor Lk, the first capacitor Cr, the primary winding Np and the second switch Q2 form a resonant circuit. One end of the first inductor Lk is connected to the drain of the second switch Q2 through the first capacitor Cr, the other end of the first inductor Lk is connected to the like-named end of the primary winding Np, and the unlike-named end of the primary winding Np is connected to the source of the second switch Q2. In a possible embodiment, the first inductor Lk is the leakage inductance of the primary winding Np, and the first capacitor Cr is the resonant capacitor.

[0093] The secondary side of the flyback converter 5 includes a rectifier diode D1 and an output capacitor Co. The anode of the rectifier diode D1 is connected to the non-identical end of the secondary winding Ns, and the cathode of the rectifier diode D1 is connected to the output of the flyback converter 5. The positive pole of the output capacitor Co is connected to the output of the flyback converter 5, and the negative pole of the output capacitor Co is connected to the reference ground, and the identical end of the secondary winding Ns is also connected to the reference ground. Further, the output of the flyback converter 5 is connected to a load, and the load receives the electrical energy (e.g., voltage and current) converted by the flyback converter 5. In some examples, the electrical energy converted by the flyback converter 5 also passes through a filter before reaching the load. In some examples, the filter is a subcomponent of the flyback converter 5, an external component of the flyback converter 5, and / or a subcomponent of the load. In any case, the load can use the filtered or unfiltered electrical energy from the flyback converter 5 to perform a function. Optionally, the load can include, but is not limited to, a computing device and related components, such as a microprocessor, an electrical component, a circuit, a laptop computer, a desktop computer, a tablet computer, a mobile phone, a battery, a speaker, a lighting unit such as an LED, a component related to an automobile / ship / aerospace / train, a motor, a transformer, or any other type of electrical device and / or circuit that receives voltage or current from a flyback converter.

[0094] The current sampling unit 53 is connected to the resonant loop to sample the loop current (denoted as I Lr ) in the resonant loop to obtain a first sampling signal V RS . In an exemplary embodiment of the present application, the current sampling unit 53 includes a sampling resistor Rs connected in series to the resonant loop, and the current sampling unit 53 samples the voltage across the sampling resistor Rs to obtain the first sampling signal V RS , which is equivalent to representing the loop current I Lr in the resonant loop. It can be understood that in other embodiments of the present application, other conventional current sampling structures can also be used to sample the loop current I Lr in the resonant loop, and the present application does not limit this.

[0095] The zero-crossing detection unit 52 is used to obtain the zero-crossing time of the excitation current (denoted as I Lm ) in the resonant loop.

[0096] Exemplarily, in a possible embodiment of the present application, the zero-crossing detection unit 52 is provided with an integration circuit and a comparison circuit. In each switching cycle, the integration circuit is configured to time-integrate a predetermined parameter of the flyback converter 5 during the conduction period of the first switch Q1 and the conduction period of the second switch Q2 respectively, to obtain a corresponding first integration result and a second integration result. At this time, the first integration result and the second integration result can correspond to the voltage signal output by the integration circuit. The comparison circuit is configured to compare the first integration result and the second integration result, and obtain the zero-crossing time ZCD of the excitation current I Lm at the moment when the second integration result is the same as the first integration result.

[0097] Based on the working principle of the transformer 51, it can be known that the voltage across the primary winding Np and the voltage across the secondary winding Ns in the transformer 51 have a certain proportional relationship with each other. Even when an auxiliary winding Na is provided in the transformer 51, the voltage across the primary winding Np, the voltage across the secondary winding Ns and the voltage Vaux across the auxiliary winding Na in the transformer 51 also have a certain proportional relationship with each other. Further, in some embodiments, the predetermined parameter for time integration by the zero-crossing detection unit 52 is the voltage across any winding in the transformer 51, as shown in FIG. 5. Exemplarily, taking the voltage across the auxiliary winding Na as the predetermined parameter, the predetermined parameter can be directly or indirectly sampled by the zero-crossing detection unit 52. Alternatively, the voltage across the auxiliary winding Na can be sampled by the zero-crossing detection unit 52 after being divided by a resistor, or the voltage across the auxiliary winding Na can be directly sampled by the zero-crossing detection unit 52. Figure 2

[0098] In another possible embodiment of the present application, the zero-crossing detection unit 52 is provided with a preset capacitor. In each switching cycle, the zero-crossing detection unit 52 charges the preset capacitor based on the current on the auxiliary winding Na in the transformer 51 during the conduction period of the first switch Q1, and discharges the preset capacitor based on the current on the auxiliary winding Na in the transformer 51 after the conduction of the second switch Q2. In addition, the zero-crossing detection unit 52 is also provided with a corresponding voltage detection unit to detect the voltage across the preset capacitor in real time after the conduction of the second switch Q2, and obtain the zero-crossing time ZCD of the excitation current I Lm when the voltage across the preset capacitor is detected to be lower than a preset voltage value.

[0099] The control circuit 54 is connected to the control terminals of the first switch Q1 and the second switch Q2, to control the conduction state of the first switch Q1 and the second switch Q2.

[0100] ​In the present application, the control circuit 100 is also connected with the current sampling unit 53 and the zero-crossing detection unit 52, so as to turn off the first switch tube Q1 when the first sampling signal V RS reach the reference value, turn off the second switch tube Q2 after the zero-crossing moment ZCD of the magnetizing current I Lm in the resonant circuit (for example, start timing at the zero-crossing moment ZCD of the magnetizing current I Lm in the resonant circuit, and turn off the second switch tube Q2 when the timing value reaches a preset time parameter, or turn off the second switch tube Q2 when the current value of the magnetizing current I Lm reaches a preset negative value, so as to facilitate subsequent realization of zero-voltage turn-on of the first switch tube Q1), so as to control the average value of the magnetizing current I Lm constant in each switching cycle.

[0101] In a switching cycle, if the dead time between the turn-on of the first switch tube Q1 and the turn-on of the second switch tube Q2 is ignored, when the converter is in a steady state, the average voltage on the primary winding Np of the converter is 0, and thus:

[0102] VC=VS=D*Vin.............(1),

[0103] wherein VC is the voltage across the first capacitor Cr, VS is the voltage across the diode D11 in the secondary side of the converter, D is the duty cycle of the control signal of the first switch tube Q1, and Vin is the input voltage of the converter.

[0104] When the first switch tube Q1 is turned on and the second switch tube Q2 is turned off, the inductance value of the first inductor Lk is much smaller than the inductance value of the primary winding Np (denoted as Lm), so the leakage inductance of the primary winding Np is ignored, and thus:

[0105] VLM1=Vin-VC-D*Vin=(1-D)*Vin.............(2),

[0106] wherein VLM1 is the voltage value across the primary winding Np of the converter during the turn-on of the first switch tube Q1.

[0107] When the first switch tube Q1 is turned off and the second switch tube Q2 is turned on, the first inductor Lk and the first capacitor Cr will resonate, so the leakage inductance of the primary winding Np cannot be ignored, and thus:

[0108] VLM2=N*Vo.................(3),

[0109] wherein VLM1 is the voltage value across the primary winding Np of the converter during the turn-off of the first switch tube Q1. Vo is the output voltage of the converter.

[0110] From the volt-second balance, we have:

[0111] VLM1*Ton=VLM2*Toff (4),

[0112] where Ton is the on-time of the first switch Q1 in a switching period, and Toff is the off-time of the first switch Q1 in a switching period.

[0113] Further, by combining formula (2), formula (3) and formula (4), we have:

[0114] (1-D)*Vin*Ton=N*Vo*Toff (5),

[0115] Based on the size relationship between Ton and Toff, formula (5) can be transformed to:

[0116] (1-D)*D*Vin=N*Vo*(1-D) (6),

[0117] That is,

[0118] N*Vo=D*Vin (7).

[0119] Based on formula (7), it can be obtained that the input-output relationship of the above asymmetric half-bridge flyback converter is similar to that of the Buck topology. Further, in the present application, the average value of the magnetizing current I Lm may be used to equivalently represent the output current Io of the converter, that is, by controlling the average value of the magnetizing current I Lm in each switching period to be constant, the output current Io of the converter can be equivalently realized to be constant, and constant current control can be realized.

[0120] In the present application, on the one hand, by turning off the first switch Q1 when the first sampling signal V RS reaches the reference value, the maximum value (i.e. the peak value) of the loop current I Lr of the converter in each switching period can be controlled to be constant. On the other hand, since in each switching period, the current waveform and the current size of the loop current I Lr in the resonant loop and the magnetizing current I Lm during the on-period of the first switch Q1 are the same, by turning off the first switch Q1 when the first sampling signal V RS reaches the reference value, the maximum value (i.e. the peak value) of the magnetizing current I Lm of the converter in each switching period can also be controlled to be constant. Further, by controlling the average value of the magnetizing current I Lmthe zero-crossing time ZCD of the zero-crossing signal ZCD, the second switch Q2 is turned off, so that the excitation current I of the converter can be controlled in each switching cycle Lm is constant. Thus, the average value of the excitation current I Lm can be controlled in each switching cycle, i.e. the constant current control of the converter output current is achieved.

[0121] Further, the control circuit 54 is also used to turn on the second switch Q2 after a first dead time when the first switch Q1 is turned off, and turn on the first switch Q1 after a second dead time when the second switch Q2 is turned off or the drain-source voltage across the first switch Q1 is 0V.

[0122] With reference to Figure 3 , in the embodiment, the control circuit 54 comprises a first turn-off trigger signal generating module 541, a first turn-on trigger signal generating module 542, a second turn-off trigger signal generating module 543 and a second turn-on trigger signal generating module 544.

[0123] The first turn-on trigger signal generating module 542 is used to generate the turn-on trigger signal of the first switch Q1 based on the drain-source voltage across the first switch Q1, or generate the turn-on trigger signal of the first switch Q1 based on the turn-off trigger signal of the second switch Q2 and the second dead time. Optionally, a voltage sampling unit can be arranged in the first turn-on trigger signal generating module 542, which is used to sample and detect the drain-source voltage across the first switch Q1, and generate the trigger signal for turning on the first switch Q1 when the zero voltage time of the drain-source voltage across the first switch Q1 from high level to low level is detected. In the embodiment, when the drain-source voltage across the first switch Q1 is detected to decrease from high level to a preset voltage value, it is determined that the zero voltage time of the drain-source voltage across the first switch Q1 from high level to low level is detected. Optionally, a first delay unit or a first timing unit can also be arranged in the first turn-on trigger signal generating module 542, and the first turn-on trigger signal generating module 542 is arranged to generate the trigger signal for turning on the first switch Q1 after receiving the turn-off trigger signal of the second switch Q2 and delaying for the second dead time.

[0124] It should be noted that the sampling position of the voltage sampling unit in the first turn-on trigger signal generating module 542 is not limited in the present application, as long as the drain-source voltage across the first switch Q1 can be finally sampled.

[0125] The second turn-on trigger signal generation module 544 is used to generate a turn-on trigger signal for the second switch Q2 based on the turn-off trigger signal of the first switch Q1 and a first dead time. In this embodiment, the second turn-on trigger signal generation module 544 is provided with a second delay unit or a second timing unit, and the second turn-on trigger signal generation module 544 is connected to the first turn-off trigger signal generation module 541. After receiving the turn-off trigger signal of the first switch Q1 and delaying for the first dead time, the second turn-on trigger signal generation module 544 generates a trigger signal to control the second switch Q2 to turn on.

[0126] The second shutdown trigger signal generation module 543 is used based on the excitation current I. Lm At the zero-crossing moment, ZCD generates a turn-off trigger signal for the second switch Q2. In this embodiment, the second turn-off trigger signal generation module 543 is connected to the zero-crossing detection unit 52. Optionally, when the zero-crossing detection unit 52 detects the excitation current I... Lm After the zero-crossing time ZCD, the second turn-off trigger signal generation module 543 can be directly triggered to generate a trigger signal to control the turn-off of the second switch Q2. Alternatively, a third delay unit or a third timing unit can be set in the second turn-off trigger signal generation module 543. When the zero-crossing detection unit 52 detects the excitation current I... Lm After the zero-crossing time ZCD, the third delay unit or the third timing unit is triggered to start timing. The second turn-off trigger signal generation module 543 can generate a trigger signal to control the turn-off of the second switch Q2 when the timing value reaches a preset time value, or when the excitation current I... Lm When the current value reaches a preset negative value, a trigger signal is generated to control the second switch Q2 to turn off, which facilitates the subsequent zero-voltage turn-on of the first switch Q1.

[0127] The first shutdown trigger signal generation module 541 is used to generate a signal based on the first sampled signal V. RS The reference value is used to generate the turn-off trigger signal for the first switch Q1.

[0128] In the first embodiment of the present invention, the above-mentioned reference value is a preset fixed value. It is understood that this embodiment is only used to achieve coarse constant current control. (Reference) Figure 5 During each switching cycle, it controls the excitation current I Lm The peak value IPK is fixed to achieve the excitation current I during each switching period. Lm The average value is constant, thus effectively achieving constant current control of the output current Io of the flyback converter 5. In this embodiment, since the excitation current I is not considered... Lm The negative part of the excitation current I is only within one switching cycle. Lm The signal is processed into a triangular wave that is approximately non-negative, and the excitation current I is then...Lm The average value is 0.5*IPK, meaning the output current Io of the flyback converter 5 is approximately 0.5*IPK*Nps. Here, IPK is a preset, fixed reference value, and Nps is the turns ratio of the primary winding Np and secondary winding Ns of the transformer 51. Thus, the constant current scheme shown in this embodiment has a simple control method and requires fewer signals to be sampled during the process of achieving constant output current (only the loop current Io in the resonant circuit). Lr Sampling is easier to implement and less expensive.

[0129] Understandable Figure 5 In the middle, it represents the excitation current I. Lm The solid line of the waveform represents the loop current I. Lr The dashed lines on the waveforms overlap, such as in the waveform portion within time period t1. This overlap indicates that the signal waveforms, i.e., the current magnitudes, are the same for both waveforms within this time period; that is, the excitation current I... Lm With loop current I Lr The maximum value or peak value is the same. And Figure 5 middle, I D11 This represents the current waveform across diode D11 in the secondary side of the converter, which is the magnetizing current I. Lm With loop current I Lr The difference.

[0130] In a second embodiment of the present invention, the above reference value is based on the loop current I of the resonant circuit. Lr The error signal generated after feedback. It should be noted that the reference... Figure 6 This embodiment can achieve precise constant current control within the first time period T1 of each switching cycle (including the entire on-time of the first switching transistor Q1, for example, from the excitation current I). Lm The zero-crossing moment ZCD to the moment when the first switch Q1 is turned off is used to characterize the loop current I in the resonant circuit by sampling through the current sampling unit 53. Lr The first sampled signal V RS The first feedback signal VFB1 is fed back in real time to the closed-loop current sampling circuit in the first turn-off trigger signal generation module 541, thereby realizing the monitoring of the excitation current I. Lm Real-time feedback. Furthermore, within the second time period T2 of each switching cycle (including the entire on-time of the second switch Q2, for example, from the moment the first switch Q1 is turned off to the excitation current I...), Lm The zero-crossing time ZCD), by sampling the first sampled signal V before the first switch Q1 is turned off. RSthe peak value of the excitation current I is sampled and held, and after a certain signal processing, a second feedback signal VFB2 is obtained and fed back to the closed-loop current sampling circuit in the first off trigger signal generating module 541, realizing the peak value feedback of the excitation current I Lm . In this way, the closed-loop current sampling circuit can output a corresponding error signal as the basis for controlling the off of the first switch tube Q1 according to the real-time feedback result and the peak value feedback result of the excitation current I Lm . For example, when the first sampling signal V RS reaches the error signal value, a trigger signal for controlling the off of the first switch tube Q1 is generated. The embodiment scheme can realize full current feedback of the excitation current I Lm at different time periods (including the negative part of the excitation current I Lm ) in one switch tube period, has higher accuracy, and finally has better constant current effect.

[0131] Specifically, referring to Figure 4 , in the embodiment, the first off trigger signal generating module 541 includes an error signal generating unit 5411 and a comparison unit 5412.

[0132] The error signal generating unit 5411 is used to generate an error signal according to the first feedback signal VFB1 and the second feedback signal VFB2. The non-inverting input terminal of the comparison unit 5412 receives the error signal, the inverting input terminal of the comparison unit 5412 receives the first sampling signal V RS , and the output terminal of the comparison unit 5412 outputs the off trigger signal Vgs_Q1_off of the first switch tube Q1.

[0133] In the embodiment, the error signal generating unit 5411 further includes a first current source Iref, a second capacitor COMP, a second current source I1, a first switch K1, a third current source I2, a second switch K2, a selection unit 5414, and a feedback signal generating unit 5413.

[0134] Among them, the first current source Iref and the second capacitor COMP are connected between the power supply end and the ground end, and the first current source Iref is used to inject current to the middle node A of the first current source Iref and the second capacitor COMP. The second current source I1 and the first switch K1 are connected between the middle node A of the first current source Iref and the second capacitor COMP and the ground end, which constitutes a first current path to the middle node A, and the second current source I1 extracts current from the middle node A when the first switch K1 is closed. The third current source I2 and the second switch K2 are connected between the middle node A of the first current source Iref and the second capacitor COMP and the ground end, which constitutes a second current path to the middle node A, and the third current source I2 extracts current from the middle node A when the second switch K2 is closed.

[0135] The feedback signal generation unit 5413 is configured to generate a first feedback signal VFB1 according to the first sampling signal V RS The first feedback signal VFB1 is the first sampling signal V RS obtained by the current sampling unit 53 in real time, and the second feedback signal VFB2 is obtained by the feedback signal generation unit 5413. RS The feedback signal generation unit 5413 further comprises a sample-and-hold unit and a voltage gain unit. The sample-and-hold unit is configured to sample and hold the peak value of the first sampling signal V Lm before the first switch Q1 is turned off to obtain a second sampling signal, and the voltage gain unit is configured to perform gain processing on the second sampling signal according to a gain coefficient to obtain the second feedback signal VFB2.

[0136] In this embodiment, the gain coefficient of the voltage gain unit is 0.5.

[0137] The selection unit 5414 is configured to generate first and second selection signals that are opposite to each other according to the control signal Vgs_Q1 of the first switch Q1 and the zero-crossing time ZCD of the excitation current I Lm . When the first selection signal is valid (e.g., high), the first switch K1 is closed, and when the second selection signal is valid (e.g., high), the second switch K2 is closed, so that the error signal generation unit 5411 extracts current from the intermediate node A based on the first current path in the first time period T1 and extracts current from the intermediate node A based on the second current path in the second time period T2.

[0138] Optionally, in an embodiment of the present application, the selection unit 5414 comprises an RS flip-flop 54141 and an RS flip-flop 54142, wherein the reset end of the RS flip-flop 54141 receives the control signal Vgs_Q1, the set end of the RS flip-flop 54141 receives the zero-crossing time ZCD of the excitation current I Lm , the first output end of the RS flip-flop 54141 outputs the first selection signal, the reset end of the RS flip-flop 54142 receives the zero-crossing time ZCD of the excitation current I Lm , the set end of the RS flip-flop 54142 receives the control signal Vgs_Q1, and the first output end of the RS flip-flop 54142 outputs the second selection signal. Alternatively, in another embodiment of the present application, only one RS flip-flop can be provided in the selection unit 5414, such as the RS flip-flop 54141, and the first selection signal is outputted by the first output end of the RS flip-flop 54141, and the second selection signal is outputted by the second output end of the RS flip-flop 54141.

[0139] In this embodiment, the first current source Iref is a constant current source for providing a fixed reference current, the second current source I1 is a voltage-controlled current source controlled by the first feedback signal VFB1, and the third current source I2 is a voltage-controlled current source controlled by the second feedback signal VFB2. That is, the first feedback signal VFB1 can control the size of the extraction capability of the second current source I1 when extracting current from the intermediate node A, thereby realizing real-time feedback of the excitation current I Lm ; the second feedback signal VFB2 can control the size of the extraction capability of the third current source I2 when extracting current from the intermediate node A, thereby realizing peak feedback of the excitation current I Lm . Further, the error signal generation unit 5411 generates an error signal across the second capacitor COMP by charging the second capacitor COMP with the first current source Iref, the second current source I1, and the third current source I2.

[0140] In this embodiment, the capacitance value of the second capacitor COMP can serve as a reference for controlling the turn-off of the first switch tube Q1, which determines the turn-off time of the first switch tube Q1. Therefore, by changing the capacitance value of the second capacitor COMP, the turn-off time of the first switch tube Q1 can be adjusted. In actual applications, the capacitance value of the second capacitor COMP should be reasonably set according to specific requirements.

[0141] Further, the control circuit 54 further includes a first RS flip-flop 545, a first driver 546, a second RS flip-flop 547, and a second driver 548. The reset end of the first RS flip-flop 545 is connected to the output end of the first turn-off trigger signal generation module 541, the set end of the first RS flip-flop 545 is connected to the output end of the first turn-on trigger signal generation module 542, and the output end of the first RS flip-flop 545 is connected to the control end of the first switch tube Q1 via the first driver 546 to provide the first control signal Vgs_Q1 to the first switch tube Q1.

[0142] The reset end of the second RS flip-flop 547 is connected to the output end of the second turn-off trigger signal generation module 543, the set end of the second RS flip-flop 547 is connected to the output end of the second turn-on trigger signal generation module 544, and the output end of the second RS flip-flop 547 is connected to the control end of the second switch tube Q2 via the second driver 548 to provide the second control signal Vgs_Q2 to the second switch tube Q2.

[0143] Further, the present application also discloses a constant current control method of a flyback converter, which can be applied to the flyback converter shown in Figures 2 to 6 .

[0144] As shown in Figure 7 , in each switching period, the constant current control method includes the following steps:

[0145] In step S1, a loop current in the resonant loop is sampled to obtain a first sampling signal.

[0146] In the embodiment, step S1 can be understood by referring to the foregoing description of the current sampling unit 53, which will not be repeated here.

[0147] In step S2, a zero-crossing time of the excitation current in the resonant loop is obtained.

[0148] In a possible embodiment of the present application, the method for determining the zero-crossing time of the excitation current comprises: integrating a predetermined parameter of the flyback converter with respect to time during the conduction of the first switch tube to obtain a first integral result; integrating the predetermined parameter of the flyback converter with respect to time after the conduction of the second switch tube to obtain a second integral result, and obtaining the zero-crossing time of the excitation current when the second integral result reaches the first integral result, wherein the transformer comprises a primary winding, a secondary winding and an auxiliary winding, and the predetermined parameter is a voltage across any winding of the transformer. In another possible embodiment of the present application, the method for determining the zero-crossing time of the excitation current comprises: charging a preset capacitor based on a current on the auxiliary winding of the transformer during the conduction of the first switch tube; discharging the preset capacitor based on the current on the auxiliary winding of the transformer after the conduction of the second switch tube, and obtaining the zero-crossing time of the excitation current when a voltage across the preset capacitor is lower than a preset voltage value. Step S2 can be understood by referring to the foregoing description of the zero-crossing detection unit 52, which will not be repeated here.

[0149] In step S3, the first switch tube is turned off when the first sampling signal reaches a reference value, and the second switch tube is turned off after the zero-crossing time of the excitation current in the resonant loop, so as to control the average value of the excitation current to be constant.

[0150] In the present application, optionally, turning off the second switch tube after the zero-crossing time of the excitation current in the resonant loop comprises: turning off the second switch tube after a preset time from the zero-crossing time of the excitation current; or turning off the second switch tube when the excitation current reaches a preset negative value.

[0151] In the first embodiment of the present application, the reference value is a preset fixed value. It can be understood that the embodiment scheme is only used to realize rough constant current control.

[0152] In the second embodiment of the present application, the reference value is a value obtained according to the loop current I Lr the error signal generated after feedback. It should be noted that the embodiment scheme can realize accurate constant current control.

[0153] In the precise constant current control scheme, the turning off the first switch tube when the first sampling signal reaches the reference value includes: setting the first current source, the second current source and the third current source to charge the second capacitor, so as to obtain the error signal between the two ends of the second capacitor; comparing the first sampling signal and the error signal, and turning off the first switch tube when the first sampling signal reaches the error signal. Wherein, the first current source injects current into the second capacitor, the second current source and the third current source extract current from the second capacitor at different times, and the second current source is controlled by the first feedback signal, and the third current source is controlled by the second feedback signal.

[0154] Further, the first feedback signal is the first sampling signal. The method for obtaining the second feedback signal includes: sampling and holding the peak value of the first sampling signal before the first switch tube is turned off to obtain the second sampling signal; and setting a gain coefficient to perform gain processing on the second sampling signal to obtain the second feedback signal. In this way, full current feedback of the excitation current I Lm can be realized. In this embodiment, the gain coefficient is 0.5. Step S3 can be understood with reference to the foregoing description of the first turn-on trigger signal generation module 541 and the second turn-on trigger signal generation module 543, which will not be repeated here.

[0155] Further, the constant current control method of the application further includes: turning on the second switch tube after the first dead time of the first switch tube being turned off; and turning on the first switch tube after the second dead time of the second switch tube being turned off or when the drain-source voltage between the two ends of the first switch tube is lower than the preset voltage value. The specific description can be understood with reference to the foregoing description of the first turn-on trigger signal generation module 542 and the second turn-on trigger signal generation module 544, which will not be repeated here.

[0156] In summary, in each switching period of the converter, the first switch tube is turned off when the loop current in the resonant loop of the primary side reaches the reference value, and the second switch tube is turned off at the zero-crossing moment of the excitation current, so that the maximum and minimum values of the excitation current are constant when the excitation current changes, and the average value of the excitation current in each switching tube period is constant, i.e. the constant current control of the output current is equivalent. The control method is simple, and the constant current effect is better. At the same time, when the voltage conversion is realized, no additional resonant inductor and devices such as optocoupler and isolation operational amplifier are needed, which can reduce the cost and the size of the circuit.

[0157] Finally, it should be noted that: obviously, the above embodiments are only examples for clearly illustrating the application, and are not limitations on the embodiments. Based on the above description, other different forms of changes or variations can be made by those skilled in the art. Here, it is not necessary and impossible to enumerate all the embodiments. The obvious changes or variations derived therefrom are still within the protection scope of the application.

Claims

1. A constant current control method of a flyback converter, the flyback converter comprising a first switch and a second switch constituting a half bridge, a first inductor, a first capacitor and a primary winding in a transformer constituting a resonance circuit with the second switch, wherein, The constant current control method comprises the following steps in each switching cycle: sampling a loop current in a resonant loop to obtain a first sampling signal; obtaining a zero-crossing moment of an excitation current in the resonant loop; turning off the first switch tube when the first sampling signal reaches a reference value, and turning off the second switch tube after the zero-crossing moment of the excitation current to control the average value of the excitation current to be constant.

2. The constant current control method according to claim 1, wherein The reference value is a preset fixed value.

3. The constant current control method according to claim 1, wherein The reference value is an error signal generated according to the loop current feedback.

4. The constant current control method according to claim 3, wherein The step of turning off the first switch tube when the first sampling signal reaches the reference value comprises the following steps: charging a second capacitor by using a first current source, a second current source and a third current source to obtain the error signal across the second capacitor; comparing the first sampling signal and the error signal, and turning off the first switch tube when the first sampling signal reaches the error signal. The first current source injects current into the second capacitor, the second current source and the third current source extract current from the second capacitor at different times, the second current source is controlled by a first feedback signal, and the third current source is controlled by a second feedback signal.

5. The constant current control method according to claim 4, wherein The first feedback signal is the first sampling signal. The second feedback signal is obtained by the following method: sampling and holding a peak value of the first sampling signal before the first switch tube is turned off to obtain a second sampling signal; performing gain processing on the second sampling signal by using a gain coefficient to obtain the second feedback signal. The method for determining the zero-crossing moment of the excitation current comprises the following steps:

6. The constant current control method of claim 1, wherein, integrating a predetermined parameter of the flyback converter with respect to time during the conduction of the first switch tube to obtain a first integral result; integrating the predetermined parameter of the flyback converter with respect to time after the conduction of the second switch tube to obtain a second integral result, and obtaining the zero-crossing moment of the excitation current when the second integral result reaches the first integral result. The transformer comprises a primary winding, a secondary winding and an auxiliary winding, and the predetermined parameter is a voltage across any winding of the transformer. The method for determining the zero-crossing moment of the excitation current comprises the following steps:

7. The constant current control method of claim 1, wherein, charging a preset capacitor based on a current on the auxiliary winding of the transformer during the conduction of the first switch tube; discharging the preset capacitor based on the current on the auxiliary winding of the transformer after the conduction of the second switch tube, and obtaining the zero-crossing moment of the excitation current when a voltage across the preset capacitor is lower than a preset voltage value. The step of turning off the second switch tube after the zero-crossing moment of the excitation current in the resonant loop comprises the following steps:

8. The constant current control method according to any one of claims 6 and 7, wherein, turning off the second switch tube after a preset time from the zero-crossing moment of the excitation current; or turning off the second switch tube when the excitation current reaches a preset negative value. The constant current control method further comprises the following steps:

9. The constant current control method of claim 1, wherein, turning on the second switch tube after a first dead time when the first switch tube is turned off; turning on the first switch tube after a second dead time when the second switch tube is turned off or when a drain-source voltage across the first switch tube is lower than a preset voltage value. The constant current control method further comprises the following steps:

10. A flyback converter, wherein, a transformer comprising a primary winding and a secondary winding; ​ a first switch and a second switch connected in series between an input terminal and a reference ground; a first inductor and a first capacitor connected with the primary winding and the second switch to form a resonant circuit; a current sampling unit connected with the resonant circuit to sample a loop current in the resonant circuit to obtain a first sampling signal; a zero-crossing detection unit to obtain a zero-crossing time of the excitation current in the resonant circuit; a control circuit connected with control terminals of the first switch and the second switch to control conduction states of the first switch and the second switch, wherein the control circuit is connected with the current sampling unit and the zero-crossing detection unit to turn off the first switch when the first sampling signal reaches a reference value and to turn off the second switch after the zero-crossing time of the excitation current to control an average value of the excitation current to be constant.

11. The flyback converter of claim 10, wherein, the control circuit is further configured to turn on the second switch after a first dead time when the first switch is turned off; and the control circuit is further configured to turn on the first switch after a second dead time when the second switch is turned off or when a drain-source voltage across the first switch is lower than a preset voltage value. the control circuit comprises:

12. The flyback converter of claim 11, wherein, a first conduction trigger signal generation module configured to generate a conduction trigger signal of the first switch based on the drain-source voltage across the first switch or to generate the conduction trigger signal of the first switch based on a turn-off trigger signal of the second switch and the second dead time; a first turn-off trigger signal generation module configured to generate a turn-off trigger signal of the first switch based on the first sampling signal and the reference value; a second conduction trigger signal generation module configured to generate a conduction trigger signal of the second switch based on the turn-off trigger signal of the first switch and the first dead time; a second turn-off trigger signal generation module configured to generate a turn-off trigger signal of the second switch based on the zero-crossing time of the excitation current. the reference value is a preset fixed value.

13. The flyback converter of claim 10, wherein, the reference value is an error signal generated according to a feedback of the loop current.

14. The flyback converter of claim 10, wherein, the first turn-off trigger signal generation module comprises:

15. The flyback converter of claim 12, wherein, an error signal generation unit configured to generate an error signal according to a first feedback signal and a second feedback signal; and a comparison unit having a positive input terminal receiving the error signal, a negative input terminal receiving the first sampling signal, and an output terminal outputting the turn-off trigger signal of the first switch. the error signal generation unit comprises:

16. The flyback converter of claim 15, wherein, a first current source and a second capacitor connected between a power supply terminal and a ground terminal, the first current source being configured to inject a current to a middle node of the first current source and the second capacitor; a second current source and a first switch connected between the middle node and the ground terminal, the second current source being configured to extract a current from the middle node when the first switch is turned on; a third current source and a second switch connected between the middle node and the ground terminal, the third current source being configured to extract a current from the middle node when the second switch is turned on; and a fourth current source and a third switch connected between the middle node and the ground terminal, the fourth current source being configured to extract a current from the middle node when the first switch is turned off. The selection unit is configured to generate first and second selection signals that are mutually opposite according to the control signal of the first switch tube and the zero-crossing moment of the excitation current, the first selection signal being effective to control the first switch to be closed, and the second selection signal being effective to control the second switch to be closed. The feedback signal generation unit is configured to generate the first and second feedback signals according to the first sampling signal. The second current source is a voltage-controlled current source controlled by the first feedback signal, and the third current source is a voltage-controlled current source controlled by the second feedback signal, and the first, second and third current sources jointly charge the second capacitor to generate the error signal across the second capacitor.

17. The flyback converter of claim 16, wherein, The feedback signal generation unit comprises: The sample-and-hold unit is configured to sample and hold the peak value of the first sampling signal before the first switch tube is turned off to obtain a second sampling signal; The voltage gain unit is configured to perform gain processing on the second sampling signal according to a gain coefficient to obtain the second feedback signal, The first feedback signal is a first sampling signal obtained by real-time sampling of the current sampling unit.

18. A lighting system, wherein, The rectifier circuit is configured to rectify an input AC power supply to output a first voltage signal; The power factor correction circuit is connected with the rectifier circuit and configured to perform power factor correction on the first voltage signal and output a second voltage signal; and The flyback converter of any one of claims 10-17 is configured to implement constant current driving of an LED load. ​

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