Control Method of Power Conversion Device

Abstract

The present disclosure provides a control method for a power conversion device, the power conversion device including a first bridge arm, a second bridge arm, and two transformers. The first bridge arm includes first to third switching tubes connected in series, and the second bridge arm includes fourth to sixth switching tubes connected in series. The windings of the two transformers are connected in series between the nodes of the two bridge arms. The control method includes: controlling the first and fourth switching tubes to operate with a duty cycle and the switching timings are out of phase with each other by 180 degrees; controlling the control signals of the third and fourth switching tubes to be complementary, and the control signals of the sixth and first switching tubes to be complementary; and when the duty cycle is less than or equal to 0.5, controlling the second and fourth switching tubes to switch synchronously, and the fifth and first switching tubes to switch synchronously, and when the duty cycle is greater than 0.5, controlling the second and sixth switching tubes to switch synchronously, and the fifth and third switching tubes to switch synchronously.

Description

Technical Field

[0001] The present disclosure relates to a control method, particularly to a control method for a power conversion device. Background Art

[0002] With the rapid development of the Internet and artificial intelligence, the demand for power supplies with high efficiency and high power density is increasing. To meet the increasing required power, the existing approach is to increase the bus voltage from 12V to 54V to reduce bus losses and costs, and then add a bus converter to step down the bus voltage to 12V for power supply.

[0003] In order to obtain a higher system conversion efficiency, for voltage conversion applications with an input of 40V - 60V and an output of 12V, the following two schemes are mostly adopted for bus converters in the prior art. Scheme one selects two magnetically integrated BUCK step-down converters with a 180-degree phase shift to adjust the output voltage by regulating the duty cycle. The circuit implementation of scheme one is simple, but the duty cycle is too small, the voltage stress of the switching tube is high, and the volume of the magnetic component is large, resulting in poor performance of the converter. Scheme two selects a magnetically integrated hard-switching full-bridge converter to adjust the output voltage by regulating the turns ratio of the primary and secondary sides of the transformer and the duty cycle. The circuit implementation of scheme two is simple and has high reliability, but its duty cycle is limited within 0.5, and the volume of the magnetic component and the voltage stress of the secondary switching tube cannot be further reduced, restricting the performance of the converter. Moreover, in scheme two, due to the large number of turns of the transformer, the number of layers of the required PCB is large, resulting in a significant increase in the cost of the converter.

[0004] Therefore, how to develop a control method for a power conversion device that can improve the above-mentioned prior art is an urgent need at present. Summary of the Invention

[0005] The purpose of the present disclosure is to provide a control method for a power conversion device, which can make the transformer inductance volt-second in the power conversion device lower, so the voltage stress of the switching tube is low, and the volume of the magnetic component is small. Therefore, magnetic components with a smaller volume and switching tubes with low withstand voltage can be correspondingly adopted, thereby reducing costs and increasing power density.

[0006] To achieve the above object, the present disclosure provides a control method, which is configured on a power conversion device. The power conversion device includes a first arm, a second arm and two transformers. The first arm includes a first switch, a second switch and a third switch connected in series. There are a first primary side node and a first secondary side node between the first switch, the second switch and the third switch respectively. The second arm includes a fourth switch, a fifth switch and a sixth switch connected in series. There are a second primary side node and a second secondary side node between the fourth switch, the fifth switch and the sixth switch respectively. Each transformer includes a primary side winding and a secondary side winding. The two primary side windings of the two transformers are connected in series between the first primary side node and the second primary side node, and the two secondary side windings of the two transformers are connected in series between the first secondary side node and the second secondary side node. The control method includes: controlling the first and fourth switches to operate at a duty cycle, and controlling the switching timings of the first and fourth switches to be out of phase by 180 degrees; controlling the control signals of the third switch and the fourth switch to be complementary, and controlling the control signals of the sixth switch and the first switch to be complementary; and when the duty cycle is less than or equal to 0.5, controlling the second switch and the fourth switch to conduct and turn off synchronously, and controlling the fifth switch and the first switch to conduct and turn off synchronously, and when the duty cycle is greater than 0.5, controlling the second switch and the sixth switch to conduct and turn off synchronously, and controlling the fifth switch and the third switch to conduct and turn off synchronously. BRIEF DESCRIPTION OF THE DRAWINGS

[0007] Figure 1 is a schematic circuit structure diagram of a power conversion device according to a preferred embodiment of the present disclosure.

[0008] Figure 2 illustrates Figure 1 the exciting inductance of the transformer of the power conversion device in

[0009] Figure 3 is Figure 1 a schematic diagram of the main waveforms of the power conversion device in when the duty cycle is less than or equal to 0.5.

[0010] Figure 4A , Figure 4B , Figure 4C and Figure 4D are Figure 1 schematic diagrams of the operating modes of the power conversion device in when the duty cycle is less than or equal to 0.5.

[0011] Figure 5 is Figure 1 a schematic diagram of the main waveforms of the power conversion device in when the duty cycle is greater than 0.5.

[0012] Figure 6A , Figure 6B , Figure 6C andFigure 6D Schematic diagram of the operating mode of the power conversion device Figure 1 when the duty cycle is greater than 0.5.

[0013] Figure 7A 、 Figure 7B 、 Figure 7C 、 Figure 8 、 Figure 9 and Figure 10A Schematic diagrams of circuit structures of various variations of the power conversion device Figure 1 of

[0014] Figure 10B and Figure 10C show different implementation modes of winding the windings of the transformer Figure 10A of

[0015] Figure 11 Schematic diagram of the circuit structure of a variation of the power conversion device Figure 1 when the duty cycle is less than or equal to 0.5.

[0016] Figure 12 Schematic diagram of the circuit structure of the power conversion device according to another preferred embodiment of the present disclosure.

[0017] Figure 13 and Figure 14 Schematic diagrams of circuit structures of different variations of the power conversion device Figure 12 of

[0018] Figure 15 Schematic diagram of the flow of the control method of the power conversion device according to a preferred embodiment of the present disclosure.

[0019] Among them, the reference numerals are explained as follows:

[0020] 1: Power conversion device

[0021] 11: DC voltage source

[0022] Vin: Input voltage

[0023] Vin+: Positive input terminal

[0024] Vin-: Negative input terminal

[0025] 12: First bridge arm

[0026] Q1: First switching transistor

[0027] Q2: Second switching transistor

[0028] Q3: Third switching transistor

[0029] A1: First primary side node

[0030] B1: First secondary side node

[0031] 13: Second bridge arm

[0032] Q4: Fourth switching transistor

[0033] Q5: Fifth switching transistor

[0034] Q6: Sixth switching transistor

[0035] A2: Second primary side node

[0036] B2: Second secondary side node

[0037] T1, T2: Transformer

[0038] T1A, T2A: Primary side winding

[0039] A3: Third primary side node

[0040] T1B, T2B: Secondary side winding

[0041] B3: Third secondary side node

[0042] Co: Output capacitor

[0043] Vo: Output voltage

[0044] Vo+: Positive output terminal

[0045] Vo-: Negative output terminal

[0046] D: Duty cycle

[0047] C1: First capacitor

[0048] C2: Second capacitor

[0049] L1, L2: Magnetizing inductance

[0050] t0, t1, t2, t3, t4, t5, t6, t7, t8: Moments

[0051] PWM1: First control signal

[0052] PWM2: Second control signal

[0053] PWM3: Third control signal

[0054] PWM4: Fourth control signal

[0055] iT1B, iT2B: Current

[0056] Ts: Switching period

[0057] D1: First diode

[0058] D2: The second diode

[0059] D3: The third diode

[0060] D4: The fourth diode

[0061] P: The clamping node

[0062] C3: The third capacitor

[0063] C4: The fourth capacitor

[0064] C5: The fifth capacitor

[0065] C6: The sixth capacitor

[0066] C7: The seventh capacitor

[0067] A4, A5: Nodes

[0068] 2, 3: The magnetic core components

[0069] 21, 22, 31, 32: The winding posts

[0070] 23: The middle post

[0071] 33: The side post

[0072] d1, d2: The air gaps

[0073] d3, d4: The distances

[0074] S1, S2, S3, S4: Steps Detailed implementation manners

[0075] Some typical embodiments embodying the features and advantages of the present disclosure will be described in detail in the following description. It should be understood that the present disclosure can have various variations in different aspects, all of which do not depart from the scope of the present disclosure, and the descriptions and illustrations therein are for illustrative purposes in nature and not intended to limit the present disclosure.

[0076] Figure 1 It is a schematic circuit diagram of a power conversion device according to a preferred embodiment of the present disclosure. As Figure 1 shown, the power conversion device 1 includes an input end, a first bridge arm 12, a second bridge arm 13, two transformers T1, T2 and an output capacitor Co, wherein the input end is electrically connected to a DC voltage source 11 and has an input positive terminal Vin+ and an input negative terminal Vin-, the DC voltage source 11 provides an input voltage Vin, and the input voltage Vin is a DC voltage.

[0077] The first bridge arm 12 is connected in parallel to the input terminal and includes a first switch Q1, a second switch Q2, and a third switch Q3 that are serially coupled. The first switch Q1 and the third switch Q3 are electrically connected to the positive input terminal Vin+ and the negative input terminal Vin- respectively. There is a first primary side node A1 between the first switch Q1 and the second switch Q2, and a first secondary side node B1 between the second switch Q2 and the third switch Q3.

[0078] The second bridge arm 13 is connected in parallel to the input terminal and includes a fourth switch Q4, a fifth switch Q5, and a sixth switch Q6 that are serially coupled. The fourth switch Q4 and the sixth switch Q6 are electrically connected to the positive input terminal Vin+ and the negative input terminal Vin- respectively. There is a second primary side node A2 between the fourth switch Q4 and the fifth switch Q5, and a second secondary side node B2 between the fifth switch Q5 and the sixth switch Q6. The switches in the first bridge arm 12 and the second bridge arm 13 can be, for example but not limited to, MOSFETs, SiC switches, or GaN switches. In some embodiments, the third switch Q3 and the sixth switch Q6 can be diodes.

[0079] Each transformer (T1, T2) includes a primary side winding (T1A, T2A) and a secondary side winding (T1B, T2B), and the primary side windings (T1A, T2A) and the secondary side windings (T1B, T2B) of each transformer (T1, T2) are mutually coupled. The two primary side windings T1A and T2A of the two transformers T1 and T2 are serially coupled between the first primary side node A1 and the second primary side node A2, and the opposite ends of the two primary side windings T1A and T2A are electrically connected to form a third primary side node A3. The two secondary side windings T1B and T2B of the two transformers T1 and T2 are serially coupled between the first secondary side node B1 and the second secondary side node B2, and the opposite ends of the two secondary side windings T1B and T2B are electrically connected to form a third secondary side node B3. The turns ratio of the primary side windings T1A and T2A to the secondary side windings T1B and T2B is N:N:1:1, where N is a positive integer.

[0080] Both ends of the output capacitor Co are electrically connected to the positive output terminal Vo+ and the negative output terminal Vo- respectively, where the positive output terminal Vo+ is coupled to the third secondary side node B3, and the negative output terminal Vo- is coupled to the negative input terminal Vin-. The positive output terminal Vo+ and the negative output terminal Vo- form an output terminal, and the voltage between the positive output terminal Vo+ and the negative output terminal Vo- is the output voltage Vo.

[0081] In some embodiments, the power conversion device 1 further includes a controller (not shown), and the controller is configured to control the operation of the switches in the first bridge arm 12 and the second bridge arm 13.

[0082] In Figure 1In the illustrated embodiment, the power conversion device 1 further includes a first capacitor C1 and a second capacitor C2. The two ends of the first capacitor C1 are electrically connected to the input positive terminal Vin+ and the third primary side node A3 respectively, and the two ends of the second capacitor C2 are electrically connected to the third primary side node A3 and the third secondary side node B3 respectively. In addition, in this embodiment, the two transformers are the first transformer T1 and the second transformer T2 respectively. The primary side winding T1A of the first transformer T1 is electrically connected between the first primary side node A1 and the third primary side node A3, and the secondary side winding T1B of the first transformer T1 is electrically connected between the second secondary side node B2 and the third secondary side node B3. The primary side winding T2A of the second transformer T2 is electrically connected between the second primary side node A2 and the third primary side node A3, and the secondary side winding T2B of the second transformer T2 is electrically connected between the first secondary side node B1 and the third secondary side node B3.

[0083] Figure 2 Illustrates Figure 1 The equivalent circuit structure schematic diagram of the power conversion device in, where Figure 2 The exciting inductance of the transformer is shown in. As Figure 2 Shown, the exciting inductance L1 of the first transformer T1 can be equivalently connected in parallel with the secondary side winding T1B of the first transformer T1, and the exciting inductance L2 of the second transformer T2 can be equivalently connected in parallel with the secondary side winding T2B of the second transformer T2. However, it is not limited thereto. In some other embodiments, the exciting inductance L1 of the first transformer T1 can also be equivalently connected in parallel with the primary side winding T1A of the first transformer T1, and the exciting inductance L2 of the second transformer T2 can also be equivalently connected in parallel with the primary side winding T2A of the second transformer T2.

[0084] The following will be described separately Figure 1 The working modes and waveforms of the power conversion device 1 when the duty cycle D is less than or equal to 0.5 and the duty cycle D is greater than 0.5.

[0085] Figure 3 Is Figure 1 The main waveform schematic diagram of the power conversion device when the duty cycle is less than or equal to 0.5, Figure 4A , Figure 4B , Figure 4C And Figure 4D Is Figure 1 The working mode schematic diagram of the power conversion device when the duty cycle is less than or equal to 0.5. As Figure 3As shown, the time period from t0 to t8 is a working cycle. The first switching transistor Q1 and the fourth switching transistor Q4 operate with a duty cycle D. The switching timings of the first switching transistor Q1 and the fourth switching transistor Q4 are phase-shifted by 180 degrees from each other. The control signals of the third switching transistor Q3 and the fourth switching transistor Q4 are complementary, and the control signals of the sixth switching transistor Q6 and the first switching transistor Q1 are complementary. The second switching transistor Q2 conducts and turns off synchronously with the fourth switching transistor Q4, and the fifth switching transistor Q5 conducts and turns off synchronously with the first switching transistor Q1. To achieve the foregoing switching operations, in some embodiments, a first control signal PWM1, a second control signal PWM2, a third control signal PWM3, and a fourth control signal PWM4 may be output by a controller to control the switching transistors. The first control signal PWM1 and the second control signal PWM2 have a duty cycle D and are phase-shifted by 180 degrees from each other. The third control signal PWM3 is complementary to the first control signal PWM1, and the fourth control signal PWM4 is complementary to the second control signal PWM2. The first switching transistor Q1 and the fifth switching transistor Q5 are controlled by the first control signal PWM1, the second switching transistor Q2 and the fourth switching transistor Q4 are controlled by the second control signal PWM2, the sixth switching transistor Q6 is controlled by the third control signal PWM3, and the third switching transistor Q3 is controlled by the fourth control signal PWM4.

[0086] Please refer to Figure 3 and Figure 4A , during the time period from t0 to t1, the first, third, and fifth switching transistors Q1, Q3, and Q5 are in the conducting state, and the second, fourth, and sixth switching transistors Q2, Q4, and Q6 are in the off state. The corresponding working mode is as shown in Figure 4A . The primary side windings T1A and T2A and the secondary side windings T1B and T2B are connected in series in sequence to form a series branch. At this time, the input voltage Vin is directly applied across both ends of the series branch. The input voltage Vin is superimposed between the input negative terminal Vin- and the first primary side node A1 through the conduction of the first switching transistor Q1 and the fifth switching transistor Q5. Since the third switching transistor Q3 is conducting, the voltage across both ends of the secondary side winding T2B of the second transformer T2 is clamped by the output voltage Vo. Also, because the primary side winding T2A of the second transformer T2 is coupled with the secondary side winding T2B and the turns ratio is N:1, the voltage across both ends of the primary side winding T2A of the second transformer T2 is clamped by N*Vo. In addition, because the primary side winding T1A of the first transformer T1 is coupled with the secondary side winding T1B and the turns ratio is N:1, the voltage drop across both ends of the secondary side winding T1B of the first transformer T1 is (Vin - 2*N*Vo) / 2, and the magnetizing current flowing through the first transformer T1 increases in the positive direction, that is, the current iT1B flowing through the secondary side winding T1B increases in the positive direction, where Figure 4AThe forward direction of the current iT1B is marked by an arrow line. Accordingly, the magnetizing inductor of the first transformer T1 stores energy. During the period from time t0 to t1, the volt-second of the first transformer T1 is (Vin - 2*N*Vo)*D*Ts, where Ts is a switching period (i.e., equal to the duration from time t0 to t8). Additionally, during the period from time t0 to t1, the magnetizing current flowing through the second transformer T2 decreases in the positive direction, that is, the current iT2B flowing through the secondary side winding T2B decreases in the positive direction, where Figure 4A The forward direction of the current iT2B is marked by an arrow line. Accordingly, the magnetizing inductor of the second transformer T2 releases energy to the output terminal. In summary, during the period from time t0 to t1, the power conversion device 1 receives power from the input terminal and stores energy in the first transformer T1 while outputting power to the output terminal.

[0087] Please refer to Figure 3 and Figure 4B , during the period from time t2 to t3, the third and sixth switching tubes Q3 and Q6 are in the on state, and the first, second, fourth, and fifth switching tubes Q1, Q2, Q4, and Q5 are in the off state. The corresponding operating mode is as shown in Figure 4B . The input terminal is disconnected, and the magnetizing inductors of the first transformer T1 and the second transformer T2 both decrease in the positive direction. During the period from time t2 to t3, the power conversion device 1 transfers the energy stored in the transformers to the output terminal.

[0088] Please refer to Figure 3 and Figure 4C , during the period from time t4 to t5, the second, fourth, and sixth switching tubes Q2, Q4, and Q6 are in the on state, and the first, third, and fifth switching tubes Q1, Q3, and Q5 are in the off state. The corresponding operating mode is as shown in Figure 4CAs shown. The primary side windings T2A and T1A and the secondary side windings T2B and T1B are connected in series in sequence to form a series branch. At this time, the input voltage Vin is directly applied across both ends of this series branch, and the input voltage Vin is superimposed between the input negative terminal Vin- and the second primary side node A2 through the conduction of the fourth switch Q4 and the sixth switch Q6. Since the sixth switch Q6 is conducting, the voltage across both ends of the secondary side winding T1B of the first transformer T1 is clamped by the output voltage Vo. Also, because the primary side winding T1A of the first transformer T1 is coupled with the secondary side winding T1B and the turns ratio is N:1, the voltage across both ends of the primary side winding T1A of the first transformer T1 is clamped by N*Vo. In addition, because the primary side winding T2A of the second transformer T2 is coupled with the secondary side winding T2B and the turns ratio is N:1, the voltage drop across both ends of the secondary side winding T2B of the second transformer T2 is (Vin - 2*N*Vo) / 2, and the magnetizing current flowing through the second transformer T2 increases positively. Accordingly, the magnetizing inductor of the second transformer T2 stores energy. During the period from time t4 to t5, the volt-second of the second transformer T2 is (Vin - 2*N*Vo)*D*Ts. Additionally, during the period from time t4 to t5, the magnetizing current flowing through the first transformer T1 decreases positively. Accordingly, the magnetizing inductor of the first transformer T1 releases energy to the output terminal. In summary, during the period from time t4 to t5, the power conversion device 1 receives power from the input terminal, and while outputting power to the output terminal, stores energy for the second transformer T2.

[0089] Please refer to Figure 3 and Figure 4D , during the period from time t6 to t7, the third and sixth switches Q3 and Q6 are in the conducting state, and the first, second, fourth, and fifth switches Q1, Q2, Q4, and Q5 are in the off state. The corresponding operating mode is as Figure 4D shown. Since Figure 4D the operating mode shown is the same as Figure 4B the operating mode shown, it will not be elaborated here.

[0090] During Figure 3 , the periods from time t1 to t2, from time t3 to t4, from time t5 to t6, and from time t7 to t8 are all dead times. During the dead times, commutation is carried out only through the parasitic diodes or parasitic capacitances of the switches.

[0091] When the duty cycle D is less than or equal to 0.5, within one switching period Ts, that is Figure 3During the period from time t0 to t8, the volt-seconds of the exciting current flowing through the first transformer T1 during the positive increasing stage and the positive decreasing stage are respectively (Vin - 2 * N * Vo) * 0.5 * D * Ts and Vo * (1 - D) * Ts. Through the turns ratio relationship of the transformer and the volt-second balance of the magnetic component, we can obtain: [Vin - (N + 1) * Vo] * D * Ts / (1 + N) = Vo * (1 - D) * Ts. After simplification, the expression of the output voltage Vo can be obtained: Vo = Vin * D / (1 + N).

[0092] Figure 5 For Figure 1 the power conversion device when the duty cycle is greater than 0.5, the schematic diagram of the main waveforms Figure 6A and Figure 6B and Figure 6C and Figure 6D For Figure 1 the power conversion device when the duty cycle is greater than 0.5, the schematic diagram of the working modes. As Figure 5 shown, the period from time t0 to t8 is a working cycle. The first switch Q1 and the fourth switch Q4 operate with a duty cycle D. The switching timings of the first switch Q1 and the fourth switch Q4 are out of phase by 180 degrees. The control signals of the third switch Q3 and the fourth switch Q4 are complementary, and the control signals of the sixth switch Q6 and the first switch Q1 are complementary. The second switch Q2 and the sixth switch Q6 are synchronously turned on and off, and the fifth switch Q5 and the third switch Q3 are synchronously turned on and off. To achieve the aforementioned switching actions, in some embodiments, the switch tubes can be controlled by the first control signal PWM1, the second control signal PWM2, the third control signal PWM3, and the fourth control signal PWM4 output by the controller. The relationships between the control signals are the same as those described above, so they will not be elaborated here. However, in the case where the duty cycle is greater than 0.5, the first switch Q1 is controlled by the first control signal PWM1, the fourth switch Q4 is controlled by the second control signal PWM2, the second switch Q2 and the sixth switch Q6 are controlled by the third control signal PWM3, and the third switch Q3 and the fifth switch Q5 are controlled by the fourth control signal PWM4.

[0093] Please refer to Figure 5 and Figure 6A , during the period from time t0 to t1, the first and fourth switch tubes Q1 and Q4 are in the conducting state, and the second, third, fifth, and sixth switch tubes Q2, Q3, Q5, and Q6 are in the non-conducting state. The corresponding working mode is as Figure 6AAs shown. At this time, the input terminal transmits power to the output terminal through the primary side windings T1A and T2A and the second capacitor C2. The exciting currents flowing through the first transformer T1 and the second transformer T2 increase in the positive direction at the same time, and the two transformers store energy. In this case, the exciting inductances can be respectively equivalent to being connected in parallel with the primary side windings T1A and T2A. During the period from time t0 to t1, the power conversion device 1 receives power from the input terminal and outputs power to the output terminal.

[0094] Please refer to Figure 5 and Figure 6B , during the period from time t2 to t3, the first, third, and fifth switching tubes Q1, Q3, and Q5 are in the conducting state, and the second, fourth, and sixth switching tubes Q2, Q4, and Q6 are in the off state. The corresponding operating mode is as Figure 6B shown. Similar to the Figure 4A shown operating mode, in the Figure 6B shown operating mode, the voltage drop across the first transformer T1 is (Vin - 2 * N * Vo) / 2, the exciting current flowing through the first transformer T1 increases in the positive direction, and the first transformer T1 stores energy. The volt-second of the first transformer T1 is (Vin - 2 * N * Vo) * D * Ts. The exciting current flowing through the second transformer T2 decreases in the positive direction, and the second transformer T2 releases energy to the output terminal. Accordingly, during the period from time t2 to t3, the power conversion device 1 receives power from the input terminal and stores energy in the first transformer T1 while outputting power to the output terminal. Different from the Figure 4A shown operating mode, in the Figure 6B shown operating mode, the first capacitor C1 and the second capacitor C2 are used to provide a path for the exciting currents of the two transformers and absorb a part of the energy.

[0095] Please refer to Figure 5 and Figure 6C , during the period from time t4 to t5, the first and fourth switching tubes Q1 and Q4 are in the conducting state, and the second, third, fifth, and sixth switching tubes Q2, Q3, Q5, and Q6 are in the off state. The corresponding operating mode is as Figure 6C shown. Since the Figure 6C shown operating mode is the same as the Figure 6A shown operating mode, it will not be elaborated here.

[0096] Please refer to Figure 5 and Figure 6D , during the period from time t6 to t7, the first, third, and fifth switching tubes Q1, Q3, and Q5 are in the off state, and the second, fourth, and sixth switching tubes Q2, Q4, and Q6 are in the conducting state. The corresponding operating mode is as Figure 6D shown. Similar to the Figure 4C shown operating mode, in the Figure 6DIn the shown operating mode, the voltage drop across the second transformer T2 is (Vin - 2 * N * Vo) / 2. The exciting current flowing through the second transformer T2 increases positively, and the second transformer T2 stores energy. The volt-second of the second transformer T2 is (Vin - 2 * N * Vo) * D * Ts. The exciting current flowing through the first transformer T1 decreases positively, and the exciting inductor of the first transformer T1 releases energy to the output terminal. Accordingly, during the time period from t6 to t7, the power conversion device 1 receives power from the input terminal, and while outputting power to the output terminal, stores energy in the second transformer T2. Different from Figure 4C the shown operating mode, in Figure 6D the shown operating mode, the first capacitor C1 and the second capacitor C2 are used to provide a path for the exciting currents of the two transformers and absorb part of the energy.

[0097] In Figure 5 it, the time periods from t1 to t2, from t3 to t4, from t5 to t6, and from t7 to t8 are all dead times. During the dead times, commutation is carried out only through the parasitic diodes or parasitic capacitors of the switching tubes.

[0098] When the duty cycle D is greater than 0.5, within one switching period Ts, that is, Figure 5 during the time period from t0 to t8 in it, the volt-seconds of the exciting current flowing through the first transformer T1 during the positive increasing stage and the positive decreasing stage are (Vin - 2 * N * Vo) * 0.5 * D * Ts and Vo * (1 - D) * Ts respectively. Through the turns ratio relationship of the transformer and the volt-second balance of the magnetic element, we can obtain: [Vin - (N + 1) * Vo] * D * Ts / (1 + N) = Vo * (1 - D) * Ts. After simplification, the expression of the output voltage Vo can be obtained: Vo = Vin * D / (1 + N).

[0099] It can be seen therefrom that the present disclosure realizes the regulation of the output voltage Vo by adjusting the turn ratios of the primary side windings T1A and T2A and the secondary side windings T1B and T2B and the duty cycle D of the switching transistors. Under the condition of the same input and output voltages, compared with the traditional BUCK converter, the inductor volt-seconds of the transformers T1 and T2 of the power conversion device 1 of the present disclosure are low, so the voltage stress of the switching transistors is low, and the volume of the magnetic components is small. Therefore, magnetic components with a smaller volume and switching transistors with a lower breakdown voltage can be correspondingly adopted, thereby reducing costs and increasing the power density. Furthermore, compared with the traditional switched full-bridge converter, the number of turns of the primary side windings T1A and T2A of the transformers T1 and T2 of the power conversion device 1 of the present disclosure is less, and the secondary side windings T1B and T2B carry continuous current when the duty cycle D is less than or equal to 0.5, and the effective value of the current is smaller, which can reduce the number of layers of the PCB. Moreover, the duty cycle D of the switching transistors of the present disclosure can be greater than 0.5, and a wider input and output voltage range can be obtained. In addition, the present disclosure can adjust N in the transformer turn ratio according to different input and output voltage ranges, and the design is relatively flexible and convenient.

[0100] In some embodiments, as Figure 7A shown, the power conversion device 1 further includes a clamping circuit, and the clamping circuit includes a clamping capacitor, a first diode D1, a second diode D2, a third diode D3, and a fourth diode D4. One end of the clamping capacitor is electrically connected to a fixed potential point, and the other end of the clamping capacitor is electrically connected to a clamping node P, where the fixed potential point can be the positive input terminal Vin+, the negative input terminal Vin-, the positive output terminal Vo+, or the positive terminal of an external voltage source. The anode and cathode of the first diode D1 are electrically connected to the clamping node P and the first primary side node A1, respectively. The anode and cathode of the second diode D2 are electrically connected to the clamping node P and the second primary side node A2, respectively. The anode and cathode of the third diode D3 are electrically connected to the first secondary side node B1 and the clamping node P, respectively. The anode and cathode of the fourth diode D4 are electrically connected to the second secondary side node B2 and the clamping node P, respectively.

[0101] In Figure 7AIn the illustrated embodiment, the clamping capacitor includes a third capacitor C3 and a fourth capacitor C4. Two ends of the third capacitor C3 are electrically connected to the positive input terminal Vin+ and the clamping node P respectively, and two ends of the fourth capacitor C4 are electrically connected to the negative input terminal Vin- and the clamping node P respectively. The voltage on the clamping node P is equal to 0.5*Vin. From the foregoing description of the operating modes, it can be known that the voltages on the first and second primary-side nodes A1 and A2 jump between 0.5*Vin and Vin, and the voltages on the first and second secondary-side nodes B1 and B2 jump between 0 and 0.5*Vin. Therefore, connecting the first and second primary-side nodes A1 and A2 and the first and second secondary-side nodes B1 and B2 to the clamping node P through diodes can effectively control the voltage stress of the switching transistors without generating additional losses.

[0102] In addition, the actual form of the clamping capacitor is not limited to Figure 7A that shown. In some other embodiments, as Figure 7B shown, the clamping capacitor only includes the third capacitor C3. Two ends of the third capacitor C3 are electrically connected to the positive input terminal Vin+ and the clamping node P respectively, and the fixed potential point connected to the third capacitor C3 can also be replaced with the positive output terminal Vo+ or the positive terminal of an external voltage source as the case may be. Also, in some other embodiments, as Figure 7C shown, the clamping capacitor only includes the fourth capacitor C4. Two ends of the fourth capacitor C4 are electrically connected to the negative input terminal Vin- and the clamping node P respectively, and the fixed potential point connected to the fourth capacitor C4 can also be replaced with the positive output terminal Vo+ or the positive terminal of an external voltage source as the case may be.

[0103] In some embodiments, as Figure 8 shown, the power conversion device 1 further includes a fifth capacitor C5, a sixth capacitor C6, and a seventh capacitor C7. The seventh capacitor C7 is connected in series between the two primary-side windings T1A and T2A as a DC-blocking capacitor to block the DC component in the current, thereby suppressing the bias magnetic field caused by the DC component. The first ends of the first capacitor C1 and the fifth capacitor C5 are both electrically connected to the positive input terminal Vin+, and the second ends of the first capacitor C1 and the fifth capacitor C5 are electrically connected to the two ends of the seventh capacitor C7 (i.e., nodes A4 and A5) respectively. The first ends of the second capacitor C2 and the sixth capacitor C6 are both electrically connected to the positive output terminal Vo+, and the second ends of the second capacitor C2 and the sixth capacitor C6 are electrically connected to the two ends of the seventh capacitor C7.

[0104] In some embodiments, on the basis of the circuit shown in Figure 8 , the foregoing clamping circuit can be further incorporated. Taking Figure 9 as an example, it combines the clamping circuit shown in Figure 7A with the circuit shown in Figure 8 . In Figure 9Among them, based on the voltage division principle of capacitors, the voltage stresses of the first, second, fourth, and fifth switching transistors Q1, Q2, Q4, and Q5 can be controlled within the rated voltage range by designing the capacitance values of capacitors C1 - C6.

[0105] In the foregoing embodiments, the first transformer T1 and the second transformer T2 may be two discrete transformers, with the primary side winding T1A coupled to the secondary side winding T1B, and the primary side winding T2A coupled to the secondary side winding T2B. However, in some embodiments, as Figure 10A shown, the four windings (such as the primary side windings T1A and T2A and the secondary side windings T1B and T2B) are all mutually coupled. To achieve Figure 10A this winding coupling method, the windings of the first transformer T1 and the second transformer T2 can be wound on the same magnetic core assembly using magnetic integration technology. The following takes Figure 10B and Figure 10C as examples to illustrate two possible winding methods and magnetic core assembly structures of the windings respectively, but the actual forms are not limited thereto.

[0106] As Figure 10B shown, the first transformer T1 and the second transformer T2 adopt a magnetic integration scheme to form an integrated magnetic component. This integrated magnetic component includes a magnetic core assembly 2, primary side windings T1A and T2A, and secondary side windings T1B and T2B. The magnetic core assembly 2 includes two winding posts 21, 22 and a middle post 23. The primary side winding T1A and the secondary side winding T1B of the first transformer T1 are wound on the winding post 21, and the primary side winding T2A and the secondary side winding T2B of the second transformer T2 are wound on the winding post 22. The length of the air gap d1 on the middle post 23 is greater than the lengths of the air gaps (not shown) on the winding posts 21 and 22. The winding directions of the windings can make the DC magnetic fluxes on the two winding posts 21 and 22 superimpose on the middle post 23, and the AC magnetic fluxes on the two winding posts 21 and 22 are partially or completely cancelled on the middle post 23. Thereby, the ripple current flowing through the windings can be significantly reduced.

[0107] As Figure 10CAs shown, the first transformer T1 and the second transformer T2 adopt a magnetic integration scheme to form an integrated magnetic component. The integrated magnetic component includes a magnetic core assembly 3, a primary side winding T1A and T2A, and a secondary side winding T1B and T2B. The magnetic core assembly 3 includes two winding columns 31, 32 and two side columns 33. The primary side winding T1A and the secondary side winding T1B of the first transformer T1 are wound around the winding column 31, and the primary side winding T2A and the secondary side winding T2B of the second transformer T2 are wound around the winding column 32. The lengths of the air gaps d2 on the two side columns 33 are greater than the lengths of the air gaps (not shown) on the two winding columns 31 and 32. The winding directions of the windings can cause the DC magnetic fluxes on the two winding columns 31 and 32 to be superimposed on the two side columns 33, and the AC magnetic fluxes on the two winding columns 31 and 32 to be partially or completely cancelled on the two side columns 33. Thus, the ripple current flowing through the windings can be significantly reduced. Compared with Figure 10B the state shown, in Figure 10C the state shown, the distance d4 between the two winding columns 31 and 32 is shorter than Figure 10B the distance d3 between the winding columns 21 and 22 shown (here, the distance between the two winding columns is equal to the distance between the central positions of the two winding columns). The AC loop of the circuit on the secondary side winding T1B and T2B sides (i.e., the loop formed by the secondary side windings T1B and T2B and the output end of the power conversion device 1) is shorter, thereby reducing the leakage inductance of the transformers T1 and T2 and reducing the winding loss.

[0108] In some embodiments, under the condition that the duty cycle D is less than or equal to 0.5, Figure 1 the first capacitor C1 and the second capacitor C2 in Figure 11 can also be omitted from the power conversion device 1, and the corresponding circuit structure is as shown in Figure 11 . And the circuit structure in

[0109] can achieve the same function through the aforementioned control method, which will not be elaborated here. Figure 1 In some embodiments, the primary side windings T1A and T2A in Figure 12 can be swapped. Specifically, as shown in Figure 1 , the primary side winding T1A of the first transformer T1 is electrically connected between the second primary side node A2 and the third primary side node A3, and the primary side winding T2A of the second transformer T2 is electrically connected between the first primary side A1 and the third primary side node A3. The set positions of the secondary side windings T1B and T2B are the same as those in Figure 12In the power conversion device 1 shown, when the duty cycle D is less than or equal to 0.5, during the conduction period of the first switching transistor Q1 and the conduction period of the second switching transistor Q2, the potential of the third primary side node A3 becomes Vin - N*Vo; and when the duty cycle D is greater than 0.5, during the conduction period of the second switching transistor Q2 and the conduction period of the third switching transistor Q3, the potential of the third primary side node A3 becomes Vin - N*Vo. Correspondingly, Figure 13 shows combining Figure 11 the circuit structure of the power conversion device shown with the aforementioned DC blocking capacitor, Figure 14 shows combining Figure 11 the circuit structure of the power conversion device shown with the aforementioned clamping circuit, and its principle and effect are the same as those described above, so they will not be elaborated here.

[0110] Figure 15 is a schematic flow chart of the control method of the power conversion device according to a preferred embodiment of the present disclosure. The control method of the present disclosure can be applied to control the power conversion devices in the aforementioned embodiments. As Figure 15 shown, the control method includes steps S1, S2, S3, and S4.

[0111] In step S1, control the first switching transistor Q1 and the fourth switching transistor Q4 to operate with a duty cycle D, and control the switching timings of the first switching transistor Q1 and the fourth switching transistor Q4 to be out of phase by 180 degrees with each other.

[0112] In step S2, control the control signals of the third switching transistor Q3 and the fourth switching transistor Q4 to be complementary, and control the control signals of the sixth switching transistor Q6 and the first switching transistor Q1 to be complementary.

[0113] In step S3, that is, when the duty cycle D is less than or equal to 0.5, control the second switching transistor Q2 and the fourth switching transistor Q4 to conduct and turn off synchronously, and control the fifth switching transistor Q5 and the first switching transistor Q1 to conduct and turn off synchronously.

[0114] In step S4, that is, when the duty cycle D is greater than 0.5, control the second switching transistor Q2 and the sixth switching transistor Q6 to conduct and turn off synchronously, and control the fifth switching transistor Q5 and the third switching transistor Q3 to conduct and turn off synchronously.

[0115] In some embodiments, the control method further includes sub-steps: generating a first control signal PWM1, a second control signal PWM2, a third control signal PWM3, and a fourth control signal PWM4 to control the switching transistors in the first and second arms 12 and 13, wherein the first and second control signals PWM1 and PWM2 have a duty cycle D and are out of phase by 180 degrees with each other, the third control signal PWM3 is complementary to the first control signal PWM1, and the fourth control signal PWM4 is complementary to the second control signal PWM2.

[0116] Further, the control method further includes sub-steps: controlling the first switch Q1, the fourth switch Q4, the sixth switch Q6, and the third switch Q3 by using the first control signal PWM1, the second control signal PWM2, the third control signal PWM3, and the fourth control signal PWM4 respectively; when the duty cycle D is less than or equal to 0.5, controlling the second switch Q2 and the fifth switch Q5 by using the second control signal PWM2 and the first control signal PWM1 respectively; and when the duty cycle D is greater than 0.5, controlling the second switch Q2 and the fifth switch Q5 by using the third control signal PWM3 and the fourth control signal PWM4 respectively.

[0117] In summary, the present disclosure provides a control method for a power conversion device, which realizes output voltage regulation by adjusting the turns ratio of the primary side winding and the secondary side winding and the duty cycle of the switch tube. Under the same input and output voltage conditions, compared with the traditional BUCK converter, the inductor volt-second of the transformer of the power conversion device of the present disclosure is low, so the voltage stress of the switch tube is low, and the volume of the magnetic component is small. Therefore, magnetic components with a smaller volume and switch tubes with a lower breakdown voltage can be correspondingly used, thereby reducing costs and increasing the power density. Furthermore, compared with the traditional full-bridge converter, the number of turns of the primary side winding of the transformer of the power conversion device of the present disclosure is less, and the secondary side winding flows through a continuous current when the duty cycle is less than or equal to 0.5, and the effective value of the current is smaller, which can reduce the number of layers of the PCB. Moreover, the duty cycle of the switch tube of the present disclosure can be greater than 0.5, and a wider input and output voltage range can be obtained. In addition, the present disclosure can adjust N in the transformer turns ratio according to different input and output voltage ranges, and the design is relatively flexible and convenient.

[0118] It should be noted that the above are only preferred embodiments proposed for explaining the present disclosure. The present disclosure is not limited to the described embodiments, and the scope of the present disclosure is determined by the appended claims. And the present disclosure can be variously modified by those skilled in the art, but all do not depart from what the appended claims intend to protect.

Claims

1. A control method, which is configured to control a power conversion device. The power conversion device includes a first bridge arm, a second bridge arm, and two transformers. The first bridge arm and the second bridge arm are connected in parallel. The first bridge arm includes a first switch tube, a second switch tube, and a third switch tube connected in series. There is a first primary side node and a first secondary side node between the first switch tube, the second switch tube, and the third switch tube respectively. The second bridge arm includes a fourth switch tube, a fifth switch tube, and a sixth switch tube connected in series. There is a second primary side node and a second secondary side node between the fourth switch tube, the fifth switch tube, and the sixth switch tube respectively. Each of the transformers includes a primary side winding and a secondary side winding. The two primary side windings of the two transformers are connected in series between the first primary side node and the second primary side node. The two secondary side windings of the two transformers are connected in series between the first secondary side node and the second secondary side node. The control method includes: Controlling the first and fourth switch tubes to operate at a duty cycle, and controlling the switching timings of the first and fourth switch tubes to be out of phase by 180 degrees with respect to each other; Controlling the control signals of the third switch tube and the fourth switch tube to be complementary, and controlling the control signals of the sixth switch tube and the first switch tube to be complementary; and When the duty cycle is less than or equal to 0.5, controlling the second switch tube and the fourth switch tube to conduct and turn off synchronously, and controlling the fifth switch tube and the first switch tube to conduct and turn off synchronously. When the duty cycle is greater than 0.5, controlling the second switch tube and the sixth switch tube to conduct and turn off synchronously, and controlling the fifth switch tube and the third switch tube to conduct and turn off synchronously.

2. The control method according to claim 1, further including sub-steps: generating a first control signal, a second control signal, a third control signal, and a fourth control signal to control the switch tubes in the first and second bridge arms, where the first and second control signals have the duty cycle and are out of phase by 180 degrees with respect to each other, the third control signal is complementary to the first control signal, and the fourth control signal is complementary to the second control signal.

3. The control method according to claim 2, further including sub-steps: using the first control signal, the second control signal, the third control signal, and the fourth control signal to control the first switch tube, the fourth switch tube, the sixth switch tube, and the third switch tube respectively.

4. The control method according to claim 2, further including sub-steps: when the duty cycle is less than or equal to 0.5, using the second control signal and the first control signal to control the second switch tube and the fifth switch tube respectively.

5. The control method according to claim 2, further including sub-steps: when the duty cycle is greater than 0.5, using the third control signal and the fourth control signal to control the second switch tube and the fifth switch tube respectively.

6. The control method according to claim 1, wherein, The power conversion device further includes an input terminal, where the input terminal has a positive input end and a negative input end. The first bridge arm and the second bridge arm are connected in parallel to the input terminal.

7. The control method according to claim 6, wherein, The power conversion device further includes an output capacitor. There is a third secondary side node between the two secondary side windings of the two transformers, and a third primary side node between the two primary side windings. Two ends of the output capacitor are electrically connected to an output positive terminal and an output negative terminal respectively. The output positive terminal is coupled to the third secondary side node, and the output negative terminal is coupled to the input negative terminal.

8. The control method according to claim 7, wherein, The two transformers are a first transformer and a second transformer respectively. The primary side winding of the first transformer is electrically connected between the first primary side node and the third primary side node. The secondary side winding of the first transformer is electrically connected between the second secondary side node and the third secondary side node. The primary side winding of the second transformer is electrically connected between the second primary side node and the third primary side node. The secondary side winding of the second transformer is electrically connected between the first secondary side node and the third secondary side node.

9. The control method according to claim 7, wherein, The two transformers are a first transformer and a second transformer respectively. The primary side winding of the first transformer is electrically connected between the second primary side node and the third primary side node. The secondary side winding of the first transformer is electrically connected between the second secondary side node and the third secondary side node. The primary side winding of the second transformer is electrically connected between the first primary side node and the third primary side node. The secondary side winding of the second transformer is electrically connected between the first secondary side node and the third secondary side node.

10. The control method according to claim 1, wherein, The third and sixth switching tubes are diodes.

11. The control method according to claim 8 or 9, wherein, The turns ratio of the primary side winding of the first transformer, the primary side winding of the second transformer, the secondary side winding of the first transformer, and the secondary side winding of the second transformer is N:N:1:1, where N is a positive integer.

12. The control method according to claim 11, wherein, An output voltage and an input voltage of the power conversion device have the following relationship: Vo = Vin·D / (1 + N) where Vo is the output voltage, Vin is the input voltage, and D is the duty cycle.

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