A linearized PGC demodulation signal processing method based on hybrid phase modulation

Abstract

The present invention discloses a linearized PGC demodulation signal processing method based on hybrid phase modulation. On the basis of sinusoidal phase modulation, triangular phase modulation is introduced to perform hybrid modulation on the laser phase modulation interferometer to obtain interference signals, which are then mixed with the carrier signal and its second-order harmonic signal to obtain a pair of phase quadrature signals; synchronous differential cross-multiplication and inverse tangent processing are performed on the quadrature signals to obtain phase signals that change with triangular phase modulation, and the demodulation phase is linearly corrected in real time using the obtained demodulation π integer phase without nonlinear error and the corresponding demodulation phase, and the demodulation nonlinear error is directly corrected by combining the π integer phase and the corrected demodulation phase. The present invention eliminates the influence of periodic nonlinear errors caused by phase modulation depth drift and carrier phase delay through linearized phase demodulation, and realizes high-precision phase demodulation.

Description

Technical Field

[0001] The invention relates to a phase demodulation method in the technical field of laser interferometry, and in particular to a linearized PGC demodulation signal processing method based on hybrid phase modulation. Background Art

[0002] Phase generated carrier (PGC) demodulation technology is widely used in the detection and acquisition of interference signal phase due to its advantages such as strong stability, good real-time performance, high sensitivity, and high signal-to-noise ratio. Traditional PGC demodulation generally includes differential cross-multiplication (PGC-DCM) algorithm and inverse tangent (PGC-Arctan) algorithm. Thanks to the PGC-DCM demodulation principle, its demodulation phase is linear, but the demodulation result will be affected by laser intensity disturbance, phase modulation depth drift, carrier phase delay, etc. Laser intensity fluctuations are caused by the intensity fluctuations of the laser source itself and environmental interference. The phase modulation depth is determined by the amplitude of the modulated signal and the half-wave voltage of the phase modulator. The carrier phase delay is caused by the time delay of the optical path and circuit propagation. Laser intensity fluctuations, phase modulation depth deviation from 2.37rad, and carrier phase delay will all lead to linear deviations in the demodulation phase.

[0003] Compared with PGC-DCM demodulation, PGC-Arctan demodulation is insensitive to laser intensity fluctuations, but it is still affected by phase modulation depth drift and carrier phase delay. Phase modulation depth deviation from 2.63rad and carrier phase delay will cause demodulation phase harmonic distortion. In most cases, phase modulation depth drift and carrier phase delay are unavoidable, resulting in periodic nonlinear errors in demodulation phase. Summary of the invention

[0004] In order to solve the problems existing in the background technology, the present invention discloses a linearized PGC demodulation signal processing method based on hybrid phase modulation, which eliminates or reduces the linear deviation and nonlinear error of PGC phase demodulation and improves the accuracy of laser phase modulation interferometry measurement.

[0005] The present invention introduces triangular phase modulation for mixed modulation based on sinusoidal phase modulation, drives the phases of PGC-Arctan and PGC-DCM demodulation to change simultaneously, obtains the demodulated π integer phase and the corresponding demodulated phase without nonlinear error in real time, uses the obtained phases to perform real-time linear correction on the demodulated phase, and finally combines the demodulated π integer phase and the corrected demodulated phase to directly correct the demodulation nonlinear error, thereby realizing high-precision phase demodulation.

[0006] The technical solution adopted by the present invention to solve its technical problem is:

[0007] 1) A direct digital frequency synthesizer and a triangle wave modulation signal generator are used to generate a mixed modulation signal V(t) of a sine wave and a triangle wave. The mixed modulation signal V(t) is expressed as follows:

[0008] V(t)=A c cos(ω c t)+A t Tri(ω t t)

[0009] Among them, A c and A t are the amplitudes of the sine wave and triangle wave signals, ω c is the frequency of the sinusoidal signal, ω t is the frequency of the triangular wave signal, t represents the time; Tri(ω t t) is a normalized triangle wave signal in the range [-1, 1];

[0010] The mixed modulation signal V(t) is amplified and applied to an external laser phase modulation interferometer to modulate the interference signal in the laser phase modulation interferometer. The laser phase modulation interferometer detects the displacement of the object to be measured to obtain a PGC interference signal. The PGC interference signal output by the laser phase modulation interferometer is sampled to collect the interference signal I(t), which is expressed as follows:

[0011]

[0012] z t =πG1A t / V π , z c =πG1A c / V π

[0013] Among them, S0 and S1 are the DC and AC component amplitudes respectively, is the phase to be measured, z t and z c are the phase modulation depths of the triangle wave and sine wave output by the triangle wave modulation signal generator and the direct digital frequency synthesizer, θ is the carrier phase delay, A t and A c are the amplitudes of the triangle wave and the sine wave respectively, G1 is the voltage amplifier gain, V π is the half-wave voltage of the electro-optic phase modulator in the laser phase modulation interferometer, J m (z c ) is the phase modulation depth z of the sine wave c The m-order Bessel function of ;

[0014] In a specific implementation, a direct digital frequency synthesizer is used to generate a sine wave carrier baseband signal cos(ωc t), the triangle wave modulation signal generator generates a triangle wave signal Tri(ω t t), the carrier baseband signal cos(ω c t) and the triangular wave signal Tri(ω t t) to obtain a mixed modulation signal V(t) of a sine wave and a triangle wave.

[0015] In the specific implementation, the mixed modulation signal V(t) is input into the electro-optic phase modulator of the laser phase modulation interferometer to modulate the interference signal to form a modulated interference signal, and the displacement of the object to be measured is detected, and the detection feedback obtains the PGC interference signal.

[0016] 2) The interference signal I(t) is respectively combined with the carrier baseband signal cos(ω c t) and the second harmonic signal cos2(ω c t) are multiplied by a multiplier and then mixed by a low-pass filter to obtain a pair of orthogonal phase signals, which are divided into orthogonal signals I1(t) and I2(t), as shown below:

[0017]

[0018] Where, B is the amplitude of the carrier base frequency signal and the second harmonic signal, and LPF[] is the low-pass filtering operation;

[0019] 3) Perform arc tangent (PGC-Arctan) and differential cross multiplication (PGC-DCM) processing on the orthogonal signals I1(t) and I2(t) at the same time:

[0020] After inverse tangent calculation and phase unwrapping, the following inverse tangent phase is obtained:

[0021]

[0022] v1=J1(z c )cosθ,v2=J2(z c )cos2θ

[0023] Where v1 and v2 are the phase modulation depth z of the sine wave respectively. c and the carrier phase delay θ determines the proportionality coefficient, is the nonlinear error at time t; represents the arc tangent phase at time t;

[0024] At the same time, after differential cross multiplication (PGC-DCM) processing, the differential cross multiplication phase obtained is expressed as:

[0025]

[0026] Where G2 is the integrator gain; represents the differential cross-multiplication phase at time t;

[0027] 4) At the initial moment, the arc tangent phase is acquired Sum differential cross-multiplication phase And at the initial moment, the arc tangent phase The integer phase counter of the π cycle is set to N, and the inverse tangent phase Sum differential cross-multiplication phase The phase signal is synchronized and matched, and the inverse tangent phase Sum differential cross-multiplication phase Will follow the triangular wave signal Tri(ω t t) Changes:

[0028] When the inverse tangent phase When the integer phase count of the π cycle becomes N+1 at time t1, the inverse tangent phase at time t1 is recorded. Sum differential cross-multiplication phase

[0029] Similarly, when the inverse tangent phase When the integer phase count of the π cycle becomes N+2 at time t2, the inverse tangent phase at time t2 is recorded. Sum differential cross-multiplication phase

[0030] After obtaining the above phase value, the inverse tangent phase is calculated as follows Perform linear correction to obtain the demodulated phase

[0031] The linear correction and The π integer phase is combined to obtain the linearized correction phase. As shown below:

[0032]

[0033] Finally, the triangular modulation phase is removed by mean filtering to obtain the final demodulated phase

[0034] The demodulated phase Used to characterize the displacement of an object in a laser phase modulation interferometer.

[0035] In a specific implementation, the phase modulation depth z c It does not have to be equal to 2.63rad, J1(z c ) does not have to be equal to J2(z c ).

[0036] In the above 4), the triangular wave Tri (ω t t) Arc tangent phase of modulation The phase change is greater than 2π, so that the triangle wave phase modulation depth z t must be greater than π to achieve the inverse tangent phase The integer phase count of the π cycle is in the triangle wave signal Tri(ω t t) can become N+1 and N+2 during the change process.

[0037] The method calculation process adopts a phase demodulation processing system, which specifically includes a first adder, a triangular wave modulation signal generator, a first differential processing module, a second differential processing module, a first multiplier, a second multiplier, a first subtractor, a second subtractor, an integral processing module, a third multiplier, a phase averaging module, a direct digital frequency synthesizer, a fourth multiplier, a fifth multiplier, a first low-pass filter, a second low-pass filter, a phase synchronization matching module, an inverse tangent calculation module, a phase unwrapping module, a third subtractor, a divider, a second adder, and a phase averaging module. The interference signal I(t) is connected to the input of the fourth multiplier and the fifth multiplier; the output of the triangle wave modulation signal generator is connected to the input of the first adder, the direct digital frequency synthesizer is connected to the first adder and the input of the fourth multiplier and the fifth multiplier, and the first adder outputs a mixed modulation signal V(t); the output of the fourth multiplier and the fifth multiplier are connected to the input of the first low-pass filter and the second low-pass filter respectively, and the output of the first low-pass filter is connected to the second differential processing module, the first multiplier and the inverse tangent calculator respectively. The output end of the second low-pass filter is connected to the input end of the first differential processing module, the second multiplier and the inverse tangent calculation module respectively; the output ends of the second differential processing module and the first differential processing module are respectively connected to the input ends of the second multiplier and the first multiplier, the output ends of the second multiplier and the first multiplier are connected to the input end of the first subtractor, the output end of the first subtractor is connected to the input end of the integral processing module, and the output end of the integral processing module is connected to the input end of the phase synchronization matching module; the output end of the inverse tangent calculation module is connected to the input end of the phase unwrapping module, and the output end of the phase unwrapping module is connected to the input end of the phase synchronization matching module; the output end of the phase synchronization matching module is respectively connected to the input ends of the second subtractor, the third subtractor and the second adder, the output end of the second subtractor is connected to the input end of the third multiplier and multiplied by π, and the output ends of the third multiplier and the third subtractor are connected to the input end of the divider; the output end of the divider is connected to the input end of the second adder, the output end of the second adder is connected to the input end of the phase averaging module, and the output end of the phase averaging module outputs the phase demodulation result.

[0038] The PGC interference signal is derived from a laser phase modulation interferometer and is a detection signal output by the laser phase modulation interferometer.

[0039] The present invention eliminates the influence of periodic nonlinear errors caused by phase modulation depth drift and carrier phase delay through linear phase demodulation, thereby realizing high-precision phase demodulation.

[0040] The present invention has the following beneficial effects:

[0041] 1) The present invention adopts a mixed phase modulation of sine wave and triangle wave, utilizes the PGC-Arctan demodulation π integer phase without nonlinear error and the corresponding PGC-DCM demodulation phase, performs real-time linear correction on the PGC-DCM demodulation phase, combines the π integer phase of PGC-Arctan demodulation and the corrected PGC-DCM demodulation phase, directly corrects the nonlinear error of PGC-Arctan demodulation, and improves the phase demodulation accuracy

[0042] 2) The present invention utilizes a linearized phase demodulation method to eliminate the nonlinear effects caused by phase modulation depth drift and carrier phase delay, without the need to obtain the definite values ​​of phase modulation depth and carrier phase delay through complex calculations, or to compensate for nonlinear errors through ellipse fitting using a large amount of pre-data.

[0043] 3) The present invention adopts hybrid phase modulation and demodulation technology, and the phase measurement has the advantages of strong anti-interference ability and large dynamic measurement range. BRIEF DESCRIPTION OF THE DRAWINGS

[0044] Figure 1 It is a structure and principle block diagram of the method of the present invention.

[0045] In the figure: 1, a first adder, 2, a triangle wave modulation signal generator, 3, a first differential processing module, 4, a second differential processing module, 5, a first multiplier, 6, a second multiplier, 7, a first subtractor, 8, a second subtractor, 9, an integral processing module, 10, a third multiplier, 11, a phase averaging module, 12, a direct digital frequency synthesizer, 13, a fourth multiplier, 14, a fifth multiplier, 15, a first low-pass filter, 16, a second low-pass filter, 17 a phase synchronization matching module, 18, an inverse tangent calculation module, 19, a phase unwrapping module, 20, a third subtractor, 21, a divider, 22, a second adder. DETAILED DESCRIPTION

[0046] The present invention will be further described below in conjunction with the accompanying drawings and embodiments.

[0047] like Figure 1 As shown, the specific implementation adopts the following phase demodulation processing system:

[0048] The system includes a first adder 1, a triangle wave modulation signal generator 2, a first differential processing module 3, a second differential processing module 4, a first multiplier 5, a second multiplier 6, a first subtractor 7, a second subtractor 8, an integral processing module 9, a third multiplier 10, a phase averaging module 11, a direct digital frequency synthesizer 12, a fourth multiplier 13, a fifth multiplier 14, a first low-pass filter 15, a second low-pass filter 16, a phase synchronization matching module 17, an inverse tangent calculation module 18, a phase unwrapping module 19, a third subtractor 20, a divider 21, and a second adder 22.

[0049] The interference signal I(t) is connected to the input ends of the fourth multiplier 13 and the fifth multiplier 14; the output of the triangle wave modulation signal generator 2 is connected to the input end of the first adder 1, the direct digital frequency synthesizer 12 is connected to the input ends of the first adder 1, the fourth multiplier 13 and the fifth multiplier 14, and the first adder 1 outputs a mixed modulation signal V(t); the output ends of the fourth multiplier 13 and the fifth multiplier 14 are respectively connected to the input ends of the first low-pass filter 15 and the second low-pass filter 16; the output end of the first low-pass filter 15 is respectively connected to the input ends of the second differential processing module 4, the first multiplier 5 and the inverse tangent calculation module 18, and the output end of the second low-pass filter 16 is respectively connected to the input ends of the first differential processing module 3, the second multiplier 6 and the inverse tangent calculation module 18; the output ends of the second differential processing module 4 and the first differential processing module 3 are respectively connected to the input ends of the second multiplier 6 and the first multiplier 5; The output ends of the multiplier 6 and the first multiplier 5 are connected to the input end of the first subtractor 7; the output end of the first subtractor 7 is connected to the input end of the integral processing module 9; the output end of the integral processing module 9 is connected to the input end of the phase synchronization matching module 17; the output end of the inverse tangent calculation module 18 is connected to the input end of the phase unwrapping module 19, and the output end of the phase unwrapping module 19 is connected to the input end of the phase synchronization matching module 17; the output end of the phase synchronization matching module 17 is respectively connected to the input ends of the second subtractor 8, the third subtractor 20 and the second adder 22, the output end of the second subtractor 8 is connected to the input end of the third multiplier 10 and multiplied by π, and the output ends of the third multiplier 10 and the third subtractor 20 are connected to the input end of the divider 21; the output end of the divider 21 is connected to the input end of the second adder 22, and the output end of the second adder 22 is connected to the input end of the phase averaging module 11, and the phase averaging module 11 outputs the phase demodulation result.

[0050] The implementation principle process of the present invention is:

[0051] like Figure 1As shown, the direct digital frequency synthesizer and the triangle wave modulation signal generator output a mixed modulation signal V(t), which is applied to the laser phase modulation interferometer after amplification. The laser phase modulation interferometer outputs a mixed phase modulation interference signal, which is sampled to obtain the interference signal I(t) after filtering and analog-to-digital conversion. The collected interference signal I(t) is expressed as follows:

[0052]

[0053] Among them, S0 and S1 are the DC and AC component amplitudes respectively, is the phase to be measured, z t =πG1A t / V π and z c =πG1A c / V π are the phase modulation depths of the triangle and sine waves, θ is the carrier phase delay, A t and A c are the amplitudes of the triangle wave and the sine wave respectively, G1 is the voltage amplifier gain, V π is the half-wave voltage of the electro-optic phase modulator, J m (z c ) is about z c The m-th order Bessel function of .

[0054] After obtaining the interference signal I(t), a mixing operation is first performed, specifically including: the interference signal I(t) is mixed with the carrier baseband signal cosω generated by the direct digital frequency synthesizer 12 by the fourth multiplier 13 and the fifth multiplier 14. c t and second harmonic signal cos2ω c t, and after filtering by the first low-pass filter 15 and the second low-pass filter 16, the sine component I1(t) and the two cosine components I2(t) are obtained to complete the mixing operation. The calculation process is as follows:

[0055]

[0056] Among them, LPF[] is a low-pass filtering operation.

[0057] After the phase quadrature signals I1(t) and I2(t) are subjected to inverse tangent calculation and phase unfolding, the following inverse tangent phase can be obtained:

[0058]

[0059] Where v1 = J1(z c )cosθ and v2=J2(z c )cos2θ is given by z c and θ determine the proportionality coefficient, is a nonlinear error. According to the frequency of the triangular wave signal, the demodulated phase can be obtained by removing the triangular wave modulation phase through mean filtering.

[0060] At the same time, after the differential cross multiplication and integration (DCM) operation, the phase obtained at this time can be expressed as:

[0061]

[0062] Where G2 is the integrator gain. Similarly, by removing the triangle wave modulation phase through mean filtering, the demodulated phase with linear deviation can be obtained.

[0063] The phase signal obtained and To synchronize The π-cycle phase counter is set to N at the initial moment. and Will follow the triangular wave signal Tri(ω t t) changes, when When the integer phase count of the π cycle becomes N+1 at time t1, and The values ​​of are recorded as and Similarly, when When the integer phase count of the π cycle becomes N+2 at time t2, and The values ​​of are recorded as and To ensure The integer phase count of the π cycle can be changed to N+1 and N+2, through Tri(ω t t) Modulated The phase change must be greater than 2π, so the triangle wave phase modulation depth z t must be greater than π. Once these phase values ​​are obtained, Perform linear correction. and The π integer phase is combined to obtain the linearized correction phase. It can be as follows:

[0064]

[0065] Finally, the triangular modulation phase is removed by mean filtering to obtain the final demodulated phase

[0066] In a specific implementation, the phase modulation depth z cIt does not have to be equal to 2.63, J1(z c ) does not have to be equal to J2(z c ).

[0067] In summary, the present invention adopts mixed phase modulation of sine wave and triangle wave, utilizes PGC-Arctan demodulation π integer phase without nonlinear error and corresponding PGC-DCM demodulation phase, performs real-time linear correction on PGC-DCM demodulation phase, combines the π integer phase of PGC-Arctan demodulation and the corrected PGC-DCM demodulation phase, directly corrects the nonlinear error of PGC-Arctan demodulation, eliminates the nonlinear influence caused by phase modulation depth drift and carrier phase delay, and improves the phase demodulation accuracy.

[0068] The above specific implementation modes are used to explain the present invention rather than to limit the present invention. Any modification and change made to the present invention within the spirit of the present invention and the protection scope of the claims shall fall within the protection scope of the present invention.

Claims

1. A linearized PGC demodulation signal processing method based on hybrid phase modulation, characterized in that The method further comprises: 1) A direct digital frequency synthesizer and a triangle wave modulation signal generator are used to generate a mixed modulation signal V(t) of a sine wave and a triangle wave. The mixed modulation signal V(t) is expressed as follows: V(t)=A c cos(ω c t)+A t Tri(ω t t) Among them, A c and A t are the amplitudes of the sine wave and triangle wave signals, ω c is the frequency of the sinusoidal signal, ω t is the frequency of the triangular wave signal, t represents the time; Tri(ω t t) is a normalized triangle wave signal in the range [-1, 1]; The mixed modulation signal V(t) is amplified and applied to the laser phase modulation interferometer. The PGC interference signal output by the laser phase modulation interferometer is sampled and the interference signal I(t) is collected, which is expressed as follows: With t =πG1A t / V π ,With c =πG1A c / V π Among them, S0 and S1 are the DC and AC component amplitudes respectively, is the phase to be measured, z t and z c are the phase modulation depths of the triangle wave and the sine wave, θ is the carrier phase delay, A t and A c are the amplitudes of the triangle wave and the sine wave respectively, G1 is the voltage amplifier gain, V π is the half-wave voltage of the electro-optic phase modulator in the laser phase modulation interferometer, J m (z c ) is the phase modulation depth z of the sine wave c The m-th order Bessel function of ; 2) The interference signal I(t) is respectively combined with the carrier baseband signal cos(ω c t) and the second harmonic signal cos2(ω c t) are multiplied by a multiplier and then mixed by a low-pass filter to obtain a pair of orthogonal phase signals, which are divided into orthogonal signals I1(t) and I2(t), as shown below: Where, B is the amplitude of the carrier base frequency signal and the second harmonic signal, and LPF[] is the low-pass filtering operation; 3) Perform arc tangent and differential cross multiplication on the orthogonal signals I1(t) and I2(t) at the same time: After inverse tangent calculation and phase unwrapping, the following inverse tangent phase is obtained: v1=J1(z c )cosθ,v2=J2(z c )cos2θ Where v1 and v2 are the phase modulation depth z of the sine wave respectively. c and the carrier phase delay θ determines the proportionality coefficient, is the nonlinear error at time t; represents the arc tangent phase at time t; At the same time, after the differential cross-multiplication process, the differential cross-multiplication phase is expressed as: Where G2 is the integrator gain; represents the differential cross-multiplication phase at time t; 4) At the initial moment, the arc tangent phase is acquired Sum differential cross-multiplication phase And at the initial moment, the arc tangent phase The integer phase counter of the π cycle is set to N, and the inverse tangent phase Sum differential cross-multiplication phase The phase signal is synchronized and matched, and the inverse tangent phase Sum differential cross-multiplication phase Will follow the triangular wave signal Tri(ω t t) Changes: When the inverse tangent phase When the integer phase count of the π cycle becomes N+1 at time t1, the inverse tangent phase at time t1 is recorded. Sum differential cross-multiplication phase Similarly, when the inverse tangent phase When the integer phase count of the π cycle becomes N+2 at time t2, the inverse tangent phase at time t2 is recorded. Sum differential cross-multiplication phase After obtaining the above phase value, the inverse tangent phase is calculated as follows Perform linear correction to obtain the demodulated phase The linear correction and The π integer phase is combined to obtain the linearized correction phase. As shown below: Finally, the triangular modulation phase is removed by mean filtering to obtain the final demodulated phase 2. The method for processing linearized PGC demodulation signals based on hybrid phase modulation according to claim 1, characterized in that: In the above 4), the triangular wave Tri (ω t t) Arc tangent phase of modulation The phase change is greater than 2π, so that the triangle wave phase modulation depth z t must be greater than π to achieve the inverse tangent phase The integer phase count of the π cycle is in the triangle wave signal Tri(ω t t) can become N+1 and N+2 during the change process.

3. The method for processing linearized PGC demodulation signals based on hybrid phase modulation according to claim 1, characterized in that: The method calculation process adopts a phase demodulation processing system, which specifically includes a first adder (1), a triangle wave modulation signal generator (2), a first differential processing module (3), a second differential processing module (4), a first multiplier (5), a second multiplier (6), a first subtractor (7), a second subtractor (8), an integral processing module (9), a third multiplier (10), a phase averaging module (11), a direct digital frequency synthesizer (12), a fourth multiplier (13), a fifth multiplier (14), a first low-pass filter (15), a second low-pass filter (16), a phase synchronization matching module (17), an inverse tangent calculation module (18), a phase unwrapping module (19), a third subtractor (20), a divider (21), and a second adder (22); The interference signal I(t) is connected to the input end of the fourth multiplier (13) and the fifth multiplier (14); the output end of the triangle wave modulation signal generator (2) is connected to the input end of the first adder (1); the direct digital frequency synthesizer (12) is connected to the first adder (1) and the input ends of the fourth multiplier (13) and the fifth multiplier (14); the first adder (1) outputs a mixed modulation signal V(t); the output ends of the fourth multiplier (13) and the fifth multiplier (14) are connected to the first low-pass filter (15) and the second low-pass filter (15) respectively. The input end of the low-pass filter (16) and the output end of the first low-pass filter (15) are respectively connected to the input ends of the second differential processing module (4), the first multiplier (5) and the inverse tangent calculation module (18); the output end of the second low-pass filter (16) is respectively connected to the input ends of the first differential processing module (3), the second multiplier (6) and the inverse tangent calculation module (18); the output ends of the second differential processing module (4) and the first differential processing module (3) are respectively connected to the input ends of the second multiplier (6) and the first multiplier (5); The output ends of the second multiplier (6) and the first multiplier (5) are connected to the input end of the first subtractor (7), the output end of the first subtractor (7) is connected to the input end of the integral processing module (9), the output end of the integral processing module (9) is connected to the input end of the phase synchronization matching module (17); the output end of the inverse tangent calculation module (18) is connected to the input end of the phase expansion module (19), the output end of the phase expansion module (19) is connected to the input end of the phase synchronization matching module (17); the output end of the phase synchronization matching module (17) is respectively connected to the input end of the phase expansion module (19); The first embodiment is a phase averaging module (11) and a phase averaging module (12) which is connected to the input end of the second subtractor (8), the third subtractor (20) and the second adder (22); the output end of the second subtractor (8) is connected to the input end of the third multiplier (10); the output ends of the third multiplier (10) and the third subtractor (20) are connected to the input end of the divider (21); the output end of the divider (21) is connected to the input end of the second adder (22); the output end of the second adder (22) is connected to the input end of the phase averaging module (11); and the output end of the phase averaging module (11) outputs the phase demodulation result.

4. The method for processing linearized PGC demodulation signals based on hybrid phase modulation according to claim 1, characterized in that: The PGC interference signal is derived from a laser phase modulation interferometer and is a detection signal output by the laser phase modulation interferometer.

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