A Method for Implementing Soft Switching in a Wide Voltage Range for a Three-Port Bidirectional DC Converter

Abstract

The present invention is a method for realizing soft switching in a wide voltage range of a three-port bidirectional DC converter. The converter includes a primary full-bridge circuit, a secondary full-bridge circuit, a secondary interleaved boost circuit, and a high-frequency transformer. When used in a power supply system for electric vehicles, the PFC port of the three-port bidirectional DC converter is connected to the AC grid through a pre-stage power factor correction circuit, and the high-voltage port and the low-voltage port are respectively connected to the power battery and the low-voltage battery. Based on the variable DC bus voltage, this method modulates the duty cycle of conduction of two power switches on the lower arm of the secondary interleaved boost circuit, the phase angle difference between the midpoint voltage of the bridge arm of the primary full-bridge circuit and the midpoint voltage of the bridge arm of the secondary full-bridge circuit, and the phase angle difference between the midpoint voltage of the bridge arm of the primary full-bridge circuit and the midpoint voltage of the bridge arm of the secondary interleaved boost circuit, so as to realize soft switching in a wide voltage range in both the charging and discharging modes of the electric vehicle and achieve efficient power transmission of the converter from light load to heavy load.

Description

Technical Field

[0001] The present invention belongs to the technical field of electric vehicle power systems, and particularly relates to a method for realizing soft switching in a wide voltage range of a three-port bidirectional DC converter. Background Art

[0002] The power system of an electric vehicle is composed of different subsystems, including an electric drive subsystem, a charging subsystem, and an auxiliary power supply subsystem. The electric drive subsystem mainly converts the high voltage of the power battery into the voltage required by the drive motor through an inverter to provide power for the vehicle; the charging subsystem mainly converts the alternating current of the power grid into direct current through a power factor correction circuit, and then charges the power battery through a DC converter; the auxiliary power supply subsystem mainly uses a DC converter to convert the high voltage of the power battery into low voltage to supply power for other electrical devices and low-voltage loads of the electric vehicle. The on-vehicle charger only charges the power battery when the electric vehicle stops, and does not play any role when the vehicle is running, becoming the "dead weight" of the system, resulting in a low utilization rate of the on-vehicle charger. Therefore, integrating the on-vehicle charger with the low-voltage converter by sharing some power devices, control circuits, printed circuit boards, and radiators has become an effective way to improve the power density of the power system, reduce the volume of the power system, and realize the lightweight of the power system.

[0003] An electric vehicle usually uses a three-port bidirectional DC converter coupled with a transformer to realize the integration of the on-vehicle charger and the low-voltage converter. The three-port bidirectional DC converter usually uses phase-shift modulation to adjust the transmission power of each port. However, in a wide voltage range, traditional phase-shift modulation will bring a high reactive current circulation, resulting in an increase in the conduction loss of the system; in addition, it will also cause the loss of the soft-switching characteristic of the converter, resulting in an increase in the switching loss, which is not conducive to the high-frequency and high-efficiency operation of the converter.

[0004] The prior art introduces pulse-width modulation on the basis of phase-shift modulation, establishes a loss model of the converter, and uses numerical calculation methods to solve the optimal efficiency operating point of the converter. However, this method is difficult to give an analytical control law, and often needs to establish a complex look-up table, which is difficult to popularize and apply in practice. The paper "Transformer-Coupled Multiport ZVS Bidirectional DC–DC Converter With Wide Input Range" proposes a soft-switching implementation method based on volt-second balance. Although this control method is simple, when the voltage range is wide, it will cause the power switch to bear a high current stress.

[0005] The present invention aims at a three-port bidirectional DC converter for integrated on-board charging and low-voltage conversion of electric vehicles, and proposes a method for realizing soft switching in a wide voltage range. Based on a variable DC bus and a PWM + phase-shift modulation mechanism, the converter achieves zero-voltage turn-on in a wide voltage range, providing an effective solution for the high-frequency and high-efficiency operation of the three-port bidirectional DC converter. Summary of the Invention

[0006] Aiming at the deficiencies of the existing technology, the technical problem to be solved by the present invention is to propose a method for realizing soft switching in a wide voltage range for a three-port bidirectional DC converter.

[0007] The technical solution adopted by the present invention to solve the above technical problem is as follows:

[0008] On the one hand, the present invention provides a method for realizing soft switching in a wide voltage range for a three-port bidirectional DC converter. The three-port bidirectional DC converter includes a primary full-bridge circuit, a secondary full-bridge circuit, a secondary interleaved boost circuit, and a high-frequency transformer. The first winding of the high-frequency transformer is connected in series between two power switches of the primary full-bridge circuit, the second winding is connected in series between two power switches of the secondary full-bridge circuit, and the third winding is connected in series between two power switches of the secondary interleaved boost circuit. A clamping capacitor is connected in parallel on the port side of the secondary interleaved boost circuit. When the three-port bidirectional DC converter is used in the power supply system of an electric vehicle, the PFC port of the three-port bidirectional DC converter is connected to the AC grid through a pre-stage power factor correction circuit, and the high-voltage port and the low-voltage port are respectively connected to the power battery and the low-voltage battery.

[0009] This method realizes soft switching in a wide voltage range for electric vehicles in both charging and discharging modes by modulating the duty cycle D, the phase angle difference φ 12 and φ 13 . D represents the duty cycle of conduction of two power switches on the lower arm of the secondary interleaved boost circuit, and φ 12 represents the phase angle difference between the midpoint voltage of the arm of the primary full-bridge circuit and the midpoint voltage of the arm of the secondary full-bridge circuit, and φ 13 represents the phase angle difference between the midpoint voltage of the arm of the primary full-bridge circuit and the midpoint voltage of the arm of the secondary interleaved boost circuit.

[0010] In the charging mode, the power battery is connected to the AC grid through the on-board charger, and the AC grid transmits electric energy to the power battery. First, the voltage of the PFC port and the high-voltage port of the converter are collected, and the reference value V of the PFC port voltage is calculated 1ref , V 1ref = N 1 / N 2 ×V 2 , N 1 , N 2They are the number of turns of the first winding and the second winding of the variable-frequency transformer, respectively, V 2 represents the high-voltage port voltage of the converter; then, the difference between the grid voltage, grid current, and the reference value and actual value of the PFC port voltage is input into the PFC controller. The PFC controller controls the pulse modulation generator to generate trigger pulses, controls the on and off of each power switch in the front-stage power factor correction circuit, and controls the PFC port voltage;

[0011] For the modulation of the duty cycle D, first, calculate the reference value V clref of the clamping capacitor voltage, V clref = N 3 / N 2 ×V 2 , where N 3 is the number of turns of the third winding of the variable-frequency transformer; then, the difference between the reference value and actual value of the clamping capacitor voltage is input into the duty cycle PI controller, and the duty cycle PI controller outputs the duty cycle D; finally, the phase-shifted pulse width modulation generator generates trigger pulses to control the on and off of each power switch in the secondary full-bridge circuit and the secondary interleaved boost circuit;

[0012] For the modulation of the phase angle difference φ 12 , collect the high-voltage port current of the converter, input the difference between the actual value and reference value of the high-voltage port current into the first phase-shift angle PI controller, and the first phase-shift angle PI controller outputs the phase angle difference φ 12 , and the phase-shifted pulse width modulation generator generates trigger pulses to control the on and off of each power switch in the primary full-bridge circuit and the secondary full-bridge circuit;

[0013] For the modulation of the phase angle difference φ 13 , collect the low-voltage port voltage of the converter, input the difference between the actual value and reference value of the low-voltage port voltage into the second phase-shift angle PI controller, and the second phase-shift angle PI controller outputs the phase angle difference φ 13 , and the phase-shifted pulse width modulation generator generates trigger pulses to control the on and off of each power switch in the primary full-bridge circuit and the secondary interleaved boost circuit;

[0014] In the discharge mode, the power battery acts as a power source to transmit electrical energy to the on-vehicle charger or the low-voltage battery; the modulation of the phase angle difference φ 13 and the duty cycle D is the same as that in the charging mode; for the modulation of the phase angle difference φ 12 , first, collect the low-voltage port voltage of the converter and calculate the reference value V 1ref of the PFC port voltage, V 1ref = N 1 / N 2 ×V 2, the difference between the actual value and the reference value of the PFC port voltage is input into the first phase-shift angle PI controller, and the first phase-shift angle PI controller outputs the phase angle difference φ 12 , the phase-shift pulse width modulation generator generates trigger pulses to control the on and off of each power switch in the primary full-bridge circuit and the secondary full-bridge circuit.

[0015] On the other hand, the present invention provides a three-port bidirectional DC converter, including a primary full-bridge circuit, a secondary full-bridge circuit, a secondary interleaved boost circuit, a high-frequency transformer, capacitors C 1 ~C 2 , a clamping capacitor C cl and an inductor L 1 ~L 4 ; the primary full-bridge circuit includes power switches S 11 ~S 14 , power switches S 11 and S 13 are located on the upper bridge arm, and power switches S 12 and S 14 are located on the lower bridge arm; the secondary full-bridge circuit includes power switches S 21 ~S 24 , power switches S 21 and S 23 are located on the upper bridge arm, and power switches S 22 and S 24 are located on the lower bridge arm; the secondary interleaved boost circuit includes power switches S 31 ~S 34 , power switches S 31 and S 33 are located on the upper bridge arm, and power switches S 32 and S 34 are located on the lower bridge arm; capacitor C 1 is connected in parallel on the port side of the primary full-bridge circuit, capacitor C 2 is connected in parallel on the port side of the secondary full-bridge circuit, and the clamping capacitor C cl is connected in parallel on the port side of the secondary interleaved boost circuit; the first winding of the high-frequency transformer is connected in series between two groups of power switches of the primary full-bridge circuit, inductor L 1 and the second winding of the frequency conversion transformer are connected in series between two groups of power switches of the secondary full-bridge circuit, inductor L 2 and the third winding of the frequency conversion transformer are connected in series between two groups of power switches of the secondary interleaved boost circuit; one end of inductor L 3 is connected to the middle position of the corresponding bridge arm of one group of power switches of the secondary interleaved boost circuit, and one end of inductor L 4 is connected to the middle position of the corresponding bridge arm of the other group of power switches of the secondary interleaved boost circuit, inductor L 3 and L 4is connected to the other end and is simultaneously connected to the positive pole of the port voltage of the secondary-side interleaved boost circuit.

[0016] Compared with the prior art, the beneficial effects of the present invention are as follows:

[0017] 1. Based on the variable DC bus voltage (the PFC port voltage of the converter), through the combination of PWM modulation and phase-shift modulation, for various working modes of electric vehicles such as charging, discharging, and driving, through digital control, the three-port bidirectional DC converter realizes soft-switching operation in a wide voltage range, and can meet the efficient power transmission of the converter from light load to heavy load. Among them, the voltage change range of the high-voltage port of the converter is 250-450V, and the voltage change range of the low-voltage port is 10.5-15V.

[0018] 2. In the electric vehicle charging mode, the power battery is connected to the AC power grid through the on-vehicle charger, and the voltage of the power battery is changing in real time. At this time, the voltage V of the power battery collected 2 controls the PFC port voltage V of the transformer through the front-stage power factor correction circuit 1 to make V 1 = N 1 / N 2 ×V 2 ; at the same time, by collecting the clamped capacitor voltage V cl , and adjusting the duty cycle D through the digital controller, so that the clamped capacitor voltage V cl = N 3 / N 2 ×V 2 (the actual V cl is obtained through V cl = V 3 / (1-D) after the duty cycle D obtained by control); finally, the voltage matching of the three ports is realized, so that the converter realizes soft switching in a wide voltage range.

[0019] 3. In the electric vehicle discharging mode, the voltage of the power battery is also changing in real time. At this time, through the voltage V of the power battery collected 2 , the phase-shift angle φ 12 is controlled and adjusted through the digital controller (at this time, φ 12 simultaneously takes into account the two tasks of voltage matching and power flow), so that V 1 = N 1 / N 2 ×V 2 ; at the same time, by collecting the clamped capacitor voltage V cl , and adjusting the duty cycle D through the digital controller, so that the clamped capacitor voltage V cl = N 3 / N 2 ×V 2(The actual V cl is obtained by passing the duty cycle D obtained after control through V cl = V 3 / (1 - D)); Finally, the voltages of the three ports are matched, enabling the converter to achieve soft switching in a wide voltage range. BRIEF DESCRIPTION OF THE DRAWINGS

[0020] Figure 1 is the topology diagram of the control system used in the present invention;

[0021] Figure 2 is the control logic timing diagram of phase - shift modulation and PWM modulation;

[0022] FIG. 3(a) is the control block diagram in the electric vehicle charging mode;

[0023] FIG. 3(b) is the control block diagram in the electric vehicle discharging mode;

[0024] FIG. 4(a) is the simulation waveform diagram of the converter in the electric vehicle charging operation state when the power battery voltage is 350V;

[0025] FIG. 4(b) is the simulation waveform diagram of the converter in the electric vehicle discharging operation state;

[0026] FIG. 4(c) is the simulation waveform diagram of the converter in the electric vehicle driving state;

[0027] FIG. 4(d) is the simulation waveform diagram of the converter in the composite state of on - vehicle charging and low - voltage conversion of the electric vehicle;

[0028] FIG. 4(e) is the simulation waveform diagram of the converter in the electric vehicle charging operation state when the power battery voltage is 400V. DETAILED DESCRIPTION OF THE EMBODIMENTS

[0029] The technical solutions of the present invention will be described in detail below in conjunction with the drawings and specific embodiments, but the protection scope of this application is not limited thereby.

[0030] The present invention provides a method for realizing soft switching in a wide voltage range for a three - port bidirectional DC converter (abbreviated as the method, see Figure 1 ~4). When the three - port bidirectional DC converter is used in the power supply system of an electric vehicle, the PFC port of the three - port bidirectional DC converter is connected to the AC power grid through a pre - stage power factor correction circuit, and the high - voltage port and the low - voltage port of the three - port bidirectional DC converter are respectively connected to the power battery and the low - voltage battery;

[0031] Figure 1Topological diagram of the control system used in the present invention. The control system includes a three-port bidirectional DC converter and a controller for realizing soft switching over a wide voltage range. The three-port bidirectional DC converter includes a primary full-bridge circuit, a secondary full-bridge circuit, a secondary interleaved boost circuit, a high-frequency transformer, capacitors C 1 ~C 2 , clamping capacitor C cl and inductors L 1 ~L 4 ; The primary full-bridge circuit includes power switches S 11 ~S 14 , power switches S 11 and S 14 constitute the first group of power switches, and power switches S 12 and S 13 constitute the second group of power switches; The secondary full-bridge circuit includes power switches S 21 ~S 24 , power switches S 21 and S 24 constitute the third group of power switches, and power switches S 22 and S 23 constitute the fourth group of power switches. The secondary interleaved boost circuit includes power switches S 31 ~S 34 , power switches S 31 and S 34 constitute the fifth group of power switches, and power switches S 32 and S 33 constitute the sixth group of power switches; Capacitor C 1 is connected in parallel at both ends of the port side of the primary full-bridge circuit, capacitor C 2 is connected in parallel at both ends of the port side of the secondary full-bridge circuit, and clamping capacitor C cl is connected in parallel at both ends of the port side of the secondary interleaved boost circuit; The frequency conversion transformer is used to connect the primary full-bridge circuit, the secondary full-bridge circuit and the secondary interleaved boost circuit. The first winding of the frequency conversion transformer is connected in series between the first group of power switches and the second group of power switches. Inductor L 1 and the second winding of the frequency conversion transformer are connected in series between the third group of power switches and the fourth group of power switches. Inductor L 2 and the third winding of the frequency conversion transformer are connected in series between the fifth group of power switches and the sixth group of power switches. One end of inductor L 3 is connected to the middle position of the corresponding bridge arm of the sixth group of power switches, and one end of inductor L 4 is connected to the middle position of the corresponding bridge arm of the fifth group of power switches. The other ends of inductors L 3 and L 4 are connected to each other and are simultaneously connected to the positive pole of the port voltage of the secondary interleaved boost circuit.

[0032] The controller for achieving soft switching in a wide voltage range includes a pulse width modulation generator, a PFC controller, a digital controller, a phase-shifted pulse width modulation generator, a primary side voltage collector, a secondary side first voltage collector, a secondary side first current collector, a secondary side second voltage collector, and a clamped capacitor voltage collector; the primary side voltage collector is used to collect the bus voltage of the primary side full-bridge circuit, that is, the PFC port voltage of the converter; the secondary side first voltage collector is used to collect the bus voltage of the secondary side full-bridge circuit, that is, the high-voltage port voltage of the converter, which is also the charging voltage of the power battery; the secondary side first current collector is used to collect the bus current of the secondary side full-bridge circuit, that is, the high-voltage port current of the converter; the secondary side second voltage collector is used to collect the port voltage of the secondary side interleaved boost circuit, that is, the low-voltage port voltage of the converter; the clamped capacitor voltage collector is used to collect the clamped capacitor voltage; each collector is connected to the digital controller; the digital controller includes a duty cycle PI controller, a first phase-shift angle PI controller, and a second phase-shift angle PI controller. The PFC controller controls the pulse modulation generator to generate trigger pulses, controls the on / off of the four power switches in the pre-stage power factor correction circuit, and modulates the PFC port voltage of the converter. The digital controller respectively performs trigger control on the power switches S 11 ~S 14 in the primary side full-bridge circuit, the power switches S 21 ~S 24 in the secondary side full-bridge circuit, and the power switches S 31 ~S 34 in the secondary side interleaved boost circuit, so that there is a phase angle difference φ h1 between the midpoint voltage v h2 of the primary side full-bridge circuit arm and the midpoint voltage v 12 of the secondary side full-bridge circuit arm, and there is a phase angle difference φ h1 between the midpoint voltage v h3 of the primary side full-bridge circuit arm and the midpoint voltage v 13 of the secondary side interleaved boost circuit arm; the conduction duty cycle of the power switches S 11 ~S 14 in the primary side full-bridge circuit and the power switches S 21 ~S 24 in the secondary side full-bridge circuit is 50%, and the conduction duty cycle of the two power switches S 31 and S 33 on the upper bridge arm of the fifth group of power switches and the sixth group of power switches is 1-D, and the conduction duty cycle of the two power switches S 32 and S 34 on the lower bridge arm is D.

[0033] Figure 2 is the control logic timing diagram of phase-shift modulation and PWM modulation. The timing of the power switches S 11 and S 14 is consistent. The power switches S12 and S 13 are in the same timing sequence. The power switch S 21 and S 24 are in the same timing sequence. The power switch S 22 and S 23 are in the same timing sequence; the power switch S 31 and S 34 are in the same timing sequence. The power switch S 32 and S 33 are in the same timing sequence. The controller controls the magnitude and direction of the power transmitted between the primary full-bridge circuit, the secondary full-bridge circuit, and the secondary interleaved boost circuit by adjusting the magnitude and polarity of the phase angle differences φ 12 and φ 13 . As shown in Figure 2 , when the phase angle difference φ 12 increases, the midpoint voltage of the bridge arm of the primary full-bridge circuit shifts to the right relative to the midpoint voltage of the bridge arm of the secondary full-bridge circuit; when the phase angle difference φ 13 decreases, the midpoint voltage of the bridge arm of the secondary interleaved boost circuit shifts to the left relative to the midpoint voltage of the bridge arm of the secondary full-bridge circuit. In actual situations, to prevent the upper and lower bridge arms from conducting simultaneously due to switching speed issues, the power switch needs to consider the dead time. Therefore, the actual duty cycle is less than the theoretical value. The smaller the dead time, the better the output waveform, but the reliability will decrease. For the convenience of description in the present invention, the on and off of the power switch includes the dead time, and the actual duty cycle should be less than the described duty cycle.

[0034] The modulation process for achieving soft switching in a wide voltage range will be described separately according to the two working modes of electric vehicle charging and discharging below; the PFC port voltage of the converter is denoted as V 1 , the high-voltage port voltage and current of the converter are respectively denoted as V 2 and I 2 , the low-voltage port voltage of the converter is denoted as V 3 , and the clamping capacitor voltage is denoted as V cl ;

[0035] Figure 3(a) is the control block diagram of the electric vehicle charging mode. In the charging mode, the power battery is connected to the AC grid through the on-vehicle charger, and the AC grid transmits electrical energy to the power battery; the front-stage power factor correction circuit is cascaded with the primary full-bridge circuit of the three-port bidirectional DC converter. The front-stage power factor correction circuit consists of four power switches T 1 ~T 4 . First, the PFC port voltage V 1 and the high-voltage port voltage V 2 of the converter are collected, and the PFC port voltage reference value V 1ref is calculated according to the turns ratio of the first winding and the second winding of the frequency conversion transformer. V 1ref = N 1 / N2 ×V 2 ,N 1 、N 2 are the number of turns of the first winding and the second winding of the variable-frequency transformer respectively; then, the grid voltage V AC , the grid current I AC and the difference between the reference value and the actual value of the PFC port voltage are input into the PFC controller. The PFC controller controls the pulse modulation generator to generate trigger pulses and controls the on / off of the four power switches T 1 ~T 4 to control the PFC port voltage V 1 .

[0036] For the modulation of the duty cycle D, first, calculate the reference value V clref of the clamping capacitor voltage according to the turns ratio of the third winding and the second winding of the variable-frequency transformer, V clref =N 3 / N 2 ×V 2 , N 3 is the number of turns of the third winding of the variable-frequency transformer; then, input the difference between the reference value and the actual value of the clamping capacitor voltage into the duty cycle PI controller, and the duty cycle PI controller outputs the duty cycle D; finally, the phase-shifted pulse width modulation generator generates trigger pulses according to the control logic timing to control the on / off of each power switch in the secondary interleaved boost circuit. By modulating the duty cycle D, control the clamping capacitor voltage V cl to achieve port voltage matching and enable the converter to achieve soft switching in a wide voltage range. The duty cycle PI controller adopts the zero-pole compensation method, and its transfer function is s represents the frequency domain operator.

[0037] Since the charging mode should maintain constant current charging, for the modulation of the phase angle difference φ 12 , collect the high-voltage port current I 2 of the converter, input the difference between the actual value and the reference value (given by the power battery) of the high-voltage port current into the first phase-shifted angle PI controller, and the first phase-shifted angle PI controller outputs the phase angle difference φ 12 . The phase-shifted pulse width modulation generator generates trigger pulses according to the control logic timing to control the on / off of each power switch in the primary full-bridge circuit and the secondary full-bridge circuit, and modulate the high-voltage port current I 2 of the converter, that is, control the charging current of the power battery. Since there are high ripple components in the high-voltage port current of the converter, a first-order filter with a cut-off frequency of 1 kHz is used for smoothing. The K p of the first phase-shifted angle PI controller is 0.037, and K i is 485.3.

[0038] For the modulation of the phase angle difference φ 13 collect the low-voltage port voltage V of the converter 3 , input the difference between the actual value and the reference value (given by the low-voltage load) of the low-voltage port voltage into the second phase-shifting angle PI controller, and the second phase-shifting angle PI controller outputs the phase angle difference φ 13 . The phase-shifted pulse width modulation generator generates trigger pulses according to the control logic timing to control the on and off of each power switch in the primary full-bridge circuit and the secondary interleaved boost circuit. Through the modulation of the phase angle difference φ 13 , control the low-voltage port voltage V of the converter 3 . The K p of the second phase-shifting angle PI controller is 0.008, and K i is 38.08.

[0039] Figure 3(b) is the control block diagram of the electric vehicle discharging mode. In the discharging mode, the power battery connected to the high-voltage port of the converter serves as the power source to transmit electrical energy to the on-board charger or the low-voltage battery. For the modulation of the phase angle difference φ 13 and the duty cycle D, it is the same as the charging mode, that is, control the clamping capacitor voltage V cl by adjusting the duty cycle D, and control the low-voltage port voltage V 13 by adjusting the phase angle difference φ 3 .

[0040] For the modulation of the phase angle difference φ 12 , first, collect the low-voltage port voltage V of the converter 2 , calculate the PFC port voltage reference value V 1ref according to the turns ratio of the first winding and the second winding of the frequency conversion transformer, V 1ref = N 1 / N 2 × V 2 . Input the difference between the actual value and the reference value of the PFC port voltage into the first phase-shifting angle PI controller, and the first phase-shifting angle PI controller outputs the phase angle difference φ 12 . The phase-shifted pulse width modulation generator generates trigger pulses according to the control logic timing to control the on and off of each power switch in the primary full-bridge circuit and the secondary full-bridge circuit. Through the modulation of the phase angle difference φ 12 , control the high-voltage port voltage V of the converter 1 . At this time, the K p of the first phase-shifting angle PI controller is -0.0045, and K i is -1.34.

[0041] Embodiment

[0042] The control method of the present invention will be described below in conjunction with embodiments. Set the turns ratio N of the first winding and the second winding of the transformer 12 to be 1.5, and the turns ratio N of the first winding and the third winding 13 to be 18, and the turns ratio N of the second winding and the third winding 23 to be 12.

[0043] Among them,

[0044] The circuit parameters of the three-port bidirectional DC converter are: L 1 = 5.9 μH, L 2 = 0.09 μH, L 3 = 2.2 μH, L 4 = 2.2 μH, C cl = 400 μF.

[0045] Assume that the power battery voltage V 2 is 350 V, and the low-voltage load voltage V 3 is 13.8 V. Then the PFC port voltage V of the converter 1 = N 12 × V 2 = 525 V, and the duty cycle D = 1 - N 23 × V 3 / V 2 = 0.5269, V cl = V 3 / (1 - D) = 29.17 V.

[0046] According to the above parameters, the converter is simulated in two modes of electric vehicle charging and discharging respectively, and the waveform diagrams shown in FIGS. 4(a)-(e) are obtained;

[0047] FIG. 4(a) is the simulation waveform diagram of the converter in the charging operation state of the electric vehicle by the AC power grid; at this time, the input power P of the primary full-bridge circuit 1 is 3315.4 W, the output power P of the secondary full-bridge circuit 2 is 3300 W, and the transmission power P of the secondary interleaved boost circuit 3 is 0. Electric energy is charged to the power battery on the secondary full-bridge circuit port side through the primary full-bridge circuit port side, where the phase angles of the power switches S 21 -S 24 , S 31 -S 34 all lag behind S 11 -S 14 ; φ 12 is 12.3°, φ 13 is 0.002°, i 1 is the current on the transformer side of the primary full-bridge circuit, i2 is the current of inductor L 1 i 3 is the current of inductor L 2 i 4 is the current of inductor L 3 i 5 is the current of inductor L 4 is the current; v h1 is the square wave voltage on the transformer side of the primary full-bridge circuit, v h2 is the square wave voltage on the transformer side of the secondary full-bridge circuit, v h3 is the square wave voltage on the transformer side of the secondary interleaved boost circuit. It can be seen from the simulation waveform diagram that all power switches have achieved soft switching.

[0048] Figure 4(b) is the simulation waveform diagram of the converter under the operating state of the electric vehicle discharging to the AC grid; at this time, the output power P of the primary full-bridge circuit 1 is 3284.76W, the input power P of the secondary full-bridge circuit 2 is 3300W, the transmission power P of the secondary interleaved boost circuit 3 is 0, and the power battery feeds back electric energy to the AC grid through the power factor correction circuit from the port side of the secondary full-bridge circuit to the port side of the primary full-bridge circuit, where S 21 -S 24 's phase angle leads S 11 -S 14 ; S 31 -S 34 's phase angle lags behind S 11 -S 14 ; φ 12 is -12.27°, φ 13 is 0.069°, i 1 is the current on the transformer side of the primary full-bridge circuit, i 2 is the current of inductor L 1 i 3 is the current of inductor L 2 i 4 is the current of inductor L 3 i 5 is the current of inductor L 4 is the current; v h1 is the square wave voltage on the transformer side of the primary full-bridge circuit, v h2 is the square wave voltage on the transformer side of the secondary full-bridge circuit, v h3 is the square wave voltage on the transformer side of the secondary interleaved boost circuit. It can be seen from the simulation waveform diagram that all power switches have achieved soft switching.

[0049] Figure 4(c) is the simulation waveform diagram of the converter under the driving state (discharge mode) of the electric vehicle. At this time, the input power P of the secondary full-bridge circuit 2 is 1000 W, and the output power P of the secondary interleaved boost circuit 3 is 983.3 W. The transmission power P of the primary full-bridge circuit 1 is 0. The electric energy of the power battery discharges to the low-voltage load on the port side of the secondary interleaved boost circuit through the port side of the secondary full-bridge circuit. Among them, S 21 -S 24 's phase angle leads S 11 -S 14 ; S 31 -S 34 's phase angle lags behind S 11 -S 14 ; φ 12 is -3.54°, φ 13 is 8.237°, i 1 is the current on the transformer side of the primary full-bridge circuit, i 2 is the current of the inductor L 2 , i 3 is the current of the inductor L 3 , i 4 is the current of the inductor L 4 , i 5 is the current of the inductor L 5 ; v h1 is the square-wave voltage on the transformer side of the primary full-bridge circuit, v h2 is the square-wave voltage on the transformer side of the secondary full-bridge circuit, v h3 is the square-wave voltage on the transformer side of the secondary interleaved boost circuit. It can be seen from the simulation waveform diagram that all power switches have achieved soft switching.

[0050] Figure 4(d) is the simulation waveform diagram of the converter under the vehicle-mounted charging and low-voltage conversion composite mode (charging mode). At this time, the input power P of the primary full-bridge circuit 1 is 3374.65 W, the output power P of the secondary full-bridge circuit 2 is 3000 W, and the output power P of the secondary interleaved boost circuit 3 is 300 W. Electric energy is supplied to the power battery on the port side of the secondary full-bridge circuit and the low-voltage load on the port side of the secondary interleaved boost circuit simultaneously through the port side of the primary full-bridge circuit. Among them, S 21 -S 24 , S 31 -S 34 's phase angles both lag behind S 11 -S 14 ; φ 12 is 11.98°, φ 13is 2.533°, i 1 is the current on the transformer side of the primary full-bridge circuit, i 2 is the inductor L 2 's current, i 3 is the inductor L 3 's current, i 4 is the inductor L 4 's current, i 5 is the inductor L 5 's current; v h1 is the square-wave voltage on the transformer side of the primary full-bridge circuit, v h2 is the square-wave voltage on the transformer side of the secondary full-bridge circuit, v h3 is the square-wave voltage on the transformer side of the secondary interleaved boost circuit. It can be seen from the simulation waveform diagram that all power switches have achieved soft switching.

[0051] Assume the power battery voltage V 2 is 400V, the low-voltage load voltage V 3 is 13.8V, then the PFC port voltage V of the transformer 1 = N 12 ×V 2 = 600V, the duty cycle D = 1 - N 13 ×V 3 / V 2 = 0.586, V cl = V 3 / (1 - D) = 33.33V. The simulation waveform diagram of the converter in the charging operation state shown in Figure 4(e). At this time, the input power P of the primary full-bridge circuit 1 is 3487.09W, the output power P of the secondary full-bridge circuit 2 is 3300W, the transmission power P of the secondary interleaved boost circuit 3 is 0, and the electric energy is charged to the power battery on the port side of the secondary full-bridge circuit through the port side of the primary full-bridge circuit, where S 21 -S 24 , S 31 -S 34 's phase angles all lag behind S 11 -S 14 ; φ 12 is 9.36°, φ 13 is 0.169°, i 1 is the current on the transformer side of the primary full-bridge circuit, i 2 is the inductor L 2 's current, i 3 is the inductor L 3 's current, i 4 is the inductor L 4 's current, i 5 is the inductor L5 The current; v h1 is the square-wave voltage on the transformer side of the primary full-bridge circuit, v h2 is the square-wave voltage on the transformer side of the secondary full-bridge circuit, v h3 is the square-wave voltage on the transformer side of the secondary interleaved boost circuit. It can be seen from the simulation waveform diagram that all power switches have achieved soft switching.

[0052] Matters not described in the present invention are applicable to the prior art.

Claims

1. A method for realizing soft switching in a wide voltage range of a three-port bidirectional DC converter. The three-port bidirectional DC converter includes a primary full-bridge circuit, a secondary full-bridge circuit, a secondary interleaved boost circuit, and a high-frequency transformer. The first winding of the high-frequency transformer is connected in series between two sets of power switches of the primary full-bridge circuit, the second winding is connected in series between two sets of power switches of the secondary full-bridge circuit, and the third winding is connected in series between two sets of power switches of the secondary interleaved boost circuit. A clamping capacitor is connected in parallel on the port side of the secondary interleaved boost circuit. When the three-port bidirectional DC converter is used in the power supply system of an electric vehicle, the PFC port of the three-port bidirectional DC converter is connected to the AC grid through a pre-stage power factor correction circuit, and the high-voltage port and the low-voltage port are respectively connected to the power battery and the low-voltage battery. Characterized in that, This method modulates the duty cycle D, phase angle difference φ 12 and φ 13 to achieve soft switching in a wide voltage range for electric vehicles in both charging and discharging modes; D represents the duty cycle of the conduction of two power switches on the lower arm of the secondary interleaved boost circuit, and φ 12 represents the phase angle difference between the midpoint voltage of the primary full-bridge circuit arm and the midpoint voltage of the secondary full-bridge circuit arm, and φ 13 represents the phase angle difference between the midpoint voltage of the primary full-bridge circuit arm and the midpoint voltage of the secondary interleaved boost circuit arm; In the charging mode, the power battery is connected to the AC grid through the on-vehicle charger, and the AC grid transmits electric energy to the power battery. First, the PFC port voltage and the high-voltage port voltage of the acquisition converter are collected, and the PFC port voltage reference value V 1ref , V 1ref = N 1 / N 2 ×V 2 , N 1 , N 2 are the number of turns of the first winding and the second winding of the frequency conversion transformer respectively, and V 2 represents the high-voltage port voltage of the converter. Then, the grid voltage, grid current, and the difference between the PFC port voltage reference value and the actual value are input into the PFC controller. The PFC controller controls the pulse modulation generator to generate trigger pulses, controls the on / off of each power switch in the front-stage power factor correction circuit, and controls the PFC port voltage; For the modulation of the duty cycle D, first, calculate the reference value V of the clamped capacitor voltage clref , V clref = N 3 / N 2 × V 2 , N 3 is the number of turns of the third winding of the frequency conversion transformer; then, input the difference between the reference value and the actual value of the clamped capacitor voltage into the duty cycle PI controller, and the duty cycle PI controller outputs the duty cycle D; finally, the phase-shifted pulse width modulation generator generates trigger pulses to control the on and off of each power switch in the secondary interleaved boost circuit; For the modulation of the phase angle difference φ 12 During modulation, the current of the high-voltage port of the acquisition converter is collected, and the difference between the actual value and the reference value of the high-voltage port current is input into the first phase-shift angle PI controller. The first phase-shift angle PI controller outputs the phase angle difference φ 12 Then, the phase-shift pulse width modulation generator generates trigger pulses to control the on / off of each power switch in the primary full-bridge circuit and the secondary full-bridge circuit; For the phase angle difference φ 13 modulation, the low-voltage port voltage of the acquisition converter is collected, and the difference between the actual value and the reference value of the low-voltage port voltage is input into the second phase-shifting angle PI controller, and the second phase-shifting angle PI controller outputs the phase angle difference φ 13 , the phase-shifted pulse width modulation generator generates trigger pulses to control the on and off of each power switch in the primary full-bridge circuit and the secondary interleaved boost circuit; In the discharge mode, the power battery acts as a power source and transmits electrical energy to the on-vehicle charger or the low-voltage battery; for the modulation of the phase angle difference φ 13 and the duty cycle D, it is the same as that in the charging mode; for the modulation of the phase angle difference φ 12 , first, collect the voltage at the low-voltage port of the converter and calculate the reference value V 1ref of the PFC port voltage, V 1ref = N 1 / N 2 × V 2 . Input the difference between the actual value and the reference value of the PFC port voltage into the first phase-shift angle PI controller. The first phase-shift angle PI controller outputs the phase angle difference φ 12 . The phase-shifted pulse-width modulation generator generates trigger pulses to control the on-off of each power switch in the primary full-bridge circuit and the secondary full-bridge circuit.

2. The method for realizing soft switching in a wide voltage range of a three-port bidirectional DC converter according to claim 1, Characterized in that, During the control process, the duty cycle of all power switches in the primary full-bridge circuit and all power switches in the secondary full-bridge circuit is 50%, and the duty cycle of two power switches in the upper arm of the secondary interleaved boost circuit is 1 - D.

3. The method for realizing soft switching in a wide voltage range of a three-port bidirectional DC converter according to claim 1, Characterized in that, The duty cycle PI controller adopts a zero-pole compensation method, and its transfer function is: In the formula, s represents the frequency domain operator.

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