Communication and ranging integrated target displacement measurement method based on heterodyne interference system

Abstract

The present invention discloses a communication and ranging integrated target displacement measurement method based on a heterodyne interference system, which mainly solves the problem of low ranging accuracy in the prior art. The solution is as follows: A communication and ranging integrated light beam is generated by a frequency-stabilized laser, and is divided into a reference-end light beam and a measurement-end light beam by a beam splitter prism; the reference-end light beam undergoes interference through a reference polarizer and is converted into a reference-end electrical signal by a photoelectric converter; the measurement-end light beam is divided into two light beams with different frequencies by a polarization beam splitter prism, and then is reflected by a reference corner reflector and a measurement corner reflector respectively onto a measurement polarizer to undergo interference, and is converted into a measurement-end electrical signal by a photoelectric converter; the electrical signals at the reference end and the measurement end are subjected to phase resolution to obtain two sine and cosine signals; the integer and fractional parts of the phase are calculated using these two signals, and then the relative displacement of the object is calculated through the phase. The present invention greatly improves the ranging accuracy and can be used for satellite communication and ranging integration.

Description

Technical Field

[0001] The present invention belongs to the field of communication technologies, and particularly relates to a heterodyne interference displacement measurement method, which can be used for satellite communication and ranging integration. Background Art

[0002] Modern ultra-precision instrument and equipment processing, ultra-large scale integrated circuit processing, and nano-scale measurement technologies and other international cutting-edge technologies require the strong support and fine calibration of ultra-precision measurement technologies. As an important part of ultra-precision technologies, heterodyne laser interference measurement systems have been widely applied to various ultra-precision measurement and processing fields due to their high measurement resolution and accuracy, non-contact measurement, strong anti-interference ability, and good traceability and reproducibility of measurement values.

[0003] "Research on Phase Subdivision Technology of Dual-Frequency Laser Interferometer with Picometer Resolution" proposed a dual-frequency quadrature phase-locked amplification phase measurement technology based on a heterodyne interference measurement system to achieve high-precision displacement measurement. Without considering optical errors, this method can achieve a measurement accuracy of the picometer level. However, this method can only be used to achieve a single ranging situation and cannot be used to achieve communication and ranging integration.

[0004] However, with the continuous improvement of spaceborne laser systems, relying on the carried laser communication and laser rangefinders, technical means for providing communication and relative positioning between spacecraft can be provided. Limited by the increasingly high constraints of satellite platforms on weight, volume, and power consumption, on the premise of meeting the requirements of project task indicators, the integrated design of communication and ranging is the future trend.

[0005] Richard S. Orr proposed "Combined GMSK Modulation and Pseudo-Random Code Ranging for Communication and Ranging" in 2018. In this design, the same signal light is shared for ranging and communication, and the pseudo-random code ranging method is used to achieve laser ranging. Pseudo-random code ranging is divided into two main parts: the master station and the slave station. First, at the master station, the code generator generates the clock code and the pseudo-random code, which are transmitted to the slave station after carrier modulation. Second, the code tracking loop of the slave station demodulates and calculates to recover the clock code and the pseudo-random code, thereby generating a downlink composite code at the slave station, which is transmitted back to the master station after modulation. Third, the code tracking loop of the master station analyzes the clock code and the pseudo-random sub-code, and obtains the rough measurement and fine measurement results by respectively comparing the phases of the clock code and the pseudo-random code. Finally, after a series of signal and data processing, the ranging result is obtained. Since this ranging method measures the absolute distance, its ranging accuracy can only reach the centimeter level and cannot achieve high-precision ranging. Summary of the Invention

[0006] The object of the present invention is to propose a communication and ranging integrated target displacement measurement method based on a heterodyne interference system in view of the deficiencies of the above-mentioned existing technologies, so as to realize the measurement of relative displacement, improve the displacement measurement accuracy, and make it reach the nanometer level.

[0007] To achieve the above object, the technical solution of the present invention is as follows:

[0008] 1. A communication and ranging integrated target displacement measurement method based on a heterodyne interference system, the system includes a frequency-stabilized laser, a beam splitter prism BS, a polarization beam splitter prism PBS, a reference corner reflector P r , a measurement corner reflector P m , a reference polarizer PA1, a measurement polarizer PA2, a photoelectric conversion device PT and a field programmable gate array FPGA, characterized in that the implementation steps are as follows:

[0009] (1) Generate two orthogonally polarized light beams with frequencies f1 and f2 respectively through a frequency-stabilized laser, and use the light beam with frequency f2 as a carrier to modulate the baseband data d(t) through binary phase shift keying BPSK to obtain the modulated light beam E2(t), and then synthesize it with the light beam with frequency f1 into a light beam E and emit it;

[0010] (2) Divide the emitted light beam E into a measurement end and a reference end through a common non-polarized state power beam splitter prism BS;

[0011] (3) Interfere the reference end light beam directly through the reference polarizer PA1 at an angle of 45 with the polarized light, and then obtain the reference end electrical signal f r ;

[0012] (4) Obtain the measurement end electrical signal f m :

[0013] (4a) Direct the measurement end light beam onto the polarization beam splitter prism PBS to divide the light beam E into orthogonally polarized light beams with frequencies f1 and f2 respectively;

[0014] (4b) Transmit the light beam with frequency f1 to the fixed reference corner reflector P r , and then reflect it back to the polarization beam splitter prism PBS; transmit the light beam with frequency f2 to the movable measurement corner reflector P m , obtain a light beam with a frequency changed to f2±Δf and then reflect it back to the polarization beam splitter prism PBS, where; is the frequency difference caused by the influence of the Doppler frequency shift, v is the moving speed of the measurement corner reflector P m , and λ is the wavelength of the laser emitted by the frequency-stabilized laser;

[0015] (4c) The two beams returning to the polarization beam splitter PBS are interfered by the measuring polarizer PA2, and then converted by the photoelectric converter PT to obtain the measuring end electrical signal f m ;

[0016] (6) Two orthogonal signals sin and cos are generated inside the FPGA through programming and are respectively compared with the reference end electrical signal f r and the measuring end electrical signal f m Perform mixing to obtain four mixed signals sinA1, cosA1, sinB1 and cosB1;

[0017] (7) Mix the mixed signals sinA1 and cosA1, sinB1 and cosB1 respectively, and perform low-pass digital filtering on the mixing results inside the FPGA to obtain two sinusoidal signals sinA3 and sinB3;

[0018] (8) The four mixed signals sinA1 and cosA1, sinB1 and cosB1 are squared and then subtracted, and then low-pass digitally filtered inside the FPGA to obtain two cosine signals cosA3 and cosB3;

[0019] (9) Phase calculation is performed on the two sine signals sinA3 and sinB3 and the two cosine signals cosA3 and cosB3 in the FPGA to obtain the two sine and cosine signals sinC and cosC;

[0020] (10) The relative distance between the targets to be measured is calculated based on the two sine and cosine signals sinC and cosC.

[0021] (10a) The two sine and cosine signals sinC and cosC are processed by a four-fine resolution phase processing method to calculate the integer part of the phase;

[0022] (10b) Divide the two sine and cosine signals to obtain the signal tan, and use the cordic algorithm to obtain the fractional part of the phase;

[0023] (10c) Add the fractional part of the phase to the integer part of the phase to get the overall phase value By overall phase value Calculate the relative distance s.

[0024] Compared with the prior art, the present invention has the following advantages:

[0025] First, the present invention uses a heterodyne interferometry system for integrated communication and ranging. Compared with the traditional method of using pseudo-random code ranging to achieve integrated communication and ranging, it not only greatly improves the ranging accuracy, that is, the ranging accuracy is increased from cm level to nm level, but also can realize the measurement of relative displacement;

[0026] Second, since a new operation of performing difference after mixing and squaring once is added in the phase resolution part of the present invention, the problem that the binary phase shift keying (BPSK) modulation method is prone to cause discontinuous phase of the ranging signal and unable to perform phase resolution is solved. The heterodyne interference measurement system can be used in the case of communication ranging integration without being affected by the communication end, and high-precision relative displacement measurement can be achieved. Description of the Drawings

[0027] Figure 1 is the schematic diagram of the existing heterodyne interference system;

[0028] Figure 2 is the schematic diagram of the implementation of the present invention;

[0029] Figure 3 is the ranging accuracy diagram of implementing communication ranging integration by the existing method;

[0030] Figure 4 is the ranging accuracy diagram of implementing communication ranging integration by the method of the present invention. Detailed Embodiment

[0031] The following further describes the embodiments and effects of the present invention in detail with reference to the drawings:

[0032] The specific implementation of this example is based on the heterodyne interference system to measure the displacement of the communication ranging integration target.

[0033] Referring to Figure 1 , the heterodyne interference system used in this example includes a frequency-stabilized laser LASER, a beam splitter prism BS, a reference polarizer PA1, a polarization beam splitter prism PBS, a reference corner reflector P r , a measurement corner reflector P m , a measurement polarizer PA2, a photoelectric conversion device PT, and a field programmable gate array FPGA, where:

[0034] The frequency-stabilized laser LASER is used to generate a polarized light beam E containing two different frequencies and orthogonal polarization directions.

[0035] The beam splitter prism BS is located on the right side of the frequency-stabilized laser LASER and is used to split the light beam emitted by the frequency-stabilized laser LASER into two light beams with perpendicular propagation directions, namely the reference end light beam E r and the measurement end light beam E m ;

[0036] The reference polarizer PA1 is located directly below the beam splitter prism BS and is used to change the polarization directions of the two polarization lights with orthogonal polarization directions in the reference end light beam E r so that they interfere;

[0037] The polarization beam splitter prism PBS is located on the right side of the beam splitter prism BS and is used to split the measurement-end beam E m into two polarized beams E1(t) and E2(t) with perpendicular propagation directions and perpendicular polarization directions;

[0038] The reference corner reflector P r is located directly above the polarization beam splitter prism PBS and is used to reflect the polarized beam E1(t) back to the polarization beam splitter prism PBS;

[0039] The measurement corner reflector P m is located on the right side of the polarization beam splitter prism PBS and maintains a horizontal movement state from left to right, and is used to reflect the polarized beam E2(t) back to the polarization beam splitter prism PBS;

[0040] The measurement polarizer PA2 is located on the right side of the reference polarizer PA1 and is used to change the polarization directions of the two beams reflected back to the polarization beam splitter prism PBS, so that they interfere;

[0041] The optoelectronic conversion device PT is located below the reference polarizer PA1 and the measurement polarizer PA2 and is used to convert the two beams after interference at the reference end and the measurement end into electrical signals;

[0042] The field programmable gate array FPGA is connected to the output signal of the optoelectronic conversion device PT and is used to perform operations on the electrical signals after optoelectronic conversion inside the FPGA to calculate the relative displacement of the measurement corner reflector P m ;

[0043] Refer to Figure 2 , the implementation steps of heterodyne interference displacement measurement based on the above system in this example are as follows:

[0044] Step 1, generate the composite beam E.

[0045] 1.1) Generate two orthogonally polarized light beams E1(t) and E2(t) with frequencies f1 and f2 respectively through the frequency-stabilized laser LASER:

[0046] E1(t) = cos(w1t)

[0047] E2(t) = cos(w2t)

[0048] where w1 and w2 are the angular frequencies of the polarized light beams E1(t) and E2(t) respectively;

[0049] 1.2) Use the light beam with frequency f2 as the carrier to modulate the baseband data d(t) through binary phase shift keying BPSK to obtain the modulated light beam E2′(t):

[0050]

[0051] 1.3) For the convenience of subsequent description, rewrite E2′(t) as:

[0052]

[0053] where D(t) represents D r (t) and D m (t) respectively according to the asynchrony of the baseband data d(t) between the transmitter and the receiver in actual communication, that is, the baseband data of the transmitter is represented as D r (t), and the baseband data of the receiver is represented as D m (t);

[0054] 1.4) Synthesize the rewritten light beam E2′(t) and the light beam E1(t) with a frequency of f1 into a light beam E.

[0055] Step 2, obtain the reference - end electrical signal f r .

[0056] 2.1) Divide the path of the synthesized light beam E transmitted through the beam - splitting prism BS into a measurement end and a reference end, that is, regard the light beam E r continuing to transmit forward through the beam - splitting prism BS as the reference end, and regard the light beam E m transmitted vertically downward after being reflected by the beam - splitting prism BS as the measurement end;

[0057] 2.2) Interfere the reference - end synthesized light beam E r directly through the reference polarizer PA1 to obtain a reference - end interference light beam: E R (t) = D r (t)cos(w1t - w2t), where w1 and w2 are the angular frequencies of the polarized light beams E1(t) and E2(t) respectively, and D r (t) represents the baseband data of the transmitter:

[0058] 2.3) Convert the obtained interference light beam E R (t) into a reference - end electrical signal f r through the optoelectronic conversion device PT again. The reference - end electrical signal f r is expressed as:

[0059] f r = D r (t)cos(w r t)

[0060] where w r = w1 - w2 is the angular frequency of the reference - end electrical signal f r .

[0061] Step 3, obtaining the electrical signal f at the measurement end m .

[0062] 3.1) Direct the beam E at the measurement end m directly onto the polarization beam splitter prism PBS, obtaining orthogonal beams E1(t) and E2′(t) with frequencies f1 and f2 respectively;

[0063] 3.2) Transmit the beam E1(t) with frequency f1 to the fixed reference corner reflector P r , and then reflect it back to the polarization beam splitter prism PBS;

[0064] 3.3) Transmit the beam E2′(t) with frequency f2 to the movable measurement corner reflector P m , obtaining a beam E3(t) with a frequency changed to f2±Δf:

[0065] E3(t) = D m (t)cos((w2 + Δw)t)

[0066] where, is the frequency difference caused by the Doppler frequency shift, v is the moving speed of the measurement corner reflector P m , λ is the wavelength of the laser emitted by the frequency-stabilized laser, w2±Δw is the angular frequency of the beam E3(t), and D m (t) represents the baseband data at the receiving end:

[0067] 3.3) Reflect the beam E3(t) with frequency changed by the measurement corner reflector P m back to the polarization beam splitter prism PBS, and interfere it with the beam E1(t) returned to the polarization beam splitter prism PBS in step 3.2) through the measurement polarizer PA2, obtaining a measurement-end interference beam: E M (t) = D m (t)cos(w1t - (w2±Δw)t);

[0068] 3.4) Convert the measurement-end interference beam E M (t) through the optoelectronic conversion device PT to obtain the electrical signal f at the measurement end m :

[0069] f m = D m (t)cos(w m t)

[0070] where, w m = w1 - (w2±Δw) is the angular frequency of the electrical signal f at the measurement end m .

[0071] Step 4, obtain two sinusoidal signals sinA3 and sinB3.

[0072] 4.1) Generate two orthogonal signals sin = sin(w3t) and cos = cos(w3t) inside the FPGA through programming, and mix them with the electrical signals f r and f m at the measurement end and the reference end respectively to obtain four mixed-frequency signals sinA1, cosA1, sinB1, and cosB1, which are respectively expressed as follows:

[0073]

[0074] where w3 is the angular frequency of the two orthogonal signals sin and cos generated inside the FPGA;

[0075] 4.2) Mix the mixed-frequency signals sinA1 and cosA1, sinB1 and cosB1 respectively again to obtain two sinusoidal mixed-frequency signals sinA2 and sinB2:

[0076]

[0077] 4.3) Perform low-pass digital filtering on the sinusoidal mixed-frequency signals sinA2 and sinB2 inside the FPGA to obtain two sinusoidal signals sinA3 and sinB3, which are respectively expressed as follows:

[0078]

[0079] Step 5, obtain two cosine signals cosA3 and cosB3.

[0080] 5.1) Square the above four mixed-frequency signals sinA1, cosA1, sinB1, and cosB1 respectively to obtain four squared signals sinA′1, cosA′1, sinB′1, and cosB′1, which are respectively expressed as follows:

[0081]

[0082] 5.2) Subtract the four squared signals sinA1′ and cosA1′, sinB1′ and cosB1′ respectively to obtain two cosine mixed-frequency signals cosA2 and cosB2, which are respectively expressed as follows:

[0083]

[0084] 5.3) Perform low-pass digital filtering on the cosine mixed-frequency signals cosA2 and cosB2 inside the FPGA to obtain two cosine signals cosA3 and cosB3, which are respectively expressed as follows:

[0085]

[0086] Step 6, phase resolution.

[0087] 6.1) Perform phase resolution on the two sine signals sinA3 and sinB3 and the two cosine signals cosA3 and cosB3 inside the FPGA to obtain two sine and cosine signals sinC and cosC, which are respectively expressed as follows:

[0088]

[0089] 6.2) Calculate the integer part of the phase through the phase processing method of four fine-resolution phase discrimination for the two sine and cosine signals sinC and cosC:

[0090] 6.2.1) Assume that the object movement direction is positive, continuously detect the change of the sign bits of the two orthogonal signals sinC and cosC. Suppose the detected change trend is 11 - 10 - 00 - 01 - 11 cycling in turn, and define this direction as the positive direction; if the object movement direction is negative, the change trend of the sign bits of the two orthogonal signals sinC and cosC is 10 - 11 - 01 - 00 - 10 cycling in turn. Using this method, it is possible to distinguish whether the object is moving forward or backward:

[0091] 6.2.2) Detect the change of the sign bits of the two orthogonal signals sinC and cosC in real time to determine the phase change amount, that is, each time the sign bits of the two orthogonal signals sinC and cosC change, the corresponding phase changes by 90°;

[0092] 6.2.3) Calculate the integer part of the phase according to the object movement direction:

[0093] If the object is moving forward, add the phase change amount to the current integer part of the phase;

[0094] If the object is moving backward, subtract the phase change amount from the current integer part of the phase;

[0095] 6.3) Obtain the fractional part of the phase through the cordic algorithm:

[0096] 6.3.1) Divide the two sine and cosine signals sinC and cosC to obtain the tangent signal tan, and use it as the input of the cordic algorithm;

[0097] 6.3.2) Define the fractional part of the phase as the size of the angle between the vector in the two-dimensional coordinate system and the x-axis, and initialize the vector as Y0, so that Y0 coincides with the x-axis;

[0098] 6.3.3) Rotate Y0 counterclockwise by arctan(2 0) Obtain the vector Y1 after one rotation by rotating by a certain angle, and compare the ratio value1 of the abscissa to the ordinate of the vector Y1 with the input tangent signal tan:

[0099] If tan > value1, the next rotation is counterclockwise; otherwise, the next rotation is clockwise.

[0100] 6.3.4) Rotate the vector Y1 after one rotation by arctan(2 1 ) degrees to obtain the vector Y2 after two rotations, and so on. After n rotations, obtain the vector Y after n rotations n , and compare the ratio valuen of the abscissa to the ordinate of Y n with the input tangent signal tan:

[0101] If tan = valuen, add up the rotation angles each time, which is the fractional part of the phase.

[0102] Otherwise, continue to rotate the vector Y n after n rotations until tan = valuen is satisfied.

[0103] Step 7, relative displacement calculation.

[0104] 7.1) Add the fractional part of the phase to the integer part of the phase to obtain the overall phase value

[0105] 7.2) Calculate the relative displacement s through the overall phase value to complete the displacement measurement of heterodyne interference:

[0106]

[0107] where λ is the wavelength of the beam emitted by the frequency-stabilized laser.

[0108] The effects of the present invention can be further illustrated by the following simulation results.

[0109] One, simulation conditions

[0110] Use the matlab simulation software, set the frequency of the reference-end electrical signal f r to 10 MHZ, the frequency of the measurement-end electrical signal f m to 10.005 MHZ, the corresponding moving speed of the corner reflector P m is 1.5 mm / s, and the frequencies of the sine and cosine signals sin and cos generated inside the FPGA are set to 20 MHZ; set the sampling rate to 100 MHZ, the sampling resolution to 13 bits, select the FIR low-pass filter, set the window function to the Chebyshev window, the cut-off frequency to 8 MHZ, and the simulation time to 1 ms.

[0111] II. Simulation Content

[0112] Simulation 1: Under the above simulation conditions, the ranging accuracy graph of realizing communication and ranging integration by the existing method is as follows Figure 3 , where the abscissa represents the simulation time and the ordinate represents the ranging accuracy. From Figure 3 , it can be seen that the ranging accuracy of realizing communication and ranging integration by the existing method is at the micron level.

[0113] Simulation 2: The ranging accuracy graph of realizing communication and ranging integration by the method of the present invention is as follows Figure 4 , where the abscissa represents the simulation time and the ordinate represents the ranging accuracy. From Figure 4 , it can be seen that the ranging accuracy of realizing communication and ranging integration by the method of the present invention is at the picometer level. If an optical device part is introduced and a series of optical error compensation operations are performed, the ranging accuracy can theoretically reach the nanometer level.

[0114] Comparison Figure 3 and Figure 4 , the method of the present invention can improve the ranging accuracy of the traditional method from the micron level to the picometer level or the nanometer level, indicating that the present invention has extremely high displacement measurement accuracy.

Claims

1. A method for measuring the displacement of a communication and ranging integrated target based on a heterodyne interference system, the system comprising a frequency-stabilized laser, a beam splitting prism BS, a polarization beam splitting prism PBS, a reference corner reflector P r , a measurement corner reflector P m , a reference polarizer PA1, a measurement polarizer PA2, a photoelectric conversion device PT, and a field programmable gate array FPGA, characterized in that The implementation steps are as follows: (1) Generate two orthogonally polarized light beams E1(t) and E2(t) with frequencies f1 and f2 respectively by a frequency-stabilized laser. Use the light beam with frequency f2 as the carrier wave to modulate the baseband data d(t) through binary phase shift keying (BPSK) to obtain the modulated light beam E2′(t), and then combine it with the light beam E1(t) with frequency f1 to emit a combined light beam E; (2) Divide the path of the combined beam E transmitted through the beam splitter prism BS into a measurement end and a reference end, that is, the beam E that continues to be transmitted forward through the beam splitter prism BS r As the reference end, the beam E that is reflected by the beam splitter prism BS and then transmitted vertically downward m As the measurement end; (3) Interfere the reference-end light beam directly through the reference polarizer PA1, and then obtain the reference-end electrical signal f through the optoelectronic conversion device PT r ; (4) Obtain the electrical signal f at the measurement end m : (4a) Direct the measured light beam onto a polarization beam splitter prism (PBS) to split the light beam E into two orthogonal light beams with frequencies f1 and f2 respectively; (4b) Transmit the light beam with frequency f1 to the fixed reference corner reflector P r , and then reflect it back to the polarization beam splitter PBS; transmit the light beam with frequency f2 to the movable measurement corner reflector P m , obtain the light beam with frequency changed to f2±Δf and then reflect it back to the polarization beam splitter PBS, where; is the frequency difference caused by the Doppler frequency shift, v is the moving speed of the measurement corner reflector P m , and λ is the wavelength of the laser emitted by the frequency-stabilized laser; (4c)Interfere the two light beams that return to the polarization beam splitter PBS through the measurement polarizer PA2, and then convert them through the optoelectronic conversion device PT to obtain the electrical signal f at the measurement end m ; (6) Two orthogonal signals sin and cos generated internally in the FPGA through programming, and electrically connecting them to the reference terminal and the measurement terminal signals f r and f m respectively for mixing to obtain four mixed signals sinA1, cosA1, sinB1, and cosB1; (7) Mix the mixing signals sinA1 and cosA1, sinB1 and cosB1 respectively, and perform low-pass digital filtering on the mixing results inside the FPGA to obtain two sine signals sinA3 and sinB3; (8) Square and then subtract the above four mixing signals sinA1 and cosA1, sinB1 and cosB1 respectively, and then perform low-pass digital filtering inside the FPGA to obtain two cosine signals cosA3 and cosB3; (9) Perform phase calculation on the two sine signals sinA3, sinB3 and the two cosine signals cosA3, cosB3 inside the FPGA to obtain two sine and cosine signals sinC and cosC; (10) Calculate the relative distance between the measured targets based on the two sine and cosine signals sinC and cosC; (10a) Calculate the integer part of the phase through a four-fine-resolution phase processing method for the two sine and cosine signals sinC and cosC; (10b) Divide the two sine and cosine signals to obtain the signal tan, and obtain the fractional part of the phase through the cordic algorithm; (10c) Add the fractional part of the phase to the integer part of the phase to obtain the overall phase value Based on the overall phase value Calculate the relative distance s.

2. According to the method described in claim 1, wherein The modulated light beam E2′(t) obtained in step (1) is expressed as follows: Among them, D(t) is expressed as D r (t) and D m (t) respectively according to the asynchrony of the baseband data d(t) between the transmitter and the receiver under actual communication.

3. According to the method of claim 1, wherein In step (3), the reference light beam is directly interfered through a reference polarizer PA1 at an angle of 45 degrees with the polarized light, and the implementation is as follows: (3a) Assume that the light wave of the polarized light beam with frequency f1 is E1(t), and the light wave of the polarized light beam with frequency f2 is E2′(t). The expressions of E1 and E2 are as follows: where w1 and w2 are the angular frequencies of the optical waves E1 and E2 of the polarized light beams, respectively, and D r (t) represents the baseband data at the transmitting end: (3b) The two polarized light beams of optical waves E1(t) and E2′(t) are guided through the reference polarizer PA1 on the same plane to generate interference, resulting in an interference light beam: E R (t) = D r (t) cos(ω1t - ω2t).

4. According to the method as claimed in claim 1, wherein, In step (4c), the two light beams returned to the polarization beam splitter prism PBS are interfered through a measurement polarizer PA2, and the implementation is as follows: (4c1) Assume that the light wave of the polarized light beam with frequency f1 is expressed as E1, and the light wave of the polarized light beam with frequency f2±Δf is expressed as E3. The optical vibration equations of the two light beams are expressed as: wherein, w1 and w2+△w are the angular frequencies of the optical waves E1 and E3 of the polarized light beam, respectively, and D m (t) represents the baseband data at the receiving end: (4c2) Two polarized light beams E1(t) and E3 are guided through a measuring polarizer PA2 to generate interference in the same plane, resulting in an interference beam: E M (t) = D m (t) cos(ω1t - (ω2 + Δω)t).

5. According to the method as claimed in claim 1, wherein The four mixing signals obtained in step (6) are respectively expressed as follows: where, w r and w m are the angular frequencies of the reference terminal electrical signal f r and the measurement terminal electrical signal f m respectively, w3 is the angular frequency of the quadrature signals sin and cos generated inside the FPGA, D r (t) represents the baseband data at the transmitting end: D m (t) represents the baseband data at the receiving end:

6. According to the method described in claim 1, characterized in that, The two sine signals sinA3 and sinB3 obtained in step (7) are respectively expressed as follows: where, w r and w m are the angular frequencies of the reference - end electrical signal f r and the measurement - end electrical signal f m respectively, w3 is the angular frequency of the quadrature signals sin and cos generated inside the FPGA, D r (t) represents the base - band data at the transmitting end: D m (t) represents the base - band data at the receiving end:

7. According to the method described in claim 1, characterized in that, The two cosine signals cosA3 and cosB3 obtained in step (8) are respectively expressed as follows: where, w r and w m are the angular frequencies of the reference - end electrical signal f r and the measurement - end electrical signal f m respectively, w3 is the angular frequency of the quadrature signals sin and cos generated inside the FPGA, D r (t) represents the base - band data at the transmitting end: D m (t) represents the base - band data at the receiving end:

8. According to the method described in claim 1, wherein The two sine and cosine signals sinC and cosC obtained in step (9) are expressed as follows: sinC = sin2(w m t - w r t) = sin[2(w m t - w3t) - 2(w r t - w3t)] = sin2(w m t - w3t)cos2(w r t - w3t) - cos2(w m t - w3t)sin2(w r t - w3t) = 32[cosB3sinA3 - sinB3cosA3] cosC = cos2(w m t - w r t) = cos[2(w m t - w3t) - 2(w r t - w3t)] = cos2(w m t - w3t)cos2(w r t - w3t)+sin2(w m t - w3t)sin2(w r t - w3t) = 64sinB3×sinA3 + 16cosB3×cosA3 Among them, w r and w m are the angular frequencies of the reference terminal electrical signal f r and the measurement terminal electrical signal f m respectively, and w3 is the angular frequency of the orthogonal signals sin and cos generated inside the FPGA.

9. According to the method as claimed in claim 1, wherein, In step (10a), based on the two-channel sine and cosine signals sinC and cosC, the integer part of the phase is calculated through a four-fine-resolution phase processing algorithm, and the implementation is as follows: (10a1) Assume that the object's moving direction is positive. Continuously detect the changes in the sign bits of the two orthogonal signals sinC and cosC. Suppose the change trend is found to be 11 - 10 - 00 - 01 - 11 in a cyclic manner. This direction is defined as the positive direction. If the object's moving direction is negative, the change trend of the sign bits of the two orthogonal signals sinC and cosC is 10 - 11 - 01 - 00 - 10 in a cyclic manner. Using this method, it is possible to distinguish whether the object is moving forward or backward: (10a2) Continuously detect the changes in the sign bits of the two orthogonal signals sinC and cosC to determine the phase change amount. That is, each time the sign bits of the two orthogonal signals sinC and cosC change, the corresponding phase changes by 90°; (10a3) Calculate the integer part of the phase according to the object's moving direction: If the object is moving forward, add the phase change amount to the current integer part of the phase; If the object is moving backward, subtract the phase change amount from the current integer part of the phase.

10. According to the method described in claim 1, wherein In step (10b), divide the two-channel sine and cosine signals sinC and cosC to obtain the signal tan, and use it as the input of the cordic algorithm. The fractional part of the phase, that is, the arctangent value of the signal tan, is obtained through the cordic algorithm. The implementation is as follows: (10b1) Divide the two-channel sine and cosine signals sinC and cosC to obtain the tangent signal tan, and use it as the input of the cordic algorithm; (10b2) Define the fractional part of the phase as the angle between the vector in the two-dimensional coordinate system and the x-axis. Initialize the vector as Y0 so that Y0 coincides with the x-axis; (10b3) Rotate Y0 counterclockwise by arctan(2 0 ) degrees to obtain the vector Y1 after the first rotation. Compare the ratio value1 of the abscissa to the ordinate of the vector Y1 with the input tangent signal tan: If tan > value1, the next rotation is counterclockwise. Conversely, the next rotation is clockwise; (10b4) Rotate the vector Y1 after the first rotation by arctan(2 1 ) degrees to obtain the vector Y2 after the second rotation, and so on. After n rotations, the vector Y after n rotations is obtained n , and compare the ratio valuen of the abscissa to the ordinate of Y n with the input tangent signal tan: If tan = valuen, add the rotation angles each time, which is the fractional part of the phase; Otherwise, the vector Y after n rotations n is further rotated until tan = value n is satisfied.

11. According to the method described in claim 1, characterized in that In step (10c), based on the overall phase value calculate the relative distance s, and the formula is as follows: In the formula, λ is the wavelength of the beam emitted by the frequency-stabilized laser.

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