Permanent magnet synchronous motor speed loop control method based on differential feedforward and parameter self-tuning

Abstract

The application discloses a permanent magnet synchronous motor speed loop control method based on differential feedforward and parameter self-tuning. Under the premise of knowing motor pole pair number, rotational inertia and permanent magnet flux linkage, a zero-pole cancellation speed loop parameter self-tuning scheme is adopted. The speed filter delay, speed control algorithm calculation delay and current loop equivalent delay are fully considered and added up as the effective delay time of the speed loop. The stiffness coefficient and the medium frequency phase angle coefficient are introduced to adjust the system stiffness and the phase angle margin of the speed loop respectively. The self-tuning parameters are discretely designed through a bilinear transformation mode to obtain the final speed loop PI control parameters. Meanwhile, the feedforward compensation of the speed loop is obtained by combining the low-pass filter-based differential feedforward algorithm. The parameter self-tuning process of the method is simple, the differential feedforward compensation is stable and reliable, one-key parameter self-tuning can be realized, the performance of the self-tuned parameters is better, the method has universality, the control parameters match the control system and the control effect is excellent.

Description

Technical Field

[0001] This invention belongs to the field of permanent magnet synchronous motor control technology, and in particular, it is a speed loop control method for permanent magnet synchronous motors that combines differential feedforward based on low-pass filtering and parameter self-tuning based on zero-pole cancellation. Background Technology

[0002] Servo control systems, as one of the supporting technologies for modern industrial automation and motion control, are widely used in industrial fields such as machine tools, textiles, robotic arms, and robots, as well as military fields such as aviation, aerospace, and marine applications, due to their advantages such as high-precision control performance, high dynamic performance, and wide application range. On the one hand, the speed loop control of traditional permanent magnet synchronous servo motors generally adopts PI controllers. However, due to the complexity of actual working conditions and the tediousness of manual parameter tuning, existing technologies require highly specialized knowledge from the tuning personnel, are time-consuming, and have high labor costs. Moreover, the performance of manually tuned parameters is not necessarily optimal and often fails to meet performance requirements. When load conditions change or the system filter delay changes, repeated adjustments may be necessary. Therefore, parameter self-tuning is of great significance for the mass application and general-purpose design of permanent magnet synchronous servo motor drivers.

[0003] On the other hand, due to system delay issues, the PI parameters derived from parameter self-tuning often exhibit a high amplitude frequency but a significantly lower phase frequency. Appropriate feedforward compensation for the speed loop can effectively solve this problem. Therefore, a safe, reliable, versatile, and effective speed loop control strategy is urgently needed. Summary of the Invention

[0004] The purpose of this invention is to address the shortcomings of existing technologies by providing a speed loop control method for permanent magnet synchronous motors based on differential feedforward and parameter self-tuning, which can achieve speed loop control when the number of pole pairs P of the motor is known. n Moment of inertia J, permanent magnet flux linkage ψ f Under the premise of [specific conditions], the PI parameters of the speed loop with better performance are tuned, and the bandwidth and performance of the speed loop are further improved by combining differential feedforward with low-pass filtering.

[0005] The technical solution to achieve the purpose of this invention is: a speed loop control method for permanent magnet synchronous motors based on differential feedforward and parameter self-tuning, characterized by including: speed loop differential feedforward compensation based on low-pass filtering, given the known number of pole pairs P of the motor. n Moment of inertia J, permanent magnet flux linkage ψ f Given the speed command value ω* m Differential feedforward compensation is performed on the speed loop to improve its bandwidth. Specifically, this includes the following steps:

[0006] Step 1: Construct the given current equation for the permanent magnet synchronous motor. This equation includes a feedforward term for the speed given differential and a PI feedback term for the speed error.

[0007] Step 2: Obtain the feedforward term using a differential feedforward method based on low-pass filtering;

[0008] Step 3: Obtain the PI feedback term based on parameter self-tuning of zero-pole cancellation.

[0009] Furthermore, step 1 specifically includes the following steps:

[0010] Step 1-1, based on the mechanical motion equation of the motor:

[0011]

[0012] The relationship between the motor speed and current is obtained as follows:

[0013]

[0014] Where, ω m It is the mechanical angular velocity of the motor, i q J is the electric current, K is the moment of inertia. t =1.5P n ψ f It is the motor torque coefficient, T L Where B is the load torque and B is the viscous friction coefficient.

[0015] Steps 1-2: Define the speed tracking error Pick in For a given rotational speed, Given a current, then

[0016]

[0017] Among them, K p K is the proportionality coefficient. i The integral coefficient;

[0018] Steps 1-3, rearranging equation (3), yield the given current:

[0019]

[0020] Ignoring the effects of load torque and friction, rewriting equation (4) yields:

[0021]

[0022] In the formula, For the feedforward term, k' p e+k' i ∫edt is the PI feedback term, k'p and k' i Here are the velocity loop PI parameters, where k' p k' is the proportional gain of the speed loop controller. i This represents the integral coefficient of the speed loop controller.

[0023] Furthermore, the step 2, which involves obtaining the feedforward term using a differential feedforward method based on low-pass filtering, specifically includes:

[0024] Step 2-1, using the differential of the given velocity Feedforward compensation is performed, and backward differential is used to... Discretization yields:

[0025]

[0026] in, The speed difference between adjacent control cycles is given. Given a value for the velocity at the current time k, Given the velocity at time k-1 in the previous moment, T ns For speed loop control cycle;

[0027] Step 2-2: Use a first-order low-pass filter to measure the speed difference between adjacent control cycles. Filtering is performed, with the first-order low-pass filter shown below:

[0028]

[0029] Among them, Y n Y is the output of the first-order low-pass filter at the current moment. n-1 X is the output of the first-order low-pass filter at the previous time step. n Let be the input to the first-order low-pass filter at the current moment, and 'a' be the coefficient of the first-order low-pass filter. The cutoff frequency of the first-order low-pass filter is determined by... To design, T s To control the cycle, The speed difference between adjacent control cycles before low-pass filtering. The speed difference between adjacent control cycles after low-pass filtering;

[0030] Step 2-3, based on steps 2-1 and 2-2, updates the velocity differential feedforward term as follows:

[0031]

[0032] Furthermore, step 3, which involves obtaining the PI feedback term through parameter self-tuning based on zero-pole cancellation, specifically includes:

[0033] Step 3-1, establish the transfer function of the speed loop controller as follows:

[0034]

[0035] Among them, K PN τ is the proportional coefficient of the speed loop controller. N The integral time constant of the speed loop controller;

[0036] Step 3-2: Introduce the stiffness coefficient α, and design the controller parameters according to the design rules of the symmetric optimal method:

[0037]

[0038] Among them, T NN T is the system reset time. Σ,N This is the effective delay time of the speed loop;

[0039] At the same time, the gain crossover frequency ω of the speed loop control system will be... cN Designed as follows:

[0040]

[0041] Phase margin φ RN Designed as follows:

[0042]

[0043] Step 3-3, using zero-pole cancellation to achieve self-tuning of velocity loop parameters, specifically includes:

[0044] Design the normalized gain crossover frequency Ω:

[0045]

[0046] A bandwidth gain factor γ is introduced, which is calculated from the normalized gain crossover frequency Ω, as shown below:

[0047] γ=max{γ BA ,γ Bφ} (14)

[0048] in,

[0049]

[0050] γ Bφ =Ω Bφ sinΩ Bφ (16)

[0051] Where, γ BA , These are the bandwidth gain coefficients for the amplitude frequency and phase frequency, respectively, in Ω. BA , These are the normalized gain crossover frequencies corresponding to the amplitude frequency and phase frequency, respectively;

[0052] Steps 3-4: Calculate the proportional gain of the speed loop controller based on the bandwidth gain coefficient γ.

[0053]

[0054] Among them, K PN The proportional coefficient of the speed loop controller, i.e., k' p ;

[0055] Steps 3-5 introduce the intermediate frequency bandwidth parameter h:

[0056]

[0057] Calculate the integral coefficients of the speed loop controller based on the intermediate frequency bandwidth parameter h:

[0058]

[0059] Among them, K IN k' is the integral coefficient of the speed loop controller. i .

[0060] Furthermore, the effective delay time T of the speed loop Σ,N for:

[0061] T E,N =T FN +T AN +T EI (20)

[0062] In the formula, T FN For a first-order time delay with a time constant, T AN To calculate the delay, T EI This is the equivalent delay for the current loop.

[0063] Furthermore, the intermediate frequency bandwidth parameter h in steps 3-5 should satisfy:

[0064] Furthermore, at the cutoff frequency ω cN At this point, the hysteresis phase angle θ provided by the PI controller for the entire servo system PI Determined by the intermediate frequency phase angle factor h,

[0065] Furthermore, step 3 also includes:

[0066] Steps 3-6 involve discretizing the self-tuning parameters after pole-zero cancellation using a bilinear transformation, mapping the transfer function of equation (9) from the s-plane to the z-plane, i.e.

[0067] The discretized velocity loop PI parameters are shown below:

[0068]

[0069] Among them, K PNdis K INdis These are the discretized velocity loop PI parameters, corresponding to k' respectively. p and k' i .

[0070] Compared with the prior art, the significant advantages of this invention are:

[0071] (1) Under the condition of known motor parameters, the parameter self-tuning process of this invention is simple, the differential feedforward compensation is stable and reliable, one-click parameter self-tuning can be realized, the tuned parameter performance is better, and it has versatility.

[0072] (2) The present invention adopts velocity loop differential feedforward compensation based on low-pass filtering, which avoids the instability caused by differentiation or even control divergence, making the feedforward compensation more stable and reliable.

[0073] (3) The present invention adopts a speed loop parameter self-tuning scheme with zero-pole cancellation. By designing the zero point of the PI controller to eliminate the poles corresponding to the large time constant, and fully considering the influence of system delay on speed loop control, the control parameters can still match when the speed filter parameters are adjusted, and the control effect is good.

[0074] (4) The parameter self-tuning adopted in this invention introduces stiffness coefficient and intermediate frequency phase angle coefficient, which fully considers the influence of load condition changes and system filter delay changes on control parameters. The stiffness of the system can be changed by adjusting the stiffness coefficient and the phase margin can be changed by adjusting the intermediate frequency phase angle coefficient, so as to obtain the parameters of the required control performance.

[0075] The present invention will now be described in further detail with reference to the accompanying drawings. Attached Figure Description

[0076] Figure 1 This is a speed loop control block diagram based on differential feedforward and parameter self-tuning.

[0077] Figure 2 This is the flowchart of parameter self-tuning control.

[0078] Figure 3 This is the flowchart of differential feedforward control.

[0079] Figure 4 This is the velocity loop phase margin design diagram.

[0080] Figure 5 This is the speed loop bandwidth design diagram.

[0081] Figure 6 These are the closed-loop Bode plots of velocity under different stiffness coefficients, where... Figure 6 (a) shows the amplitude-frequency response under different stiffness coefficients. Figure 6 (b) shows the phase frequency characteristics under different stiffness coefficients.

[0082] Figure 7 The closed-loop Bode plots of velocity are given for different intermediate frequency phase angles with α=3. Figure 7 (a) shows the amplitude-frequency response under different intermediate frequency phase angle coefficients. Figure 7 (b) shows the phase frequency characteristics under different intermediate frequency phase angle coefficients.

[0083] Figure 8 The velocity closed-loop Bode plots are given for different feedforward ratios α=3 and h=5, where... Figure 8 (a) shows the amplitude-frequency response under different feedforward ratios. Figure 8 (b) shows the phase frequency characteristics under different feedforward ratios.

[0084] Figure 9 The Bode plots are velocity closed-loop diagrams with different velocity filtering delays (α=3, h=5, 25% feedforward). Figure 9 (a) shows the amplitude-frequency response under different speed filtering delays. Figure 9 (b) shows the phase frequency characteristics under different speed filtering delays. Detailed Implementation

[0085] To make the objectives, technical solutions, and advantages of this application clearer, the following detailed description is provided in conjunction with the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and not intended to limit the scope of this application.

[0086] This invention provides a speed loop control method for permanent magnet synchronous motors based on differential feedforward and parameter self-tuning. The basic principle is as follows: Given the number of pole pairs P of the motor... n Moment of inertia J, permanent magnet flux linkage ψ f Under the premise of zero-pole cancellation, a speed loop parameter self-tuning scheme is adopted, which fully considers the speed filtering delay, speed control algorithm calculation delay, and current loop equivalent delay, and sums them into the effective delay time T of the speed loop. Σ,N Two adjustable control parameters, stiffness coefficient α and intermediate frequency phase angle coefficient h, are introduced to adjust the system stiffness and phase margin of the speed loop, respectively. Then, the self-tuning parameters after zero-pole cancellation are discretized using a bilinear transformation to obtain the final speed loop PI control parameters. The speed command is obtained, differentiated, and discretized to obtain... To improve the stability and reliability of differential feedforward compensation, a first-order low-pass filter is used. Filtering is performed Differential feedforward compensation is obtained by combining the differential feedforward algorithm. Combining differential feedforward based on low-pass filtering and parameter self-tuning based on zero-pole cancellation yields a complete speed loop control method.

[0087] In one embodiment, a speed loop control method for a permanent magnet synchronous motor based on differential feedforward and parameter self-tuning is provided, and its control block diagram is shown below. Figure 1 As shown, the method specifically includes the following steps:

[0088] (1) According to the mechanical motion equation of the motor:

[0089]

[0090] The relationship between the motor speed and current is obtained as follows:

[0091]

[0092] Where, ω m It is the mechanical angular velocity of the motor, i q J is the electric current, K is the moment of inertia. t It is the motor torque coefficient, T L It is the load torque, and B is the viscous friction coefficient.

[0093] (2) Define speed tracking error Pick in For a given rotational speed, Given a current, then

[0094]

[0095] Among them, K p K is the proportionality coefficient. i The integral coefficient;

[0096] By rearranging equation (24), we can obtain the given current:

[0097]

[0098] Ignoring the effects of load torque and friction, rewriting equation (25) yields:

[0099]

[0100] As can be seen from the above formula, given current The output of the speed loop and the input of the current loop consist of two parts: a feedforward term containing the differential of the given speed. PI feedback term k' including speed error p e+k' i ∫edt. Where the PI feedback term k'p e+k' i ∫edt can be obtained using a parameter self-tuning scheme based on zero-pole cancellation, and its control flowchart is as follows: Figure 2 As shown; feedforward term This can be achieved using a differential feedforward scheme based on low-pass filtering, and its control flowchart is as follows: Figure 3 As shown.

[0101] (3) The transfer function of the speed loop controller is established as follows:

[0102]

[0103] Among them, K PN τ is the proportional coefficient of the speed loop controller. N The integral time constant of the speed loop controller;

[0104] Introducing a stiffness coefficient α (α ranges from 2 to 4), and based on the design rules of the symmetric optimal method, the controller parameters are designed as follows:

[0105]

[0106] Among them, T NN T is the system reset time. Σ,N This is the effective delay time of the speed loop;

[0107] Effective delay time T of the speed loop Σ,N It can be divided into three parts: 1. In order to make the speed signal smoother and suppress the spurious signal from the sensor, a speed filter is used, and the speed filter uses a first-order time delay T with a time constant. FN This indicates that: 2. The speed control algorithm will cause an additional computational delay T. AN 3. Equivalent delay T of the current loop for inner loop torque generation current control EI That is, the effective delay time T of the speed loop. Σ,N As shown in the following formula:

[0108] T E,N =T FN +T AN +T EI (29)

[0109] In the formula, T FN For a first-order time delay with a time constant, T AN To calculate the delay, T EI This is the equivalent delay for the current loop.

[0110] like Figure 9 The figure shows the closed-loop Bode plot of the velocity under different velocity filter delays, i.e., the first-order time delay T caused by the use of the velocity filter. FNThe difference is evident from the Bode plot; as the velocity filter delay increases, the amplitude and phase frequencies of the velocity loop decrease synchronously.

[0111] (4) After introducing the stiffness coefficient α, the design value of the gain crossover frequency of the speed loop control system is: The design value for phase margin is: like Figure 4 The diagram shown is the velocity loop phase margin design diagram. To meet the phase margin design requirements, the stiffness coefficient α is taken as [2, 4], corresponding to a phase margin of... The design values ​​are [36.87°, 61.93°]; to achieve excellent system control, the following values ​​are used. At this time, α = 3.

[0112] (5) A zero-pole cancellation scheme is adopted, which eliminates the poles corresponding to larger time constants by designing the zero point of the PI controller to achieve self-tuning of the speed loop parameters:

[0113] Design the normalized gain crossover frequency Ω:

[0114]

[0115] A bandwidth gain factor γ is introduced, which is calculated from the normalized gain crossover frequency Ω, as shown below:

[0116] γ=max{γ BA ,γ Bφ} (31)

[0117] in,

[0118]

[0119] γ Bφ =Ω Bφ sinΩ Bφ (33)

[0120] Where, γ BA , These are the bandwidth gain coefficients for the amplitude frequency and phase frequency, respectively, in Ω. BA , These are the normalized gain crossover frequencies corresponding to the amplitude frequency and phase frequency, respectively.

[0121] Figure 5 The diagram shows the design bandwidth of the velocity loop under a specific system delay. The system's design bandwidth is positively correlated with the bandwidth gain coefficient γ and inversely proportional to the stiffness coefficient α. When the stiffness coefficient α = 3, the system's design bandwidth is 106 Hz under the current system delay.

[0122] Figure 6 The figure shows the closed-loop Bode plots of velocity under different stiffness coefficients, and... Figure 5The speed loop bandwidth design is matched, and the system's amplitude and phase frequencies are maximized at α=2, but the system will also enter the divergence region. Considering the system's phase margin, the recommended value α=3 provides the best control performance.

[0123] Calculate the proportional gain of the speed loop controller based on the bandwidth gain coefficient γ:

[0124]

[0125] Among them, K PN The proportional coefficient of the speed loop controller, i.e., k' p .

[0126] (6) To reduce the complexity of the algorithm, an intermediate frequency bandwidth parameter is introduced. At the cutoff frequency ω cN At this point, the hysteresis phase angle θ provided by the PI controller for the entire servo system PI Determined by the intermediate frequency phase angle factor h,

[0127] When the intermediate frequency phase angle coefficient h takes different values, the phase lag caused by the PI controller to the servo system varies, which in turn determines the phase margin of the system. To maximize the phase margin at the cutoff frequency, the intermediate frequency phase angle coefficient h should satisfy the following:

[0128] Calculate the integral coefficients of the speed loop controller based on the intermediate frequency bandwidth parameter h:

[0129]

[0130] Among them, K IN k' is the integral coefficient of the speed loop controller. i .

[0131] The self-tuning parameters after pole-zero cancellation are discretized using a bilinear transformation, mapping the transfer function of equation (27) from the s-plane to the z-plane, i.e.

[0132] The discretized velocity loop PI parameters are shown below:

[0133]

[0134] Among them, K PNdis K INdis These are the discretized velocity loop PI parameters, corresponding to k' respectively. p and k' i .

[0135] Figure 7The figure shows the closed-loop Bode plots of the velocity system under different intermediate frequency phase angles h. The amplitude-frequency response of the system is almost negligible due to the intermediate frequency phase angle h. However, as the intermediate frequency phase angle h increases, the integral coefficient K... IN It is inversely proportional to h, which leads to a decrease in the integral term, and thus the phase frequency of the system also decreases.

[0136] (7) Due to system delay, a single PI controller often cannot meet the phase frequency characteristics of the system. Therefore, feedforward compensation is needed to improve system performance. This is achieved by utilizing the derivative of the given speed. Feedforward compensation is performed using a backward differential method. Discretization yields:

[0137]

[0138] in, The speed difference between adjacent control cycles is given. Given a value for the velocity at the current time k, Given the velocity at time k-1 in the previous moment, T ns This is the speed loop control cycle.

[0139] (8) Using a first-order low-pass filter to... Filtering is performed to ensure the stability and reliability of the velocity loop with added differential feedforward compensation. The first-order low-pass filter is shown below:

[0140]

[0141] Among them, Y n Y is the output of the first-order low-pass filter at the current moment. n-1 X is the output of the first-order low-pass filter at the previous time step. n Let be the input to the first-order low-pass filter at the current moment, and 'a' be the coefficient of the first-order low-pass filter. The cutoff frequency of the first-order low-pass filter is determined by... To design, T s To control the cycle, The speed difference between adjacent control cycles before low-pass filtering. This is the speed difference between adjacent control cycles after low-pass filtering.

[0142] (9) The velocity differential feedforward term after discretization and low-pass filtering can be updated as shown in the following equation:

[0143]

[0144] Figure 8 The figure shows the velocity closed-loop Bode plots under different feedforward ratios for comparison. Figure 7After adding differential feedforward compensation, the system's amplitude and phase frequency characteristics are greatly improved. A larger feedforward ratio results in higher amplitude and phase frequencies. However, a large feedforward ratio can cause system oscillations or even divergence. At the recommended feedforward ratio of 25%, ... Figure 9 As shown, the system bandwidth has been increased by approximately 100%. With this, the design of the speed loop control system based on parameter self-tuning with zero-pole cancellation and differential feedforward compensation using low-pass filtering is complete.

[0145] The method described in this invention has a simple parameter self-tuning process, stable and reliable differential feedforward compensation, and can achieve one-click parameter self-tuning. The tuned parameters have good performance, are versatile, and the control parameters match the control system, resulting in excellent control effect.

[0146] The foregoing has shown and described the basic principles, main features, and advantages of the present invention. Those skilled in the art should understand that the present invention is not limited to the above embodiments. The embodiments and descriptions in the specification are merely illustrative of the principles of the invention. Any modifications, equivalent substitutions, or improvements made within the spirit and principles of the present invention without departing from its spirit and scope should be included within the protection scope of the present invention.

Claims

1. A speed loop control method for a permanent magnet synchronous motor based on differential feedforward and parameter self-tuning, characterized in that, The method includes: Step 1: Construct the given current equation for the permanent magnet synchronous motor. This equation includes a feedforward term for the speed given differential and a PI feedback term for the speed error. Step 2: Obtain the feedforward term using a differential feedforward method based on low-pass filtering; Step 3: Obtain the PI feedback term based on parameter self-tuning of zero-pole cancellation; Step 1 specifically includes the following steps: Step 1-1, based on the mechanical motion equation of the motor: The relationship between the motor speed and current is obtained as follows: Where, ω m It is the mechanical angular velocity of the motor, i q J is the q-axis current, J is the moment of inertia, and K is the q-axis current. t It is the motor torque coefficient, T L Where B is the load torque and B is the viscous friction coefficient. Steps 1-2: Define the speed tracking error Pick in For a given rotational speed, Given a current, then Among them, K p K is the proportionality coefficient. i The integral coefficient; Steps 1-3, rearranging equation (3), yield the given current: Ignoring the effects of load torque and friction, rewriting equation (4) yields: In the formula, For the feedforward term, k' p e+k' i ∫edt is the PI feedback term, k' p and k' i Here are the velocity loop PI parameters, where k' p k' is the proportional gain of the speed loop controller. i The integral coefficient of the speed loop controller; Step 3, which describes obtaining the PI feedback term through parameter self-tuning based on zero-pole cancellation, specifically includes: Step 3-1, establish the transfer function of the speed loop controller as follows: Among them, K PN τ is the proportional coefficient of the speed loop controller. N The integral time constant of the speed loop controller; Step 3-2: Introduce the stiffness coefficient α, and design the controller parameters according to the design rules of the symmetric optimal method: Among them, T NN T is the system reset time. Σ,N This is the effective delay time of the speed loop; At the same time, the gain crossover frequency ω of the speed loop control system will be... cN Designed as follows: Phase margin Designed as follows: Step 3-3, using zero-pole cancellation to achieve self-tuning of velocity loop parameters, specifically includes: Design the normalized gain crossover frequency Ω: A bandwidth gain factor γ is introduced, which is calculated from the normalized gain crossover frequency Ω, as shown below: γ=max{γ BA ,c Bφ } (11) in, c Bφ =Oh Bφ sinΩ Bφ (13) Where, γ BA , These are the bandwidth gain coefficients for the amplitude frequency and phase frequency, respectively, in Ω. BA , These are the normalized gain crossover frequencies corresponding to the amplitude frequency and phase frequency, respectively; Steps 3-4: Calculate the proportional gain of the speed loop controller based on the bandwidth gain coefficient γ. Among them, K PN The proportional coefficient of the speed loop controller, i.e., k' p ; Steps 3-5 introduce the intermediate frequency bandwidth parameter h: Calculate the integral coefficients of the speed loop controller based on the intermediate frequency bandwidth parameter h: Among them, K IN k' is the integral coefficient of the speed loop controller. i .

2. The speed loop control method for permanent magnet synchronous motors based on differential feedforward and parameter self-tuning according to claim 1, characterized in that, Step 2, which describes obtaining the feedforward term using a differential feedforward method based on low-pass filtering, specifically includes: Step 2-1, using the differential of the given velocity Feedforward compensation is performed, and backward differential is used to... Discretization yields: in, The speed difference between adjacent control cycles is given. Given a value for the velocity at the current time k, Given the velocity at time k-1 in the previous moment, T ns For speed loop control cycle; Step 2-2: Use a first-order low-pass filter to measure the speed difference between adjacent control cycles. Filtering is performed, with the first-order low-pass filter shown below: Among them, Y n Y is the output of the first-order low-pass filter at the current moment. n-1 X is the output of the first-order low-pass filter at the previous time step. n Let be the input to the first-order low-pass filter at the current moment, and 'a' be the coefficient of the first-order low-pass filter. The cutoff frequency of the first-order low-pass filter is determined by... To design, T s To control the cycle, The speed difference between adjacent control cycles before low-pass filtering. The speed difference between adjacent control cycles after low-pass filtering; Step 2-3, based on steps 2-1 and 2-2, updates the velocity differential feedforward term as follows: (19)。 3. The speed loop control method for permanent magnet synchronous motors based on differential feedforward and parameter self-tuning according to claim 1, characterized in that, The effective delay time T of the velocity loop Σ,N for: T Σ,N =T FN +T AN +T EI (20) In the formula, T FN For velocity filtering delay, T AN For calculating the delay in the speed control algorithm, T EI This is the equivalent delay for the current loop.

4. The speed loop control method for permanent magnet synchronous motors based on differential feedforward and parameter self-tuning according to claim 1, characterized in that, In step 3-2, the stiffness coefficient α takes the value of 2 to 4.

5. The speed loop control method for permanent magnet synchronous motors based on differential feedforward and parameter self-tuning according to claim 1, characterized in that, The intermediate frequency bandwidth parameter h in step 3-5 should meet the following requirements:

6. The speed loop control method for permanent magnet synchronous motors based on differential feedforward and parameter self-tuning according to claim 1, characterized in that, At the cutoff frequency ω cN At this point, the hysteresis phase angle θ provided by the PI controller for the entire servo system PI Determined by the intermediate frequency phase angle factor h, 7. The speed loop control method for permanent magnet synchronous motors based on differential feedforward and parameter self-tuning according to claim 2, characterized in that, Step 3 also includes: Steps 3-6 involve discretizing the self-tuning parameters after pole-zero cancellation using a bilinear transformation, mapping the transfer function of equation (6) from the s-plane to the z-plane, i.e. The discretized velocity loop PI parameters are shown below: Among them, K PNdis K INdis These are the discretized velocity loop PI parameters, corresponding to k' respectively. p and k' i .

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