A planar waveguide structure

Abstract

The application discloses a planar waveguide structure, which comprises a microstrip line, a dielectric layer and a metal layer, the microstrip line is used for transmitting digital signals, and the single side or double side edge of the microstrip line is arranged as a periodic structure, and the microstrip line has a constant characteristic impedance. The planar waveguide structure provided by the application has a characteristic impedance close to the characteristic impedance of a traditional microstrip line and has a strong ability of suppressing electromagnetic interference.

Description

Technical Field

[0001] This invention belongs to the field of waveguide technology, and specifically relates to a planar waveguide structure. Background Technology

[0002] Traditional microstrip lines (CMLs) have long served as the primary means of signal transmission in planar circuits. In recent years, the decreasing rise time of digital signals in high-speed planar circuits has created a conflict between miniaturization and high speed in product design. The shortened rise time of digital signals leads to severe electromagnetic interference (EMI), and crosstalk occurs in various areas of the circuit board. Therefore, for decades, numerous researchers have dedicated themselves to developing effective methods for isolating crosstalk between high-speed loops.

[0003] To quantitatively analyze the electromagnetic coupling between two planar transmission lines, some researchers have used classical transmission line theory to analyze the S-parameters between the two coupled microstrip lines; others have used circuit models and the finite-difference time-domain (FDTD) method to analyze crosstalk between the two microstrip lines. Currently, the known causes of crosstalk between adjacent loops are: 1) two parallel microstrip lines being too close together; 2) a shortened rise time of the digital signal transmitted on the microstrip line; and 3) excessively high frequency of the harmonic signal.

[0004] To mitigate electromagnetic interference (EMI), various methods for isolating crosstalk between two microstrip lines have been proposed. One of the most direct approaches is to increase the spacing between the two microstrip lines. Researchers have analyzed the impact of this spacing on far-end and near-end crosstalk. Theoretical analysis and experimental measurements show that increasing the spacing does indeed help reduce electromagnetic coupling between the two parallel microstrip lines. However, this method of isolating EMI by increasing the microstrip line spacing is no longer sufficient to meet the miniaturization requirements of today's high-speed circuit products. Introducing a grounded guard trace between two parallel microstrip lines can attract some of the electric field lines in the signal lines, effectively suppressing EMI between the two microstrip lines. Therefore, this type of grounded guard trace for EMI isolation is widely used in many commercial products. Furthermore, this type of grounded guard trace can also be applied to differential circuit systems to reduce crosstalk between differential pairs. However, this grounded guard trace is not without its drawbacks. When the spacing between adjacent ground holes in the guard trace is too large, it can cause resonant coupling between the two signal transmission lines, and as the signal frequency continues to increase, the effectiveness of the grounded guard trace in isolating EMI gradually weakens. Furthermore, when using grounding protection wires to isolate electromagnetic interference between adjacent microstrip lines, the spacing between the two microstrip lines needs to be increased to accommodate the grounding protection wires. This makes the grounding protection wires extremely unfavorable for circuit miniaturization. Summary of the Invention

[0005] The purpose of this invention is to provide a planar waveguide structure whose characteristic impedance is close to that of a traditional microstrip line and has a strong ability to suppress electromagnetic interference.

[0006] To achieve the above objectives, the technical solution adopted by the present invention is as follows:

[0007] A planar waveguide structure includes a microstrip line, a dielectric layer, and a metal layer. The microstrip line is used to transmit digital signals. One or both edges of the microstrip line are configured with a periodic structure, and the microstrip line has a constant transient impedance. The lattice constant d of the periodic structure is:

[0008]

[0009] In the formula, v is the transmission rate of the digital signal in the microstrip line, and R... T Let R be the rise time of the digital signal, and R be the rise time of the signal. T ≤30ps.

[0010] Several alternative methods are provided below, but they are not intended as additional limitations on the overall solution above. They are merely further additions or optimizations. Provided there are no technical or logical contradictions, each alternative method can be combined individually with respect to the overall solution above, or multiple alternative methods can be combined with each other.

[0011] Preferably, the microstrip line is equivalent to an equivalent network that periodically modulates the inductance and capacitance. The characteristic impedance of the equivalent network is obtained using the input impedance method based on the unit cell of the periodic structure, as follows:

[0012] The input impedance Z of a unit cell in a periodic structure in Represented as:

[0013]

[0014] Then the input impedance Z in Characteristic impedance It can be represented as:

[0015]

[0016] Among them, Z p It is the capacitance impedance, Z s and Y p These are the inductance and impedance Z. s =jωL and capacitance admittance Y p =jωC, where j is the imaginary unit, ω is the angular frequency, L is the inductance, and C is the capacitance.

[0017] Preferably, the resonant frequency of the LC circuit is defined as ω. o :

[0018]

[0019] Define Z s With Z p The square root of the product is Z. o :

[0020]

[0021]

[0022]

[0023] The characteristic impedance expression is:

[0024]

[0025] and

[0026]

[0027] in, and These are the forward Block characteristic impedance and the backward Block characteristic impedance, respectively. and respectively with Z o They differ by only one phase factor.

[0028] Preferably, the unit cell of the periodic structure contains grooves etched by the edges of the microstrip lines.

[0029] Preferably, the width of the groove is half of the lattice constant.

[0030] Preferably, the unit cell has at least one groove.

[0031] Preferably, when the two sides of the microstrip line are set as a periodic structure, the periodic structure of the two sides is symmetrical.

[0032] Preferably, when the two sides of the microstrip line are set as a periodic structure, the periodic structure of the two sides is asymmetrical.

[0033] This invention provides a planar waveguide structure, constructed by etching periodic structures along the edges of a microstrip line. By adjusting the lattice constant of the microstrip line, a specified characteristic impedance can be provided to this planar transmission line, achieving transmission characteristics surpassing those of traditional microstrip lines. Since it can provide the same characteristic impedance as traditional microstrip lines in the time domain, the characteristic impedance experienced by digital signals propagating through this type of transmission line will not differ significantly from that of traditional microstrip lines, thus achieving the effect of deceiving digital signals. Furthermore, this transmission line has a strong ability to suppress electromagnetic interference and can completely replace traditional microstrip lines in high-speed circuits. Attached Figure Description

[0034] Figure 1 (a) is a schematic diagram of the planar waveguide structure in which the microstrip line of the present invention has periodic edges on both sides;

[0035] Figure 1 (b) is a schematic diagram of the planar waveguide structure in which one side of the microstrip line of the present invention is set as a periodic structure;

[0036] Figure 1 (c) is Figure 1 (b) Equivalent circuit diagram of the planar waveguide structure;

[0037] Figure 2 (a) is a schematic diagram of the capacitance of the double-sided periodic microstrip line as a function of frequency in this invention.

[0038] Figure 2 (b) is a schematic diagram of the capacitance of a single-sided periodic microstrip line as a function of frequency in this invention.

[0039] Figure 2 (c) is a schematic diagram of the capacitance of the double-sided periodic microstrip line as a function of frequency in this invention;

[0040] Figure 2 (d) is a schematic diagram of the capacitance of a single-sided periodic microstrip line as a function of frequency in this invention.

[0041] Figure 3 (a) is a schematic diagram of the characteristic impedance of the double-sided periodic microstrip line in this invention as a function of frequency;

[0042] Figure 3 (b) is a schematic diagram of the characteristic impedance of a single-sided periodic microstrip line as a function of frequency in this invention.

[0043] Figure 3 (c) is a schematic diagram of the S-parameters of the double-sided periodic microstrip line in this invention as a function of frequency;

[0044] Figure 3 (d) is a schematic diagram of the S-parameters of a single-sided periodic microstrip line as a function of frequency in this invention;

[0045] Figure 4 (a) is a schematic diagram of the calculation results of the transient impedance of the double-sided periodic microstrip lines with lattice constants of 8.0 mm, 4.0 mm, 2.5 mm and 2.0 mm in this invention;

[0046] Figure 4 (b) is a schematic diagram of the calculation results of the transient impedance of single-sided periodic microstrip lines with lattice constants of 8.0 mm, 4.0 mm, 2.5 mm and 2.0 mm in this invention;

[0047] Figure 4 (c) is a schematic diagram of the transient impedance calculation results of the double-sided periodic microstrip line in this invention;

[0048] Figure 4 (d) is a schematic diagram of the transient impedance calculation results of a single-sided periodic microstrip line in this invention;

[0049] Figure 5 (a) is a schematic diagram of the measurement results of the transient impedance of double-sided periodic microstrip lines with lattice constants of 8.0 mm, 4.0 mm, 2.5 mm and 2.0 mm in the experiment of this invention;

[0050] Figure 5 (b) is a schematic diagram of the measurement results of the transient impedance of single-sided periodic microstrip lines with lattice constants of 8.0 mm, 4.0 mm, 2.5 mm and 2.0 mm in the experiment of this invention;

[0051] Figure 5 (c) is a schematic diagram of the measurement results of the transient impedance of the double-sided periodic microstrip line in the experiment of this invention;

[0052] Figure 5 (d) is a schematic diagram of the measurement results of the transient impedance of a single-sided periodic microstrip line in the experiment of this invention;

[0053] Figure 6 (a) is a diagram of the coupled circuit structure consisting of a double-sided periodic microstrip line and a conventional microstrip line in the experiment of this invention;

[0054] Figure 6 (b) is a diagram of the coupled circuit structure consisting of a single-sided periodic microstrip line and a conventional microstrip line in the experiment of this invention;

[0055] Figure 6 (c) is a schematic diagram of the remote crosstalk measurement results of the coupling circuit composed of periodic microstrip lines and traditional microstrip lines in the experiment of this invention.

[0056] Figure 6 (d) is a schematic diagram of the near-end crosstalk measurement results of the coupling circuit composed of periodic microstrip lines and traditional microstrip lines in the experiment of this invention. Detailed Implementation

[0057] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0058] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention pertains. The terminology used herein in the description of the invention is for the purpose of describing particular embodiments only and is not intended to limit the invention.

[0059] To overcome the shortcomings of traditional microstrip lines, this embodiment proposes a planar waveguide structure, such as... Figure 1 As shown in (a) and 1(b), the planar waveguide structure includes a microstrip line, a dielectric layer and a metal layer, and the single or double edges of the microstrip line are set as a periodic structure. The microstrip line has a constant characteristic impedance. Therefore, when the microstrip line is used to transmit digital signals, the characteristic impedance of the digital signal detection is the same as that of the uniform microstrip line, which can be used to deceive the digital signal.

[0060] The propagation time of a digital signal transmitted in a microstrip line within a single cell of a periodic structure is less than half the rise time of that digital signal. Therefore, when the digital signal is determined, the lattice constant d of the periodic structure can be obtained as:

[0061]

[0062] In the formula, v is the transmission rate of the digital signal in the microstrip line, and R... T This is the rise time of the digital signal. Setting this lattice constant allows for the determination of R... T A step signal with a rise time cannot detect impedance changes caused by continuously distributed grooves.

[0063] In this embodiment, for ease of manufacturing and research, the unit cell of the periodic structure contains grooves etched from the edges of the microstrip line. That is, the periodic structure in this embodiment is a periodic groove, and when the two sides of the microstrip line are set as periodic structures, the periodic structures of the two sides are symmetrical.

[0064] Traditional microstrip lines can obtain their characteristic impedance by extracting circuit parameters or using the power-to-voltage ratio. For the microstrip line with periodically etched grooves at its edges in this embodiment, this embodiment proposes calculating the characteristic impedance from the perspective of an equivalent circuit. Because the edges of the microstrip line are periodically etched, without considering the resistance of the conductors and leakage current, the entire microstrip line can be calculated using a single equivalent circuit. Figure 1(c) represents the equivalent network that periodically modulates the inductance and capacitance. This circuit that periodically modulates the capacitance and inductance is actually a low-pass filter. By isolating the basic unit of the periodic circuit and using the input impedance method, the characteristic impedance of the periodic network can be obtained. The input impedance Z is viewed from the left side of the equivalent circuit. in It can be represented as:

[0065]

[0066] Then Z in Characteristic impedance It can be represented as:

[0067]

[0068] Among them, Z p It is the capacitance impedance, Z s and Y p These are the inductance and impedance Z. s =jωL and capacitance admittance Y p =jωC, where j is the imaginary unit, ω is the angular frequency, L is the inductance, and C is the capacitance. The resonant frequency of an LC circuit is defined as:

[0069]

[0070] Define Z s With Z p The square root of the product is Z. o :

[0071]

[0072] make:

[0073]

[0074]

[0075] The characteristic impedance can be expressed as:

[0076]

[0077] and

[0078]

[0079] in, and These are the forward and backward Bloch characteristic impedances, respectively. With the capacitance and inductance of the periodic microstrip unit cell, the characteristic impedance can be calculated without loss. The characteristic impedance of a traditional microstrip line differs from that of a periodic microstrip line only by a phase factor. It is worth noting that when ω << ω o (i.e., low-frequency band) or under conditions where the lattice constant is much smaller than the wavelength, there is It can be seen that all the formulas for traditional microstrip lines are still valid in the low-frequency band.

[0080] This embodiment provides an experimental example, based on a circuit model ( Figure 1 The S-parameters calculated for the equivalent circuit (c) will be compared with the full-wave numerical results to examine the validity of the circuit model. Let w be the width of the microstrip line, h be the thickness of the dielectric material (dielectric layer), t be the thickness of the microstrip line and the metal layer, d be the lattice constant, and ε be the dielectric constant. r The depth of the groove is b. In the calculation, it is assumed that the width a of the groove is half of the lattice constant d.

[0081] When extracting the circuit parameters (RLGC) of a microstrip line, this embodiment uses Gauss's law to calculate the total charge Q accumulated per unit length of the microstrip line in order to extract the capacitance of the periodic microstrip line:

[0082]

[0083] in, Let V be the current flux density around the microstrip line, and S be the surface surrounding the microstrip line. Under quasi-static conditions, the potential difference V between the microstrip line and the current return path can be expressed as:

[0084]

[0085] Capacitance per unit length can be expressed as:

[0086]

[0087] The current I flowing in the microstrip line is obtained using Ampere's law:

[0088]

[0089] in, It is an electric field. For the current return path, Where C is the magnetic field strength, and C is the closed path length of the microstrip line. The magnetic flux Φ through a unit length of microstrip line is:

[0090]

[0091] in, S1 is the magnetic flux density, and S1 is the area per unit length between the microstrip line and the ground plane. The self-inductance of the microstrip line can be expressed as:

[0092]

[0093] All circuit parameters were extracted using COMSOL. To verify the theoretical calculation results, an RO4003 circuit board with a dielectric constant of ε was selected. r =3.37, the thickness of the metal film is t = 0.0175 mm, and the thickness of the dielectric material is h = 0.508 mm. To design periodic microstrip lines with arbitrary characteristic impedances, a conventional microstrip line width w0 = 1.04 mm is chosen for comparison, resulting in a characteristic impedance of 53.2 Ω at low frequencies. This choice facilitates analysis when using time-domain signals to probe the transient impedance of the transmission line.

[0094] This experiment considers periodic microstrip lines with three lattice constants: d = 0.5 mm, 1.0 mm, and 2.0 mm. To achieve a characteristic impedance close to 53.2 Ω at low frequencies, the structural parameters for the double-sided periodic microstrip line with a lattice constant of d = 0.5 mm (i.e., a microstrip line with periodic edges) are chosen as w = 1.5392 mm and b = 0.2973 w; for the periodic microstrip line with a lattice constant of d = 1.0 mm, w = 1.5496 mm and b = 0.2986 w; and for the periodic microstrip line with a lattice constant of d = 2.0 mm, w = 1.56 mm and b = 0.3 w. Similarly, the structural parameters for a single-sided periodic microstrip line with a lattice constant of d = 0.5 mm (i.e., a microstrip line with a periodic structure on one side) are w = 1.56 mm and b = 0.6 w; the structural parameters for a single-sided periodic microstrip line with d = 1.0 mm are w = 1.5912 mm and b = 0.6078 w; and the structural parameters for a single-sided periodic microstrip line with d = 2.0 mm are w = 1.5912 mm and b = 0.6013 w. The relevant structural dimensions for periodic microstrip lines are listed in Table 1.

[0095] Table 1. Structural dimensions related to periodic microstrip lines

[0096]

[0097] The calculated capacitance and inductance per unit length of the subwavelength periodic microstrip line selected in this experiment are shown in the figure. Figure 2 , Figure 2(a) and (b) show the capacitance per unit length of BSPML and USPML with periods d = 0.5 mm, 1.0 mm, and 2.0 mm, respectively, as a function of frequency. For high-speed circuit systems, the calculation results of capacitance and inductance per unit length at low frequencies are of primary concern. In the low-frequency range, for example at f = 0.05 GHz, the capacitance per unit length of BSPML with a lattice constant d = 0.5 mm is C = 0.12352 pF / mm, the capacitance per unit length of BSPML with a lattice constant d = 1.0 mm is C = 0.11855 pF / mm, and the capacitance per unit length of conventional microstrip line (CML) is C = 0.10171 pF / mm. As the lattice constant increases, the capacitance per unit length of periodic microstrip lines gradually decreases, while conventional microstrip lines have the smallest capacitance value. Furthermore, these three subwavelength periodic microstrip lines and conventional microstrip lines exhibit similar frequency-dependent capacitance patterns in the low-frequency range. At a frequency of f = 0.05 GHz, the capacitance per unit length of a USPML with a lattice constant of d = 0.5 mm is C = 0.12759 pF / mm, and the capacitance per unit length of a USPML with a lattice constant of d = 1.0 mm is C = 0.12405 pF / mm.

[0098] Figure 2 (c) and (d) show the variation of inductance per unit length of a subwavelength periodic microstrip line with frequency. At a frequency f = 0.05 GHz, the inductance per unit length of a BSPML with a lattice constant d = 0.5 mm is L = 0.34301 nH / mm, and the capacitance per unit length of a BSPML with a lattice constant d = 1.0 mm is L = 0.32897 nH / mm. The inductance per unit length of a conventional microstrip line is L = 0.28793 nH / mm. Introducing a subwavelength periodic structure at the edge of a periodic microstrip line can effectively increase the inductance of the microstrip line. When the microstrip line has a large self-inductance, it has a small mutual inductance with adjacent microstrip lines, thus effectively reducing electromagnetic interference. At a frequency f = 0.05 GHz, the inductance per unit length of a USPML with a lattice constant d = 0.5 mm is L = 0.3547 nH / mm, and the inductance per unit length of a USPML with a lattice constant d = 1.0 mm is L = 0.34621 nH / mm. With the capacitance and inductance per unit length, Z can be obtained from equation (4). o To ensure the circuit model provides accurate S-parameter calculations, the resistance R of the metal conductor in the microstrip line is provided by the perturbation method, and the conductance G is calculated using the following formula:

[0099] G=Cωtanδ (16)

[0100] In the formula, C is the capacitance per unit length of the microstrip line, ω is the angular frequency of the electromagnetic wave, and tanδ is the tangent loss of the material.

[0101] Figure 3 (a) and Figure 3 (b) is Z obtained using circuit parameters. o Variation with frequency. For BSPMLs, at low frequencies (e.g., f = 0.05 GHz), the Z-terminus is d = 0.5 mm, 1.0 mm, 2.0 mm. o The impedances are 52.696Ω, 52.68Ω, and 52.872Ω respectively, compared to the Z-axis of traditional microstrip lines. o The largest difference between 53.207Ω and 0.527Ω is only 0.527Ω. USPMLs, at low frequencies (e.g., f = 0.05GHz), have lattice constants of d = 0.5mm, 1.0mm, and 2.0mm. o The impedances are 52.78Ω, 52.829Ω, and 53.129Ω respectively, compared to the Z-axis of traditional microstrip lines. o The largest difference between Ω and 53.207Ω is only 0.427Ω. It can be seen that by adjusting the geometric parameters of SPMLs with different lattice constants, they can be made to match the Z-axis of a traditional microstrip line with a width of w0 = 1.04 mm. o The numerical difference between them should be controlled within 0.8%. Figure 3 (c) and Figure 3 (d) shows the S-parameter simulation results of BSPML and USPML with lattice constant d = 2.0 mm, respectively. The solid line represents the full-wave calculation result, while the S-parameters calculated by the circuit model are represented by the dashed line. It can be seen that the difference between the two is very small.

[0102] Input a microstrip line with a rise time of R T The voltage step function signal can be used to detect the characteristic impedance of a conventional microstrip line or to detect impedance changes caused by width variations in the microstrip line (i.e., time-domain reflectometry). Therefore, when a microstrip line has grooves, transient impedance changes can also be detected by reflecting the voltage step signal transmitted through the microstrip line. For periodic microstrip lines, since each unit cell has at least one groove, a series of transient impedance peaks will appear along the microstrip line when a step signal passes through a microstrip line with multiple grooves. This embodiment uses a rise time R... T The step signal cannot clearly identify the property of a continuous periodic groove distribution in a microstrip line to define a maximum lattice constant d of a periodic microstrip line that can deceive digital signals. max :

[0103]

[0104] In the formula, v is the propagation rate of the step signal in the microstrip line.

[0105] When the lattice constant d of the periodic microstrip line is less than d max When the lattice constant is too small, the reflected signal within the unit cell will be masked by the rise time delay. Therefore, the reflected wave of the voltage step signal cannot resolve the impedance abrupt change caused by the adjacent grooves in the periodic structure of the microstrip line, and the entire periodic microstrip line will be considered a uniform microstrip line with a certain characteristic impedance. Taking the circuit board used in this embodiment as an example, for the rise time R... T For a step signal of 30 ps, ​​d max The value is 2.5 mm. When the lattice constant d of the periodic microstrip line is less than 2.5 mm, the step signal will not be able to distinguish between the traditional microstrip line and the periodic microstrip line.

[0106] Figure 4 This study aims to obtain the time-varying transient impedance of a periodic microstrip line by utilizing the reflected wave of a time-domain pulse. Figure 4 (a) and 4(b) are periodic grooves with lattice constants of d = 8 mm, 4 mm, 2.5 mm and 2.0 mm, respectively, for traditional microstrip lines with width w0 = 1.04 mm and length 10 cm. The groove depth is 0.3 w for BSPMLs and 0.6 w for USPMLs.

[0107] It can be seen that the periodic microstrip line with a lattice constant of d = 8.0 mm exhibits significant transient impedance oscillations before t = 0.8 ns. As the lattice constant decreases, the transient impedance oscillations rapidly decrease. When the lattice constant is less than d = 2.5 mm, the transient impedance only shows very weak ripples at the beginning. When the lattice is further reduced to d = 2.0 mm, for the double-sided periodic microstrip line, the transient impedance from t = 0.267 ns to 14.0 ns is almost a horizontal line, similar to a traditional microstrip line with a fixed characteristic impedance, whose average characteristic impedance is Z. o =66.31Ω.

[0108] Figure 4 (c) and (d) represent the transient impedance changes over time obtained from the reflected signal after a step signal with a rise time of 30 ps is input to two-sided and one-sided periodic microstrip lines with lattice constants d = 0.5 mm, 1.0 mm, and 2.0 mm, respectively. For a conventional microstrip line with a width w0 = 1.04 mm, the transient impedance varies between 53.0 Ω and 53.26 Ω from time t = 0.292 ns to 1.228 ns. The change in transient impedance within this time interval can almost be considered as a horizontal line, and the midpoint value is taken as the characteristic impedance Z. o = 53.1653Ω. When the transient impedance of a transmission line does not change with time, it can be considered as its characteristic impedance.

[0109] Furthermore, when the loss is very small, the result obtained through time-domain signal reflection techniques can be considered as the characteristic impedance of the transmission line at low frequencies. Figure 3 (a) The characteristic impedance Z of a conventional microstrip line with a width w0 = 1.04 mm at low frequencies o =53.207Ω, with an error of only 0.078% compared to the characteristic impedance obtained from time-domain reflectometry.

[0110] To make the characteristic impedance of a double-sided subwavelength periodic microstrip line with a lattice constant of d = 0.5 mm approach that of a conventional microstrip line with a width of w0 = 1.04 mm, this embodiment selects a linewidth of w = 1.5392 mm and a groove depth of b = 0.4976 mm for the periodic line. The transient impedance obtained by time-domain reflectometry varies between 52.82337 Ω and 53.29868 Ω in the time range of t = 0.292 ns to 1.50386 ns. The midpoint value is taken as the characteristic impedance Z. o =53.1187Ω. From the perspective of time-domain analysis, the characteristic impedance constructed by a double-sided periodic microstrip line with a lattice constant of d = 0.5 mm differs from that of a conventional microstrip line by only 0.08%.

[0111] Aside from the time delay caused by the periodic microstrip line, the characteristic impedance experienced by the digital signal is almost the same when propagating on these two types of microstrip lines. For a single-sided subwavelength periodic microstrip line with a lattice constant of d = 0.5 mm, if the width of the microstrip line is chosen to be w = 1.56 mm and the groove depth is b = 0.936 mm, the characteristic impedance Z obtained using time-domain reflectometry is... o =52.955Ω. That is, this experiment proves that under the condition of satisfying equation (1), by selecting a periodic microstrip line with an appropriate width, a periodic microstrip line with a characteristic impedance that is very close to that of a traditional microstrip line can be obtained.

[0112] To verify the accuracy of equation (1), a microstrip line with a width of w0 = 1.04 mm and a length of 10 cm was first selected to design single-sided and double-sided periodic microstrip lines with lattice constants of d = 8 mm, 4 mm, 3 mm, 2.5 mm and 2 mm, respectively. The groove depth of the double-sided groove periodic structure was b = 0.3 w0, while the groove depth of the single-sided periodic structure was b = 0.6 w0. A voltage step signal with a rise time of 30 ps was input into the periodic microstrip line to detect the characteristic impedance of the periodic microstrip line.

[0113] Figure 5 (a) and (b) show the measurement results of the transient impedance of a two-sided and a single-sided periodic microstrip line with a width of w0 = 1.04 mm, respectively. The data obtained from theoretical calculations are in high agreement with the experimental results. Figure 5(c) and (d) show the transient impedance measurements performed using a time-domain reflectometer on the double-sided and single-sided grooved periodic microstrip lines, respectively, after selecting the dimensions in Table 1. The measured characteristic impedance value for a conventional microstrip line with a width w0 = 1.04 mm is Z. o =51.99Ω, lattice constant d = 2.0mm, and characteristic impedance Z of a double-sided periodic microstrip line with a width w = 1.56mm. o =51.749Ω, lattice constant d = 1.0mm, and characteristic impedance Z of a single-sided periodic microstrip line with a width w = 1.5912mm. o =51.7319Ω. It can be seen that when a periodic microstrip line with a suitable width is selected and the standard of equation (1) is met, the result obtained from the time-domain reflection measurement shows that its transient impedance is almost a horizontal line and is very close to the transient impedance of a traditional microstrip line.

[0114] In addition, to verify the ability of the periodic microstrip line to suppress both long-range and near-end crosstalk, this experiment measured long-range crosstalk. A voltage step signal with a rise time of 30 ps and an amplitude of 0.2 V was input to one port of the periodic microstrip line. The circuit for measuring the time-domain signal is as follows. Figure 6 As shown in (a) and (b). Figure 6 (a) is a coupling circuit composed of a double-sided periodic microstrip line and a conventional microstrip line. Figure 6 (b) is a coupling circuit composed of a single-sided periodic microstrip line and a conventional microstrip line. Figure 6 (c) is the result of remote crosstalk measurement of the coupled microstrip circuit.

[0115] The coupling region of the two parallel microstrip lines is 10 cm, and the spacing between the two microstrip lines is 1.04 mm (i.e., the linewidth of a conventional microstrip line). It can be seen that at time t = 0.8687 ns, the long-range crosstalk (FEXT) of the two conventional microstrip lines is -0.06866 V, and the crosstalk accounts for approximately 34.33% of the incident signal amplitude. If one conventional microstrip line is replaced with a double-sided periodic microstrip line with a lattice constant of 0.5 mm, the FEXT becomes -0.02118 V, and the crosstalk accounts for approximately 10.59% of the incident signal amplitude. If one conventional microstrip line is replaced with a single-sided periodic microstrip line with a lattice constant of 0.5 mm, the FEXT becomes -0.008 V, and the crosstalk accounts for approximately 4% of the incident signal amplitude. This periodic etching structure at the edges of the microstrip lines can effectively reduce electromagnetic interference from the original 34.33% to 4%, and the long-range crosstalk is reduced to about one-tenth of its original value. Figure 6(d) shows the measurement results of near-end crosstalk. For two parallel conventional microstrip lines, the measurement result at t = 1.0 ns is 0.00617 V. If one of the parallel conventional microstrip lines is replaced with a double-sided periodic microstrip line with a period of d = 0.5 mm, the measurement result of NEXT is 0.00475 V. If it is replaced with a single-sided periodic microstrip line with a period of d = 0.5 mm, the measured result of NEXT is 0.00375 V.

[0116] This experiment demonstrates that the planar waveguide structure provided in this embodiment, by adjusting the microstrip line width, groove depth, and lattice constant, can achieve a characteristic impedance similar to that of traditional microstrip lines (between 52.7Ω and 53.2Ω), making the periodic grooves imperceptible to high-speed digital signals. Furthermore, the experiment shows that the planar waveguide structure of this embodiment can reduce crosstalk with adjacent transmission microstrip lines, thus enabling it to completely replace traditional microstrip lines in high-speed circuit systems. Moreover, this embodiment provides a definitive formula for the characteristic impedance of a lossless periodic transmission line based on microwave network technology, representing a valuable advancement for the complete replacement of traditional microstrip lines with periodic microstrip lines.

[0117] The technical features of the above embodiments can be combined in any way. For the sake of brevity, not all possible combinations of the technical features in the above embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0118] The embodiments described above are merely illustrative of several implementations of the present invention, and while the descriptions are specific and detailed, they should not be construed as limiting the scope of the invention. It should be noted that those skilled in the art can make various modifications and improvements without departing from the concept of the present invention, and these modifications and improvements all fall within the scope of protection of the present invention. Therefore, the scope of protection of the present invention should be determined by the appended claims.

Claims

1. A planar waveguide structure comprising a microstrip line, a dielectric layer and a metal layer, the microstrip line being arranged to transmit a digital signal, characterized in that, The single-side or double-side edge of the microstrip line is set as a periodic structure, and the microstrip line has a constant transient impedance, and a lattice constant d of the periodic structure is: where v is the transmission rate of the digital signal in the microstrip line, R T is the rise time of the digital signal, and R T ≤ 30 ps.

2. The planar waveguide structure of claim 1, wherein, The microstrip line is equivalent to an equivalent network periodically modulated by inductance and capacitance, and a characteristic impedance of the equivalent network is obtained based on a unit cell of the periodic structure by using an input impedance method, as follows: Input impedance Z of a unit cell of a periodic structure in is represented as: The input impedance Z in characteristic impedance may be expressed as: where Z p is the capacitive impedance, Z s and Y p are the inductive impedance Z s = jcoL and the capacitive admittance Y p = jcoC, respectively, j is the imaginary unit, co is the angular frequency, L is the inductance, and C is the capacitance.

3. The planar waveguide structure of claim 2, wherein, The resonant frequency of the LC circuit is defined as ω o : Definition of Z s Z is multiplied by p Z is the quantity that is squared o : Let: The characteristic impedance expression is: and wherein Zf and Zr are the forward and backward Bloch characteristic impedances, respectively, are the forward and backward Bloch characteristic impedances, respectively, and are only a phase factor apart from Z, respectively.

4. The planar waveguide structure of claim 1, wherein, The unit cell of the periodic structure contains a groove etched by the edge of the microstrip line.

5. The planar waveguide structure of claim 4, wherein, The width of the groove is half of the lattice constant.

6. The planar waveguide structure of claim 4, wherein, The unit cell has at least one groove.

7. The planar waveguide structure of claim 1, wherein, When the double-side edge of the microstrip line is set as a periodic structure, the periodic structure of the double-side edge has symmetry.

8. The planar waveguide structure of claim 1, wherein, When the double-side edge of the microstrip line is set as a periodic structure, the periodic structure of the double-side edge has asymmetry.

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