Dual Active Bridge Converter with Bidirectional Differential Power Transfer and Control Method

Abstract

The present invention provides a dual-active-bridge converter and a control method with bidirectional differential power transfer, including an input side, a transformer, and an output side; the input side includes six switching tubes S1 - S6, and the six switching tubes S1 - S6 are connected in series in pairs to form three groups and are connected in parallel on the power supply side; the output side includes a symmetrical full bridge, and the symmetrical full bridge includes two full bridges with the same structure. The full bridge includes two diodes connected in series and two switching tubes connected in series, and the two diodes connected in series and the two switching tubes connected in series are connected in parallel; the transformer includes a first transformer and a second transformer. The first group of series-connected switching tubes S1, S2 and the second group of series-connected switching tubes S3, S4 are connected to a full bridge through the first transformer, and the second group of series-connected switching tubes S3, S4 and the third group of series-connected switching tubes S5, S6 are connected to another full bridge through the second transformer. The present invention has the characteristics of high efficiency, bidirectional differential power flow, etc., and realizes the two-quadrant operation of the converter with a relatively low switching device capacity.

Description

Technical Field

[0001] The present invention relates to the field of electronic circuits, and in particular to a dual active bridge converter topology with bidirectional differentiated power transfer capability. Background Art

[0002] According to the 2022 annual report of the Global Wind Energy Council, the total installed capacity of global offshore wind power in 2021 was 21.1GW, an increase of 210% compared with 6.8GW in 2020. Among them, my country's total installed capacity increased from 3.8GW to 16.9GW. According to the forecast of the GWEC organization, the number of global offshore wind power installed capacity will continue to increase every year, and the total annual installed capacity will reach 54.8GW by 2030. With the gradual improvement of the development and utilization of offshore wind energy resources, offshore resources have also received more and more attention.

[0003] Common technologies for offshore wind power transmission are divided into two categories: high voltage AC (HVAC) and high voltage DC (HVDC). For offshore wind power that is more than 70 to 100 kilometers offshore, HVAC will cause many problems: power loss increases sharply with the length of the transmission line, and eventually exceeds the HVDC system; the submarine cable-to-ground capacitance generates a large amount of reactive power, which is difficult to compensate; compared with the HVDC system, additional safety factors such as harmonics, frequency deviation, and three-phase imbalance need to be considered, and it is difficult to achieve fault ride-through, which seriously affects system reliability. Therefore, for offshore wind power, the HVDC system has become the first choice.

[0004] The existing traditional HVDC usually uses DC-AC inverters to boost the voltage, collects it to the high-voltage AC bus, and then rectifies it for output by AC-DC rectifiers. However, the area available for offshore wind power is very limited, and these converters are large and difficult to transport to offshore substations for installation, so they cannot be used for offshore wind power. In order to overcome this problem, it is necessary to use a pure DC transmission system and introduce DC-DC converters with high power density, high efficiency, and high step-up ratio.

[0005] For a dual active bridge (DAB) DC-DC converter that uses a high-frequency transformer as the core device, by changing the external phase shift angle and internal phase shift angle between the primary and secondary full bridges, bidirectional power transfer of the converter can be achieved while ensuring zero-voltage turn-on of the switching tubes. With the above advantages, DAB has been widely used in HVDC. However, for the special application scenario of offshore wind power, the functions of DAB cannot be fully utilized, that is, under normal operating conditions, the power flows unidirectionally, from the offshore wind farm to the onshore power grid; reverse power transfer only exists in a few working conditions such as the restart of shutdown fans and the startup of auxiliary equipment, and the startup power only accounts for less than 5% of the converter capacity. The redundancy of the reverse power transfer capacity is relatively large, bringing unnecessary costs to the construction and maintenance of the coastal converter station.

[0006] Literature: F. Li, Y. Li and X. You, "Optimal Dual-Phase-Shift Control Strategy of an Isolated Buck–Boost Converter With a Clamped Inductor," in IEEE Transactions on Power Electronics, vol. 33, no. 6, pp. 5374-5385, June 2018, doi: 10.1109 / TPEL.2017.2732439. discloses an isolated buck-boost topology applying SAB, which reduces the requirement for the number of switching tubes to a certain extent and can meet the case of unidirectional power transfer, but cannot achieve bidirectional power transfer. Summary of the Invention

[0007] Aiming at the defects in the prior art, the purpose of the present invention is to provide a dual active bridge converter and control method for bidirectional differential power transfer.

[0008] A dual active bridge converter for bidirectional differential power transfer according to the present invention includes an input side, a transformer and an output side;

[0009] The input side includes six switching tubes S 1 -S 6 , and six switching tubes S 1 -S 6 are connected in series in pairs to form three groups and are connected in parallel on the power supply side;

[0010] The output side includes a symmetric full bridge, and the symmetric full bridge includes two full bridges with the same structure. The full bridge includes two diodes connected in series and two switching tubes connected in series, and the two diodes connected in series and the two switching tubes connected in series are connected in parallel;

[0011] The transformer includes a first transformer and a second transformer. A first group of series-connected switching tubes S 1 , S 2 and a second group of series-connected switching tubes S 3 , S 4 are connected to a full bridge through the first transformer. A second group of series-connected switching tubes S 3 , S 4 and a third group of series-connected switching tubes S 5 , S 6 are connected to another full bridge through the second transformer.

[0012] Preferably, one end of the primary winding of the first transformer is connected between two series-connected switching tubes S 1 , S 2 of the first group, and the other end is connected between two series-connected switching tubes S 3 , S 4 of the second group;

[0013] One end of the primary winding of the second transformer is connected between two series-connected switching tubes S 3 , S 4 of the second group, and the other end is connected between two series-connected switching tubes S 5 , S 6 of the third group.

[0014] Preferably, one end of the secondary winding of the first transformer is connected between two series-connected switching tubes Q 1 , Q 2 of a full bridge, and the other end is connected between two series-connected diodes D 1 , D 2 of the full bridge;

[0015] One end of the secondary winding of the second transformer is connected between two series-connected switching tubes Q 3 , Q 4 of another full bridge, and the other end is connected between two series-connected diodes D 3 , D 4 of the other full bridge.

[0016] Preferably, the other end of the secondary winding of the first transformer is connected to the other end of the secondary winding of the second transformer through a capacitor.

[0017] Preferably, the full bridge further includes a capacitor, and the capacitor is connected in parallel with the two series-connected diodes and the two series-connected switching tubes.

[0018] Preferably, the capacitors of the two full bridges are connected in series.

[0019] Preferably, it further includes a PI regulator for regulating each of the switching tubes S 1 -S 6 and the phase shift angle of the switching tube Q 1 -Q 4 .

[0020] According to a control method of a bidirectional differential power transfer dual active bridge converter provided by the present invention, the voltage of the series-connected capacitors is maintained constant through a voltage equalization control strategy.

[0021] Preferably, forward power transfer is achieved through the control method. The voltage equalization control strategy introduces the duty cycle difference of the switching tubes of two full bridges as a control variable, and forward power transfer is achieved by adjusting the phase shift angle.

[0022] Preferably, reverse power transfer is achieved through the control method. By adjusting the phase shift angle of each of the switching tubes S 1 -S 6 and the switching tube Q 1 -Q 4 , reverse power transfer is achieved.

[0023] Compared with the prior art, the present invention has the following beneficial effects:

[0024] The present invention has the ability of bidirectional differential power transfer from the wind farm side to the grid side and from the grid side to the wind farm side.

[0025] The present invention replaces half of the switching tubes with diodes, reducing costs and the volume of the converter station.

[0026] Based on the converter modulation strategy, the present invention can achieve forward and reverse power regulation of the proposed topology.

[0027] By selecting appropriate parameters, the present invention can enable the switching devices of the converter to operate within the soft switching region under normal far-sea offshore wind power transmission conditions, reducing the power loss and heat dissipation requirements of the converter.

[0028] The present invention has characteristics such as high efficiency and bidirectional power differential flow. It realizes the two-quadrant operation of the converter while making full use of the switching devices. BRIEF DESCRIPTION OF THE DRAWINGS

[0029] By reading the detailed description of the non-limiting embodiments with reference to the following drawings, other features, objects, and advantages of the present invention will become more apparent:

[0030] Figure 1 is the converter topology diagram;

[0031] Figure 2 is the key waveform diagram of the forward mode;

[0032] Figure 3 It is the key waveform diagram of the reverse mode;

[0033] Figures 4a to 4f It is the current flow diagram of different stages in the forward mode;

[0034] Figures 5a to 5f It is the current flow diagram of different stages in the reverse mode;

[0035] Figure 6 It is the soft-switching operating range diagram in the forward mode;

[0036] Figure 7 It is the soft-switching operating range diagram in the reverse mode;

[0037] Figure 8a 、 Figure 8b It is the curve of the average value of leakage inductance current - phase shift angle relationship and the soft-switching critical point diagram with different k values in the forward mode;

[0038] Figure 9a 、 Figure 9b It is the curve of the average value of leakage inductance current - phase shift angle relationship and the soft-switching critical point diagram with different k values in the reverse mode;

[0039] Figure 10 It is the control block diagram in the forward mode and the schematic diagram of the duty cycle and phase shift angle generation of the switch tube pulse signal;

[0040] Figure 11 It is the control block diagram in the reverse mode and the schematic diagram of the duty cycle and phase shift angle generation of the switch tube pulse signal;

[0041] Figure 12 It is the simulation waveform diagram in the forward mode;

[0042] Figure 13 It is the simulation waveform diagram in the reverse mode;

[0043] Figure 14 It is the soft-switching diagram of the switch tube in the forward mode;

[0044] Figure 15 It is the soft-switching diagram of the switch tube in the reverse mode;

[0045] Figure 16 It is the waveform diagram of the output-side series capacitor voltage balance. Specific implementation manners

[0046] The present invention will be described in detail below with reference to specific embodiments. The following embodiments will help those skilled in the art to further understand the present invention, but do not limit the present invention in any form. It should be noted that for those of ordinary skill in the art, without departing from the concept of the present invention, several changes and improvements can still be made. These all belong to the protection scope of the present invention.

[0047] A dual-active-bridge converter topology with bidirectional differential power transfer provided by the present invention is as follows Figure 1 shown, including an input side, a transformer, and an output side.

[0048] The input side includes six switching tubes S: S 1 , S 2 , S 3 , S 4 , S 5 , S 6 . The six switching tubes are connected in series in pairs to form three groups and are connected in parallel to the power supply side. The output side includes a symmetric full-bridge, and the symmetric full-bridge includes two full-bridges with the same structure. The full-bridge includes two series-connected diodes and two series-connected switching tubes Q. The two series-connected diodes and the two series-connected switching tubes are connected in parallel, and are respectively denoted as D 1 , D 2 , D 3 , D 4 and Q 1 , Q 2 , Q 3 , Q 4 . The input side is respectively connected to the full-bridge on the output side through two high-frequency transformers with a turns ratio of 1:k, where the connection method of the same-name ends of the upper and lower side windings is opposite. The transformer includes a first transformer and a second transformer. The first group of series-connected switching tubes S and the second group of series-connected switching tubes S are connected to a full-bridge through the first transformer, and the second group of series-connected switching tubes S and the third group of series-connected switching tubes S are connected to another full-bridge through the second transformer.

[0049] Specifically, one end of the primary winding of the first transformer is connected between the two series-connected switching tubes S 1 , S 2 of the first group, and the other end is connected between the two series-connected switching tubes S 3 , S 4 of the second group; one end of the primary winding of the second transformer is connected between the two series-connected switching tubes S 3 , S 4 of the second group, and the other end is connected between the two series-connected switching tubes S 5 , S 6 of the third group. One end of the secondary winding of the first transformer is connected between the two series-connected switching tubes Q 1 , Q 2 of a full-bridge, and the other end is connected between the two series-connected diodes D 1 , D 2 of the said full-bridge; one end of the secondary winding of the second transformer is connected between the two series-connected switching tubes Q 3 , Q 4 of another full-bridge, and the other end is connected between the two series-connected diodes D of the said another full-bridge3 , D 4 Between

[0050] The full bridge further includes a capacitor, which is connected in parallel with two series-connected diodes and two series-connected switching transistors, and the capacitors of the two full bridges are connected in series. The other end of the secondary winding of the first transformer is connected to the other end of the secondary winding of the second transformer through a capacitor, which plays a role in blocking direct current during reverse power flow.

[0051] Basic working principle

[0052] Forward mode:

[0053] When the topology is in forward power flow, among the three primary arms, the working states of arm A and arm C are exactly the same, while the working state of arm B is opposite to that of the above-mentioned arms, forming a power flow path on the primary side. The two high-frequency transformers operate in parallel, and the two half-active full bridges on the secondary side maintain the same working state. At this time, the trigger signals of the two switching transistors of the secondary full bridge need to be kept symmetrical. The specific gate trigger signal setting and device input conditions Figure 2 As shown.

[0054] Stage 1: t 0 -t 1 Time period

[0055] Taking the arrival time of the pulse signals of S 1 , S 4 , S 5 as the starting point of the steady-state working cycle. Before the signal arrives, Q 2 , Q 4 are in the conducting state. At this time, the primary current flows through the body diodes of S 1 , S 4 , S 5 . The secondary current flows through the body diodes of Q 2 and D 1 . At this time, the voltage applied to the primary leakage inductance of the transformer is V i +V o / 2n, and the current rises rapidly from a negative value until the current becomes 0, and the current flow direction changes. The expression for the rate of change of the leakage inductance current i 1 is:

[0056]

[0057] Stage 2: t 1 -t 2 Time period

[0058] When the current flow direction is positive, since the turn-on signal has been applied to Q 2 before, the primary current flows from S 1 , S 4 , S5 The body diode of [component] is commutated to the channel. For the secondary side, the current flows through Q 2 , via D 2 for freewheeling. The secondary side of the transformer is in a short - circuit state, and the voltage applied across the leakage inductance of the primary side of the transformer is V i , and the rising speed of the current value slows down. The expression for the rate of change of the leakage inductance current i 1 is:

[0059]

[0060] Stage 3: t 2 - t 3 time period

[0061] When Q 2 turns off and Q 1 turns on, the current flow direction is positive. At this time, it can only freewheel through the body diode of Q 1 . The current flows through the power grid and returns to the transformer via D 2 . The voltage applied across the inductor is V i - V o / 2n, and the current remains constant. The expression for the rate of change of the leakage inductance current i 1 is:

[0062]

[0063] Stage 4: t 3 - t 4 time period

[0064] When S 1 , S 4 , S 5 turn off, the current switches from S 1 , S 4 , S 5 to the body diode of S 2 , S 3 , S 6 . A negative voltage is applied to the primary side of the transformer from the input side, and the voltage across L 1 is - V i - V o / 2n, and the current value rapidly drops to 0. The expression for the rate of change of the leakage inductance current i 1 is:

[0065]

[0066] Stage 5: t 4 - t 5 time period

[0067] When the current reverses, the secondary side current then flows from Q 1 , Q 3The body diode commutates to the channel, and the current flows through D 1 , the secondary side is approximately short - circuited, and the voltage applied to the inductor is -V i , the current decrease rate slows down, and this process continues until Q 1 , Q 3 turns off. The expression for the rate of change of the leakage inductance current i 1 is:

[0068]

[0069] Phase 6: t 5 -t 6 time period

[0070] When Q 1 , Q 3 turns off, Q 2 , Q 4 turns on, the secondary - side current flows through the body diode of Q 2 and D 1 , the voltage applied to the inductor remains unchanged, and the current remains constant. When S 2 , S 3 , S 6 turns off, S 1 , S 4 , S 5 turns on, the process returns to Phase 1. The expression for the rate of change of the leakage inductance current i 1 is:

[0071]

[0072] The current flow directions in different time periods are as Figures 4a to 4f shown.

[0073] Taking the reference value of the rate of change of the inductor current as V i / L 1 , the per - unit values of the rate of change of the current in different phases are as follows in the table:

[0074] Stage 1 Stage 2 Stage 3 Stage 4 Stage 5 Stage 6 k+1 1 1-k -1-k -1 k-1

[0075] Under the forward - power - transfer operating condition, assuming that the circuit is at the steady - state operating point and satisfies the volt - second balance law. At the same time, define the phase - shift angle η at this time as the angle by which S 1 leads Q 1 . Let the duration of Mode 1 be t a , and the duration of Mode 2 be t b . Defining the current stress as the maximum value of the inductor current, the current stress under different transformer turns - ratio values is shown in the following formula:

[0076]

[0077] Under steady-state operation, the absolute values of the currents at times t 0 and t 3 should be equal. At the same time, from the working mode, it can be known that the sum of the times t a and t b should be equal to the duration corresponding to the phase-shift angle:

[0078]

[0079] Solving the above equations can obtain:

[0080]

[0081] Under the per-unit system, the average value of the wind farm side current in half a period is taken, that is, the period from t 0 to t 3 has three stages in total, and the calculation formula is as follows:

[0082]

[0083] Simplifying the above formula can obtain:

[0084]

[0085] It should be noted that the above formula is not applicable in the case of extremely light load or severe mismatch, that is, due to the selection of the phase-shift angle and the turns ratio, stage 2 does not exist. And such situations rarely occur in the application scenario of offshore wind power transmission, so this patent does not discuss it.

[0086] In the forward power transmission mode, the power transfer expression from the primary side to the secondary side is as follows:

[0087]

[0088] Reverse mode

[0089] When the power flows in the reverse direction, the pulses of the two switching tubes of the primary side bridge arm B are blocked, that is, this bridge arm exits the working state, and the bridge arms A and C conduct diagonally. No current can flow through the bridge arm B. Therefore, the two transformers are in series at this time. Since half of the devices in the secondary full-bridge are diodes with unidirectional conductivity, only four switching tubes in the secondary side are in the working state at this time. To meet the current flow path requirements during reverse power transmission, and the wiring methods of the upper and lower half-active bridges are basically the same, but the transformer connections are opposite, so the drive signals of the two switching tubes of the lower full-bridge need to be swapped. In the reverse direction, the working mode of the right-side topology is a three-level half-bridge. The specific gate trigger signal setting and device usage are as Figure 3 shown.

[0090] Stage 1: The period from t 0 to t 1 ​

[0091] With S 1 , S 6 The moment when the trigger signal arrives is taken as the starting point of the steady-state working cycle. At this time, the inductor current is negative, and S 1 , S 6 , Q 1 , Q 4 are all in the conducting state, and the primary current flows through the body diode of S 1 , S 6 , and the secondary current flows through the body diode of Q 1 , Q 4 The voltage applied across the leakage inductance is V i / 2 - V o / 4n, and the expression for the rate of change of the leakage inductance current i 1 is:

[0092]

[0093] Stage 2: t 1 -t 2 time period

[0094] When Q 1 , Q 4 turns off and Q 2 , Q 3 turns on, since the inductor current remains negative, the secondary current commutates from the channel of Q 1 , Q 4 to the body diode of Q 2 , Q 3 . The flow path of the primary current remains unchanged, and the voltage applied across the leakage inductance becomes V i / 2 + V o / 4n (DC-blocking capacitor voltage), and the expression for the rate of change of the leakage inductance current i 1 is:

[0095]

[0096] Stage 3: t 2 -t 3 time period

[0097] When the current rises to 0, the direction of current flow changes to the positive direction. The primary current commutates from the body diode of S 1 , S 6 to the channel, and the secondary current commutates from the body diode of Q 2 , Q 3 to the channel, and other states of the circuit remain unchanged.

[0098] Stage 4: t 3 -t 4 time period

[0099] When S 1 , S 6 is turned off, and S 2 , S 5 is turned on, the primary current commutates from the channel of S 1 , S 6 to the body diode of S 2 , S 5 . The leakage inductance voltage becomes -V i / 2 + V o / 4n, and the expression for the rate of change of the leakage inductance current i 1 is:

[0100]

[0101] Stage 5: t 4 -t 5 time period

[0102] When Q 2 , Q 3 is turned off, and Q 1 , Q 4 is turned on, since the inductor current remains positive, the secondary current commutates from the channel of Q 2 , Q 3 to the body diode of Q 1 , Q 4 . The primary current conduction path remains unchanged, and the voltage applied across the leakage inductance becomes -V i / 2 - V o / 4n (DC-blocking capacitor voltage). The expression for the rate of change of the leakage inductance current i 1 is:

[0103]

[0104] Stage 6: t 5 -t 6 time period

[0105] When the inductor current becomes 0, at this time the current direction reverses. The primary current commutates from the body diode of S 2 , S 5 to the channel, and the secondary current commutates from the body diode of Q 1 , Q 4 to the channel until the current reaches the negative peak value, and then returns to Stage 1 again.

[0106] The current flow directions in different time periods are as Figures 5a to 5f shown.

[0107] Define the phase shift angle η at this time as the angle by which Q 1 leads S 1 . Similarly, in the case of reverse power flow, the following results can also be obtained. The reverse current stress takes half of the peak-to-peak value of the current:

[0108]

[0109] Write the following formula according to the principle of equal absolute value of current:

[0110]

[0111] The time of different stages during reverse power transmission is solved as follows:

[0112]

[0113] Since the above calculations are carried out under the condition of being reduced to the primary side, the calculation stage of the average current during reverse power transmission should be selected as t 2 -t 5 time period.

[0114]

[0115] Simplifying the above formula, the average primary current during reverse power transmission is as follows

[0116]

[0117] The transmission power expression is as follows:

[0118]

[0119] Differential power flow

[0120] To verify the forward and reverse power transmission capacities of the above topology, let the transformer turns ratio k be 1, and we can get:

[0121]

[0122] Analyzing the above formula, it can be seen that the power reaches the maximum value at a phase shift angle of 0.3 during forward power transmission, while it reaches the maximum value at 0.25 during reverse power transmission. The peak power ratio is as follows: The capacity peak of forward power transmission is more than three times that of the reverse, and the requirements of differential power transmission in both forward and reverse directions can be better achieved.

[0123] P pos_max :P neg_max =16:5 (24)

[0124] Under the condition of the same maximum forward transmission power, if the switching device types are the same, by analyzing the ratio of the number of switching tubes required for this topology and the traditional DAB topology when the capacity needs to be the same, it can be obtained that this topology can replace 25% of the switching tubes with diodes, reducing the number of switching devices and significantly improving the economy of far-sea wind power transmission.

[0125] Analysis of Soft-Switching Characteristics

[0126] Since there are significant differences in the circuit topologies for the forward and reverse directions, the switching characteristics need to be analyzed separately when discussing them.

[0127] When the transformer turns ratio is 1, all the switching devices on the primary side of the transformer and the switching devices in SemiAB on the secondary side are in the ZVS turn-on state. However, when the transformer turns ratio is not 1 and the load is light, some of the switching devices may lose the zero-voltage turn-on characteristic. From the appearance of the current flow path, ZVS turn-on means that before the turn-on signal of the switching device arrives, there must be current flowing through the body diode so that ZVS conduction can be achieved. If the commutation directly occurs from the channel of the complementary switching devices in the same bridge arm to its own channel, a large amount of switching losses will be generated.

[0128] The conditions for maintaining ZVS conduction of all devices when the topology is in the forward operating mode are as follows:

[0129]

[0130] For k < 1, since the per-unit value of the current change rate in stage 2 is 1, that is, t b > 0 can meet the ZVS condition. For k > 1, as long as stage 1 exists, the primary side ZVS can be guaranteed. The existence of stage 1 means that t a > 0. Substituting the above formula and solving, the soft-switching region in the forward power transfer mode is as follows:

[0131]

[0132] According to the above average current expression of the primary side, the curve is plotted as Figure 6 shown. Different values of k are taken respectively, and the curve of the output power varying with the phase-shift angle is observed and the soft-switching critical operating points are marked as Figure 8a and 8b shown.

[0133] By analogy with the forward power transfer mode, the analysis in the reverse power transfer mode shows that the conditions for the switching devices to be in the ZVS turn-on state are as follows:

[0134]

[0135] The above formula can be equivalently expressed as t b > 0 for k < 1 and t a > 0 for k > 1. The soft-switching region is as follows:

[0136]

[0137] Based on the average current expression of the primary side during reverse power transfer, the soft-switching region can be plotted as Figure 7As shown, different values are taken for k respectively, and the curve of the output power varying with the phase-shifting angle is observed and the soft-switching critical operating points are marked as Figure 9a and 9b shown.

[0138] Control strategy

[0139] Since a capacitor is connected in parallel on the grid side of the secondary S-DAB, considering the possible asymmetry characteristics of the main circuit and the drive circuit, the voltages of the two series voltage-dividing capacitors will be different, resulting in a large voltage stress on both sides of the switching tube. Therefore, a voltage-sharing control strategy needs to be introduced to maintain the capacitor voltage constant.

[0140] Since the above derivation is based on Q 1 and Q 4 having exactly the same trigger pulses, Q 2 and Q 3 having exactly the same trigger pulses, that is, it is defaulted that there are only the following two conduction modes for the secondary switching tubes:

[0141] <![CDATA[Q 1 > <![CDATA[Q 2 > <![CDATA[Q 3 > <![CDATA[Q 4 > Mode 1 Turn on Turn off Turn off Turn on Mode 2 Turn off Turn on Turn on Turn off

[0142] Under the condition of fixed duty cycle and phase-shifting angle, the system lacks adjustment freedom. Therefore, the duty cycle difference between the secondary Q 1 and Q 4 is introduced as a control variable in the voltage-sharing control strategy to stabilize the voltages of the DC-blocking capacitor and the series voltage-dividing capacitors. Q 2 and Q 3 both take the complementary signals of the trigger signals of the other switching tube in the corresponding bridge arm.

[0143] First, analyze the principle of voltage sharing of the secondary series capacitors. In the above cases of Mode 1 and Mode 2, C 1 and C 2 are put into use simultaneously. When the sum of the voltages is the same as the DC-side voltage, the voltage-dividing condition basically remains unchanged. Therefore, a new working mode needs to be introduced to adjust the output capacitor voltage:

[0144] <![CDATA[Q 1 > <![CDATA[Q 2 > <![CDATA[Q 3 > <![CDATA[Q 4 > Mode 3 Turn on Turn off Turn on Turn off Mode 4 Turn off Turn on Turn off Turn on

[0145] Keeping the duty cycles of Q 1 and Q 4 the same and adjusting the duty cycle will result in the conduction conditions of the switching tubes in two new modes.

[0146] Analysis shows that either Mode 3 or Mode 4 can adjust the voltage mismatch condition of the series voltage-dividing capacitors to balance the capacitor voltages. Use a PI regulator to adjust Q 1 and Q 4The phase-shifting angle. In the balanced state, the phase-shifting angle between the two is 0. Mode 3 can be applied to the case where the voltage of the upper series capacitor is higher than that of the lower series capacitor, and Mode 4 can be applied to the case where the voltage of the lower series capacitor is higher than that of the upper series capacitor.

[0147] Moreover, in the forward power transmission mode, the wind farm exhibits a current source characteristic externally, and a closed-loop control also needs to be introduced to ensure the stability of the voltage on the wind farm side. When the output power of the wind farm side decreases, the phase-shifting angle between S 1 and Q 1 is decreased. When the output power of the wind farm side increases, the phase-shifting angle between S 1 and Q 1 is increased. On the basis of achieving power balance, the capacitor voltage on the wind farm side, that is, the input voltage, can be ensured to be stable at the set value.

[0148] For this topology, in the forward and reverse operating modes, the control block diagrams shown in Figure 10 and Figure 11 can be adopted to achieve capacitor voltage balance:

[0149] Implementation example:

[0150] According to the Figure 1 topology, a Plecs simulation model is built. The input voltage V i = 1500V, V o = 3000V, the switching frequency is 10kHz, the dead time t d = 1μs, the transformer turns ratio k = 1, the DC-blocking capacitor C o1 = 330μF, C 1 = C 2 = 330μF, L 1 = 120μH.

[0151] 1. Forward-mode operation simulation. Set the duty cycle of each pulse signal to 0.5, and the phase-shifting angle between the primary and secondary sides to 0.3. At steady state, the average value of the primary-side wind farm side current is 127.9A, and the average value of the secondary-side grid side current is 63.51A. Since the transformer turns ratio is 1, it can be concluded that the forward transmission power is the largest at this time, and the average value is 191.85kW. The waveforms of the trigger signals of each switch tube, the voltages on both sides of the leakage inductance, and the leakage inductance current are as shown in Figure 12 . Taking S 1 as an example to observe the soft-switching situation of the switch tube, it can be found from Figure 14 that ZVS conduction is achieved.

[0152] 2. Reverse-mode operation simulation. By adjusting the pulse signal given to each switch tube with a phase shift angle of 0.25, it can be found that the average value of the primary side wind field current is -39.7 A, the average value of the secondary side grid current is -20.15 A, and the reverse transmission power is the largest, with an average value of 59.55 kW. The input side and output side currents are as Figure 13 shown. Taking S 1 as an example to observe the soft-switching situation of the switch tube, it can be found from Figure 15 that ZVS conduction is achieved.

[0153] 3. Considering the possible asymmetry of the circuit parameters and structure, change C 2 = 230 μF. At the same time, set the initial values of the two series capacitors to 1000 V and 2000 V respectively. Figure 16 For the voltage waveforms of the two series voltage-dividing capacitors, it can be found that the control strategy proposed in this patent plays a role in balancing the voltage.

[0154] The specific embodiments of the present invention have been described above. It should be understood that the present invention is not limited to the above specific embodiments, and those skilled in the art can make various changes or modifications within the scope of the claims, which do not affect the essence of the present invention. Without conflict, the embodiments of this application and the features in the embodiments can be combined with each other arbitrarily.

Claims

1. A dual-active-bridge converter with bidirectional differential power transfer, characterized in that, it includes an input side, a transformer and an output side; The input side includes six switching tubes S 1 -S 6 , and two switching tubes are connected in series to form a bridge arm, and three bridge arms are connected in parallel on the power supply side; the output side includes a symmetric full bridge, the symmetric full bridge includes two full bridges with the same structure, the full bridge includes two series-connected diodes and two series-connected switching tubes, and the two series-connected diodes and two series-connected switching tubes are in parallel; The transformer includes a first transformer and a second transformer. A first group of series-connected switching tubes S 1 , S 2 and a second group of series-connected switching tubes S 3 , S 4 are connected to a full bridge through the first transformer. A second group of series-connected switching tubes S 3 , S 4 and a third group of series-connected switching tubes S 5 , S 6 are connected to another full bridge through the second transformer; One end of the primary winding of the first transformer is connected between two series-connected switching transistors S 1 , S 2 , and the other end is connected between two series-connected switching transistors S 3 , S 4 ; One end of the primary winding of the second transformer is connected between two series-connected switching transistors S 3 , S 4 , and the other end is connected between two series-connected switching transistors S 5 , S 6 ; One end of the secondary winding of the first transformer is connected between two series-connected switching transistors Q 1 ,Q 2 of a full bridge, and the other end is connected between two series-connected diodes D 1 ,D 2 of the full bridge; One end of the secondary winding of the second transformer is connected between two series-connected switching transistors Q 3 ,Q 4 of another full bridge, and the other end is connected between two series-connected diodes D 3 ,D 4 of the other full bridge; the other end of the secondary winding of the first transformer is connected to the other end of the secondary winding of the second transformer through a capacitor; the full bridge also includes a capacitor, and the capacitor is in parallel with the two series-connected diodes and two series-connected switching tubes; the capacitors of the two full bridges are connected in series.

2. The dual-active-bridge converter with bidirectional differential power transfer according to claim 1, characterized in that, It further includes a PI regulator for regulating each of the switching transistors S 1 -S 6 , the switching transistor Q 1 -Q 4 phase shift angle.

3. A control method for the dual-active-bridge converter with bidirectional differential power transfer according to claim 1, characterized in that, the voltage of the series-connected capacitors is maintained constant through an equalizing control strategy.

4. The control method for the dual-active-bridge converter with bidirectional differential power transfer according to claim 3, characterized in that, forward power transmission is achieved through the control method, and the equalizing control strategy introduces the duty cycle difference of the switching tubes of the two full bridges as a control variable, and forward power transfer is achieved by adjusting the phase shift angle.

5. The control method for the dual-active-bridge converter with bidirectional differential power transfer according to claim 3, characterized in that, Reverse power transfer is achieved through the control method by adjusting the phase shift angles of each of the switching transistors S 1 -S 6 , the switching transistors Q 1 -Q 4 to achieve reverse power transfer.

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