Radar method and radar system with high range resolution under low signal processing load

Abstract

The invention relates to a method for a radar system for detecting the surrounding environment, the radar system having a transmitting device for radiating a transmission signal comprising a sequence of at least approximately identical individual signals, characterized in that over the sequence of individual signals, the frequency positions of the individual signals (the frequency positions are characterized in particular by their center frequencies) and their time intervals (optionally with the exception of varying and at least approximately mean-free components) vary at least approximately linearly, the relative change in the time intervals being at least approximately twice the relative change in the frequency positions and the signs of these changes being opposite.

Description

Technical Field

[0001] The present invention relates to a method for a radar system (or a method for operating a radar system) and a radar system for a driver assistance system in a motor vehicle. According to the invention, the radar system has a high range resolution with a low signal processing load. Background Art

[0002] More and more motor vehicles are equipped with driver assistance systems. These systems use sensor systems to detect the surrounding environment and, based on the traffic situation thus detected, initiate automatic vehicle reactions and / or provide instructions to the driver, particularly warnings. A distinction is made between comfort functions and safety functions.

[0003] In current developments, Full Speed ​​Range Adaptive Cruise Control (FSRA) plays a key role as a comfort function. If the traffic situation permits, the vehicle adjusts its speed to the driver's preset desired speed; otherwise, it automatically adapts the speed to the traffic situation. Furthermore, lane change assistance systems are becoming increasingly important with the expansion towards at least partially automated lane change functions (Lane Change Assist).

[0004] In addition to improving comfort, safety functions are also gaining increasing attention. Shortening the braking or stopping distance in emergency situations plays a key role. The corresponding driver assistance functions range from automatic pre-charging of the brakes to reduce braking waiting times to autonomous emergency braking.

[0005] Radar sensors are currently the primary method for these types of driver assistance systems. They operate reliably even in adverse weather conditions and, in addition to measuring the distance to an object, can also directly measure the radial relative velocity of the object using the Doppler effect. The transmission frequencies used are typically 24 GHz and 77 GHz.

[0006] The aforementioned functions require a relatively long sensor range, along with high distance measurement accuracy, resolution, and separation performance. High range resolution and separation performance are therefore also important, as they at least partially compensate for the limited angular resolution and separation performance of motor vehicle radar sensors (due to their small size). However, achieving both high range and high range resolution typically requires a high digital signal processing load, which is difficult to achieve because the corresponding signal processors currently used in motor vehicles are only available to a limited extent and / or are expensive.

[0007] DE 10 2013 200 404 A1 and WO 2018 / 086783 A1 propose methods that are intended to allow for a high range and high distance resolution with a moderate digital signal processing load. However, these methods fail to achieve the goal of high distance resolution, particularly at high relative radial velocities, and also have reduced sensitivity, i.e., range. Summary of the Invention

[0008] The object of the present invention is to provide a method for a radar sensor and a radar sensor in which, even for relatively moving objects, a large range and a high distance resolution can be achieved simultaneously with a moderate load on the digital signal processing.

[0009] This object is achieved in principle by means of a radar method or a radar system according to claims 1 and 13. Advantageous embodiments of the invention are set forth in the dependent claims. The invention shows how radar modulation and signal evaluation can be designed to achieve high measurement accuracy and high measurement resolution for the distance to an object and its relative speed.

[0010] The present invention has the advantage that currently available and relatively advantageous signal processors for motor vehicle applications can be used to implement sensors with both high range and high distance resolution, for example to enable the next generation of demanding driver assistance systems. Such signal processors generally have a simple design and, in addition, have the advantage of consuming less electrical energy.

[0011] In a method according to the present invention for detecting the surrounding environment, the radar system includes a transmitting device for radiating a transmission signal, the transmission signal comprising a sequence of at least approximately identical individual signals. Over the sequence of individual signals, the frequency positions of the individual signals (characterized in particular by their center frequencies) and their time intervals (optionally with the exception of varying and at least approximately mean-free components) vary at least approximately linearly. The relative change in the time intervals is at least approximately twice the relative change in the frequency positions, with the signs of these changes being opposite.

[0012] Suitably, the frequency position, time interval and / or phase position of the single signal may be superimposed with a random or pseudo-random component.

[0013] The frequencies of the individual signals are preferably modulated linearly, and the slope of the frequency modulation is at least approximately the same for all individual signals, wherein the individual transmission signals are frequency ramps.

[0014] According to an advantageous embodiment of the method, for K frequency ramps, hereinafter numbered with k=0, ..., K-1, I digital received values, hereinafter numbered with i=0, ..., I-1, can be obtained for each of the multiple receive channels. Subsequently, a two-dimensional discrete Fourier transform can be performed on each of the I·K received values, optionally incompletely and preferably using a one-dimensional fast Fourier transform. The dimensions resulting from the received value exponential dimension i after the transformation can be referred to as range gates j=0, ..., J-1, and the dimensions resulting from the frequency ramp dimensions can be referred to as Doppler gates l=0, ..., L-1.

[0015] Furthermore, the linear variation of the frequency position and the time intervals of the individual frequency ramps can result in sharp power peaks in the received signal of the transmitted signal reflected at the object after a two-dimensional discrete Fourier transform, if the object moves towards or away from the radar system, i.e., has a relative radial motion component.

[0016] The linear change in the frequency position of the frequency ramp can advantageously be taken into account by correcting the position of the object's power peak after a two-dimensional discrete Fourier transform in the Doppler gate dimension l primarily by a component linearly dependent on the range gate dimension j to determine the radial relative velocity of the object. The linearity factor is derived from the quotient of the change in the frequency position on the frequency ramp and the change in frequency within the reception time range during the respective frequency ramp. The position of the power peak is preferably determined by interpolation, which generally results in non-integer values ​​for the range gate dimension j and / or the Doppler gate dimension l.

[0017] Advantageously, the linear change in the frequency position of the individual frequency ramps can be taken into account by, after a one-dimensional discrete Fourier transform of the I received values ​​for each frequency ramp k=0, ..., K-1, correcting the phase of the values ​​generated in range gate dimension j by a phase component proportional to the product 2π·j·k / K, wherein the proportionality factor is essentially the quotient of the change in frequency position on the frequency ramp and the change in frequency within the reception time range during the individual frequency ramps. The correction can then be achieved by multiplying a complex pointer of length 1 with the corresponding phase.

[0018] The sequence of K individual transmission signals can be repeated cyclically, wherein the slope of the linear frequency position variation at the individual transmission signals is varied at least occasionally over the sequence (i.e., at least in one sequence or in one of the multiple sequences), in particular in order to increase the radial distance measurement accuracy and / or the relative speed measurement accuracy and / or to be more stable with respect to interference with other radar systems.

[0019] Each transmitted signal preferably represents a frequency ramp, wherein two periods with opposite slopes, i.e., slopes differing by a factor of -1, are used for precise radial distance measurement and / or relative velocity measurement of the object. The sum and difference of the positions of the object's power peaks generated within two periods after a two-dimensional discrete Fourier transform are primarily used only in the Doppler gate dimension, and not in the range gate dimension.

[0020] Furthermore, the sequence of K individual frequency ramps can be repeated cyclically, wherein, in particular for greater stability with respect to interference with other radar systems, the slope of the frequency ramps is automatically varied at least occasionally over the sequence, ie, at least within a sequence.

[0021] Advantageously, the sequence of K individual transmission signals can be repeated cyclically, wherein, in particular in order to resolve ambiguities in the determination of the radial relative velocity and / or to achieve higher stability with respect to interference with other radar systems, the averaging time interval varies at least occasionally with the sequence, i.e., at least within a sequence.

[0022] Multiple receive channels can preferably be implemented via multiple transmit and / or receive antennas. In addition to the two-dimensional discrete Fourier transformation performed using the I·K receive values, digital beamforming can also be provided on the receive channels or for generating the receive channels.

[0023] The present invention further claims protection for a radar system for detecting an environment, the radar system comprising a transmitting device for radiating a transmission signal, the transmission signal comprising a sequence of at least approximately identical individual signals. The radar system is characterized in that the frequency positions of the individual signals (characterized in particular by their center frequencies) and their time intervals (optionally with the exception of a varying and at least approximately meanless component) can be varied at least approximately linearly over the sequence of individual signals. The relative variation of the time intervals is at least approximately twice the relative variation of the frequency positions, with the signs of these variations being opposite. BRIEF DESCRIPTION OF THE DRAWINGS

[0024] Figure 1 shows an exemplary embodiment of a radar system,

[0025] Figure 2 shows the frequency of the transmitted signal with a constant frequency position according to the prior art, representing a so-called frequency ramp,

[0026] Figure 3 The absolute value spectra of the three objects after two-dimensional discrete Fourier transform and the absolute value spectra according to Figure 2 The frequency change process,

[0027] Figure 4 shows the frequency of the transmitted signal with a linearly varying frequency position,

[0028] Figure 5 The absolute value spectra of the three objects after two-dimensional discrete Fourier transform and the absolute value spectra according to Figure 4 The frequency ramp is a constant frequency ramp.

[0029] Figure 6 shows the absolute value spectrum for a selection of the spacing of the frequency ramps according to WO 2018 / 086783 A1,

[0030] Figure 7 The absolute value spectrum is shown for the inventive selection of the spacing of the frequency ramps. DETAILED DESCRIPTION

[0031] Considered Figure 1 An exemplary embodiment of a radar system is schematically shown in FIG. The radar system comprises a transmitting antenna TX0 for radiating a transmit signal and M=4 receiving antennas RX0 to RX3 for receiving the transmit signal reflected by an object; the antennas are implemented as patch antennas in planar technology on a planar circuit board 1.1, wherein the circuit board is oriented with respect to the horizontal and vertical directions in the vehicle, as shown in FIG. Figure 1 As shown. All antennas (transmitting antennas and receiving antennas) preferably have the same beam characteristics in elevation and azimuth. The four receiving antennas (and thus their phase centers, i.e., radiation centers) have the same lateral, i.e., horizontal, distance d = λ / 2 = 1.96 mm from each other, where λ = c / 76.5 GHz = 3.92 mm is the average wavelength of the radiated signal in the used 76 to 77 GHz frequency band, and c = 3*10 8 Meters per second is the speed of light.

[0032] The transmission signal radiated on the transmitting antenna is obtained from a high-frequency oscillator 1.2 in the range of 76 to 77 GHz, the frequency of which can be controlled by the voltage v 控制 The control voltage is generated in the control device 1 . 7 , wherein these control devices contain, for example, a phase-locked loop or a digital-to-analog converter, which is controlled in such a way that the frequency profile of the oscillator corresponds to the desired frequency modulation.

[0033] The signals received by the four receiving antennas are mixed in parallel in a real-valued mixer 1.3, also together with the signal from oscillator 1.2, into the low-frequency range. The received signals then pass through a bandpass filter 1.4 with the transfer function shown, an amplifier 1.5, and an analog-to-digital converter 1.6. The received signals are then further processed in a digital signal processing unit 1.8.

[0034] To be able to measure the distance of an object, such as Figure 2 As shown, the high frequency oscillator and thus the frequency f of the transmitted signalTX Very fast linear change (at T ch =51.2 microseconds, change B ch =150MHz, where the center frequency is f c =76.5GHz); Here we mention the frequency ramp (often also called "linear frequency modulation signal"). The frequency ramp is on a fixed grid T Dc = 70 microseconds; there are a total of K = 256 frequency ramps, all of which have the same frequency variation, that is, the same frequency slope and the same frequency position (that is, in particular, the same starting frequency and center frequency). In recent years, this modulation method has become increasingly popular in radars used to detect the surrounding environment of motor vehicles.

[0035] During each frequency ramp k=0, ..., K-1, the received signal of each of the M=4 analog-to-digital converters m=0, ..., M-1 is sampled (or scanned) I=256 times at intervals of 200 nanoseconds (i.e. at 5 MHz), wherein the sampling always starts at the same point in time relative to the start of the ramp (see Figure 2 ); the resulting digital sample values ​​with indices i=0, ..., i-1 are denoted by s(i,k,m). Signal acquisition is only meaningful within the timeframe in which the received signal of the object arrives within the distance range of interest—that is, after the ramp begins, one must wait for at least the propagation time corresponding to the maximum distance of interest (for a maximum distance of interest of 99 meters, this corresponds to 0.66 microseconds). It should be noted that, here and below, distance is always understood to be radial distance.

[0036] As is known from the prior art and can also be easily derived, in the case of a single point-shaped object at a distance r, the sampled signal s(i, k, m) represents a sinusoidal oscillation at index i, which can be described in a very good approximation as follows:

[0037]

[0038] That is, the oscillation frequency is proportional to the object distance r, wherein, in general, even in the case of radial relative motion of the object and the sensor, a constant distance can be assumed to a very good approximation for the frequency of the sinusoidal oscillation. However, relative motions with a radial component v are not proportional to the phase position of the sinusoidal oscillation. This has the following impacts:

[0039]

[0040] That is, the phase position changes linearly over the frequency ramp k, with the rate of change of the phase being proportional to the radial relative velocity v of the object. Due to the linearity of the receiver, in the case of multiple and / or extended objects, the sampled signal s(i, k, m) is generated as a linear superposition of sinusoidal functions of the aforementioned shape.

[0041] This signal form allows further processing using a two-dimensional Fourier transform (DFT) comprising a signal window suitable for each reception channel m, wherein the two-dimensional Fourier transform (DFT) is preferably implemented in two stages by means of two one-dimensional fast Fourier transforms (FFT = Fast Fourier Transform). After this two-dimensional discrete Fourier transform (DFT), power peaks appear in the resulting spectrum S(j,l,m), whose respective positions correspond to the distance r and the relative velocity v of the relevant object, see Figure 3 , which shows the absolute value spectrum |S(j,l,m) / A(m)| in dB, independent of the receiving channel m, of three objects with the same radar cross section, at least approximately the same azimuth, and the following distances and relative velocities: [r1=29.5 m, v1=1.09 m / s], [r2=30 m, v2=1.09 m / s], and [r3=45 m, v3=60.4 m / s]; the receiver noise is also superimposed on the object signal, which is significantly lower in the spectrum than the power peak of the object marked with the object number. The dimensions j = 0, ..., J-1 resulting from dimension i (sample value index) are represented by range gates, and the dimensions l = 0, ..., L-1 resulting from dimension k (frequency ramp) are represented by Doppler gates. This is because the position of the power peak in dimension j is primarily determined by the distance to the object, and in dimension l by the relative velocity (which is reflected by the Doppler effect). However, in this case, it is negligible that the power peak position also has a very small dependence on the other of the two physical variables, namely, distance and relative velocity. It should be noted that the velocity cannot be unambiguously calculated from the Doppler gates of the power peak, because in the proposed design, only an unambiguous range of 28 meters per second is achieved with K = L = 256 Doppler gates. Ambiguity can be introduced, for example, by the spacing T of the frequency ramps. Dc This is achieved by varying the radar cycle (as explained later). Figure 3 , the number of range gates is only J=100 and is therefore significantly smaller than the number of sample values ​​I=256; the background is that, on the one hand, the sample values ​​are real-valued, so that their spectrum is symmetrical, i.e., no additional information is contained in the upper half of the discrete Fourier transform (DFT), and on the other hand, according to Figure 1 The upper transition range of the simulated bandpass filter 1.4 has a frequency bandwidth of 1.09 MHz (corresponding to a range of 56 frequency interpolation points). ch=150MHz, the range gate width B ch / 150MHz 1m is exactly = 1 meter, so J = 100 range gates allow a maximum range of 99 meters.

[0042] from Figure 3 As can be seen, the first two objects with [r1 = 29.5 m, v1 = 1.09 m / s] and [r2 = 30 m, v2 = 1.09 m / s] cannot be separated, but rather merge into one power peak, because they have the same relative velocity and only slightly different distances. Their distance difference is 0.5 m, and thus only half a range gate. Typically, a difference of approximately two range gates is required to separate two point-like objects with the same relative velocity. For the distance separation performance of these two objects, a significantly higher modulation bandwidth B is required. ch , at least 4 times higher, that is, B ch = 600MHz, which results in a range gate width of B ch / 150MHz = 1m = 0.25m. For the same maximum sensor range of approximately 99m, each frequency ramp requires four times more sampling values. This, on the one hand, requires a faster analog-to-digital converter, and, more seriously, on the other hand, requires approximately four times more processing power and memory in the digital signal processing device.

[0043] In order to avoid this, alternative modulation forms can be used, such as are known from DE 10 2013 200 404 A1 and are described in Figure 4 Compared with the previous modulation form shown in Figure 2 The modulation form considered is one in which the only changes are the frequency positions, in particular the start and center frequencies F c The frequency position marked by (k) is now on K=256 frequency ramps with frequency B s / K, where B s =600MHz linearly; this effectively achieves a higher modulation bandwidth and thus results in significantly better distance separation performance. The spacing of the frequency ramp is constant, that is, it is constant, T D (k) = 70 microseconds. In this modulation format, the signal processing in the form of two-dimensional discrete Fourier transform (DFT) can remain unchanged. According to the above example, the absolute value spectrum |S(j,l,m) / A(m)| obtained for the three objects is Figure 5 As shown in. Figure 3Compared to the original absolute value spectrum of , the position of the power peak now shifts in the Doppler gate dimension l, but not in the range gate dimension j. This is because the linear increase in the frequency position by the frequency ramp increases the number of wave trains in the beam path from the sensor to the corresponding object and back (the wavelength decreases with increasing frequency), which is reflected in the phase position of the received value s(i,k,m) according to the formula or equation (1) The component that changes linearly in the frequency ramp k has an effect; this component is superimposed on the component caused by the radial relative motion and is also linear in k according to equation (2), so that both components have essentially the same effect, namely a shift of the power peak in the Doppler gate dimension. As will be shown later, the shift in the Doppler gate dimension l caused by the change in frequency position is approximately B of the range gate dimension j of the corresponding object. s / B ch times.

[0044] In accordance with Figure 5 It can be seen from the spectrum of [r1=29.5 m, v1=1.09 m / s] and [r2=30 m, v2=1.09 m / s] that the first two objects are now separated, i.e. form two independent power peaks, wherein the separation is carried out in the Doppler gate dimension, since the slightly different distance of 0.5 m leads to different shifts in the two Doppler gates caused by the change in frequency position (the difference in the Doppler gate dimension is higher than the difference in the range gate dimension by B). s / B ch times, i.e. 4 times higher, where the difference is half a range gate).

[0045] However, according to Figure 5 The disadvantage of the spectrum is that the third object with [r3 = 45 meters, v3 = 60.4 meters / second] no longer has a sharp power peak, but rather has a very strong spread in the Doppler gate dimension. This in turn leads to several disadvantages: first, the possible detection range is reduced (because the power level is reduced), second, the relative velocity measurement becomes less accurate (because the power peak is blurred), and third, it is no longer possible to separate another target with the same relative velocity but a slightly different distance (because the blurred power peaks overlap). The greater the absolute value of the relative velocity, the greater the dissipation of the power peak; in the case of the first two objects, the effect is still not visible because their relative velocity is very small.

[0046] It is proposed in WO 2018 / 086783 A1 that a central frequency F c The spacing T of the frequency ramps k=0, ..., K-1 of (k) D (k) is no longer kept constant, but is changed so that T D (k)·F c(k) is constant. Then, for the above example, we can get Figure 6 The absolute value spectrum |S(j,l,m) / A(m)|. According to Figure 5 The spectrum is different in that the dissipation of the power peak of the third object with high relative velocity [r3=45 m, v3=60.4 m / s] is now smaller (approximately halved), but still exists and is unacceptably large.

[0047] Therefore, a method according to the invention is now derived which prevents blurring of power peaks even at high relative speeds.

[0048] The relative time t∈[-T ch / 2, T ch / 2], for the high-frequency oscillator and the frequency f of the transmitted signal TX (t,k) is applicable to:

[0049] f TX (t,k)=F c (k)+B ch / T ch t (3)

[0050] Wherein, the center frequency F of the frequency ramp k=0, ..., K-1 c (k) is:

[0051] F c (k) = F cc +B s / K·(k-(K-1) / 2), (4)

[0052] Among them, F cc is the center frequency F c (k) The average value. By integration, the phase of the oscillator signal and the transmitted signal This is given by the following equation:

[0053]

[0054] The integration constant has no influence here and is therefore omitted.

[0055] For a single point object, the phase of the received signal at the mixer output is Produced by the phase difference between the current oscillator signal and the signal reflected from the object, it is delayed by the transmission time Δt:

[0056]

[0057] Among them, s ch Indicates the linear frequency modulation signal modulation bandwidth Bch = +1 represents an increasing frequency ramp, and = -1 represents a decreasing frequency ramp. The received signal after the mixer is also called an intermediate frequency signal (IF). The sampled signal s(i, k, m) of the relevant receiving channel m is obtained by sampling at time t∈[-T ch / 2,T ch / 2] is obtained by forming I sample values ​​with indexes i=0, ..., I-1.

[0058] The transit time Δt is calculated for an object with radial relative velocity v:

[0059] Δt=2(r c (k)+vt) / c; (7)

[0060] Among them, r c (k) is the distance of the object at the center of frequency ramp k:

[0061] r c (k) = r + v·T c (k), (8)

[0062] Where r is the average distance over all frequency ramps, T c (k) is the absolute time of the center of frequency ramp k (the absolute time of the centers of all frequency ramps is defined as 0). A constant relative speed is assumed here, since the entire sequence of K frequency ramps lasts only a short time, for example ≤ 20 ms.

[0063] After transformation and omitting negligibly small terms, the phase of the intermediate frequency signal is obtained from equations (5) to (7) as follows:

[0064]

[0065] The average intermediate frequency signal phase (ie at t=0) on the frequency ramp k is obtained as follows:

[0066]

[0067] From equation (9), the frequency of the intermediate frequency signal, that is, the intermediate frequency itself, can be derived:

[0068] f IF (t,k)=|B ch | / T ch ·2(r c (k)+v·t) / c+s ch ·(F c (k)+B ch / T ch ·t)·2v / c. (11)

[0069] For the average intermediate frequency f of the frequency ramp k IF (k) (i.e. at t = 0) yields:

[0070] f IF (k)=|B ch | / T ch 2r c (k) / c+s ch ·F c (k)·2v / c; (12)

[0071] The first component reflects the distance-dependent effect of the linear frequency modulation, the second component represents the Doppler effect, i.e. the frequency shift due to the relative motion, wherein the second component is usually significantly smaller than the distance-dependent component. In the average over all frequency ramps, the following equation is obtained with the average distance r (see equation (8)) and the average center frequency F cc (See equation (4)) the intermediate frequency f IF :

[0072] f IF =|B ch | / T ch 2r / c+s ch ·F cc 2v / c. (13)

[0073] If a one-dimensional discrete Fourier transform is formed on the frequency ramp k and the sampled signal s(i, k, m) of the receiving channel m, then at the range gate j(k) = f IF (k)*T ch The power peak is obtained at , that is, using equation (12) we get:

[0074] j(k)=|B ch |·2r c (k) / c+s ch ·F c (k)·T ch 2v / c, (14)

[0075] And in the average of all range gates, we get from (13):

[0076] j=|B ch |·2r / c+s ch ·F cc ·T ch 2v / c, (15)

[0077] This usually means that j(k) or a non-integer value of j, i.e. the actual maximum of the power peak lies between two integer range gates considered in the discrete Fourier transform (DFT), the position of which can be determined by interpolation. After the two-dimensional discrete Fourier transform (DFT), the power peak is located at the average range gate j according to equation (15). The variation of the range gate j(k) according to equation (14) on the frequency ramp k is mainly caused by the slight change in the relative speed of the range r c The effect of (k) is relatively small, however, because the distance changes only slightly over the short time span of the total K frequency ramps (typically in the range of ≤ 20 milliseconds). After the two-dimensional discrete Fourier transform (DFT), this only leads to a slight broadening of the power peak in the range gate dimension. The first component of the range gate j according to equation (15) is generated by the distance r of the object, and the second component is generated by its relative velocity v; the second component is usually much smaller than the first component, so that the range gate is mainly determined by the distance.

[0078] From the average intermediate frequency signal phase according to equation (10) Using the average distance r according to equation (8) c (k) It is concluded that:

[0079]

[0080] The first component in the frequency ramp k changes linearly (because the center frequency F c (k) Linear change). For the constant spacing of the frequency ramp studied at the beginning of this article, that is, the time T of the linear change of the center of the frequency ramp c (k), the second component is nonlinear for relative velocity v≠0, because the corresponding linear term T c (k) and F c (k) appears in the product. The nonlinear characteristics of the frequency ramp dimension k after the second one-dimensional discrete Fourier transform (DFT), the resulting Doppler gate dimension l does not produce a sharp power peak; from T c (k)·F c (k)·s ch The higher the nonlinear component of 2v / c, and thus the higher the relative speed, the more blurred the power peak (as in Figure 5 (as can also be seen in the example).

[0081] In order to avoid the ambiguity associated with the relative velocity, the second component of equation (16) It must also be linear in k, that is:

[0082] s ch ·Tc (k)·F c (k)·2v / c=(k-(K-1) / 2)·constant

[0083] By following T c (k) Solve this equation and substitute the average ramp frequency F according to equation (4) c (k), neglecting very small terms yields:

[0084] T c (k) = (k-(K-1) / 2)·T Dc / (1+(k-(K-1) / 2) / K·B s / F cc ) (17)

[0085] Among them, T Dc = constant s ch / (2v / c·F cc );

[0086] As can be seen from equation (17), the parameter T Dc is the average spacing of the frequency ramps (i.e., the average sampling time used to obtain the Doppler gate dimension by the second discrete one-dimensional Fourier transform, which symbolizes T Dc "D" in the index). Due to the modulation bandwidth B on the sequence of frequency ramps s Usually higher than the average transmission frequency F cc is much smaller, so the denominator of equation (17) is of the form (1+x), where |x| << 1, so that for example the series expansion up to the second order term 1 / (1+x) = 1-x+x 2 -+... can be used as an excellent approximation:

[0087] T c (k) = (k-(K-1) / 2)·T Dc (1-(k-(K-1) / 2) / K B s / F cc )+(k-(K-1) / 2) / K·B s / F cc ) 2 ). (18)

[0088] The time interval T between two adjacent frequency ramps D (k) = T c (k)-T c (k-1) Using equation (18) and omitting negligibly small terms, we obtain:

[0089] T D (k) = T Dc·(1-2((kK / 2) / K·B s / F cc )+3((kK / 2) / K·B s / F cc ) 2 ); (19)

[0090] Because the third component contains a very small ratio B in quadratic form s / F cc , and therefore usually better than in B s / F cc The second linear component is much smaller, so it can also be ignored or omitted:

[0091] T D (k) = T Dc ·(1-2(kK / 2) / K·B s / F cc ). (20)

[0092] Therefore, the time interval of the frequency ramp varies at least approximately linearly over the frequency ramp k. According to equation (20), the frequency ramp spacing T D (k) = T c (k)-T c The relative change of (k-1) on the frequency ramp k=1, ..., K-1 is:

[0093] (T D (k)-T Dc ) / T Dc =-2(kK / 2) / K·B s / F cc . (twenty one)

[0094] According to equation (4), the central frequency F that changes linearly on the frequency ramp k=0, ..., K-1 is obtained: c Relative change of (k):

[0095] (F c (k)-F cc ) / F cc = +(kK / 2-1 / 2) / 2) / K·B s / F cc . (twenty two)

[0096] As can be seen from equations 21 and 22, the slope of the linear relative change of the center frequency of the frequency ramp = +B s / F cc , and the slope of its time interval = -2B s / F cc, i.e. the relative change in the time interval is twice the relative change in the frequency position of the frequency ramp, wherein the signs of these changes are opposite. It should be noted that in the case of an exact determination of the time interval, for example according to equation (19), the correlation of the relative changes is not completely accurate, but only approximately given. For the above considered modulation bandwidth B s =600MHz, average frequency F cc = 76.5 GHz, the relative change in frequency position over the entire sequence of K frequency ramps is approximately 0.78%, and the relative change in their time intervals is 1.56%. It should also be noted that when the ramp spacing is designed according to WO 2018 / 086783 A1, the time intervals of the frequency ramps and the relative change in frequency position are opposite and equal in absolute value, i.e., they do not differ by a factor of 2 in absolute value.

[0097] The choice of the time interval of the frequency ramp (i.e. T according to equation (20) D (k)), which is generated after the two-dimensional discrete Fourier transform (DFT) Figure 7 The absolute value spectrum |S(j,l,m) / A(m)| is shown. Figure 6 The power peak of the third object with a high relative velocity (r3 = 45 m, v3 = 60.4 m / s) is now also sharp, meaning dissipation has been prevented. This also has an effect at a level that is approximately 2 dB higher. The two objects with the same relative velocity and only slightly different distances (r1 = 29.5 m, v1 = 1.09 m / s) and (r2 = 30 m, v2 = 1.09 m / s) remain unchanged.

[0098] Now, the position of the power peak of the object in the Doppler gate dimension must also be determined. For this purpose, the time T of the center of the frequency ramp determined above according to equation (17) is c (k) is inserted into the intermediate frequency signal phase according to equation (16) In use for center frequency F c In the case of equation (4) for (k), omitting the irrelevant constant phase component yields:

[0099]

[0100] The duration of the entire frequency ramp sequence is T s for:

[0101] T s = K·T Dc ; (twenty four)

[0102] It is again emphasized that, as required and by selecting T accordinglyc (k), which represents a linear phase change process on k.

[0103] If we now form a second one-dimensional discrete Fourier transform on the frequency ramp dimension k, then at the Doppler gate The power peak is generated at , which can be expressed by equation (23):

[0104] l = s ch ·(B s ·2r / c + F cc ·T s ·2v / c); (25)

[0105] The first component is derived from the distance r of the object, the second from its relative velocity v. Unlike the range gate j according to equation (15), which is controlled solely by the object size, ie its distance, the Doppler gate takes both relative velocity and distance into account to a similar degree.

[0106] As can be seen by comparing equations (25) and (15) for the generated Doppler gate l and range gate j, the effect of range in the Doppler gate dimension is higher than in the range gate dimension. s / |B ch times, which results in a corresponding improvement in distance separation performance.

[0107] Now we also rewrite equations (15) and (25) for the range and Doppler gates in a special way, that is, the range and relative velocity are related to their gate lengths:

[0108] j=r / R Lch +s ch v / D Lch (26)

[0109] l=s ch ·r / R Ls +s ch v / D Ls (27)

[0110] Wherein, the range and Doppler gate length are:

[0111] R Lch =c / (2|B ch |), R Ls =c / (2B s ), D Lch =c / (2F cc T ch ), D Ls =c / (2F cc T s). (28)

[0112] In sensor applications, the distance and relative velocity of an object are unknown, but the underlying task is to determine them from the position of the power peak after a two-dimensional discrete Fourier transform (DFT). Therefore, the two equations (26) and (27) are solved based on the distance r and the velocity v; this yields:

[0113] r=R L ·(jl·T ch / T s ) (29)

[0114] v=D L ·s ch (lj·B s / B ch ) (30)

[0115] Among them, the modified gate length is

[0116] R L =R Lch / (1-B s / B ch ·T ch / T s ), D L =D Ls / (1-B s / B ch ·T ch / T s ). (31)

[0117] The range gate j and Doppler gate l of an object are generally non-integer and can be determined by interpolation from the shape of the power peak in the two-dimensional discrete Fourier transform (DFT), which only provides values ​​at integer gates.

[0118] Furthermore, it must be taken into account that the Doppler gate l can usually lie within a larger value range than the unambiguous range L=K of the discrete Fourier transform (DFT); the Doppler gate can therefore only be determined from the discrete Fourier transform (DFT) up to an unknown integer multiple of K. A method for resolving the ambiguity is to vary the average frequency ramp spacing T over the radar period, similar to the method proposed in DE 10 2009 016 480 A1. Dc , i.e., in the sequence of K frequency ramps transmitted in the current radar cycle, a different T is used than in the previous sequence. Dc By changing D in Equation (27) Ls, generating a different value of the Doppler gate l in the current radar cycle than in the previous radar cycle, while the relative speed is approximately the same, enables the ambiguity to be resolved (the relative speed may vary only slightly over the radar cycle, typically around 50 milliseconds).

[0119] According to equation (30) for determining the relative velocity of an object, the frequency position (in terms of B) is taken into account by the linear variation of the frequency ramp in the following manner s ≠0 is the characteristic), that is, the component jB proportional to its range gate is subtracted from the Doppler gate l of the generated power peak. s / B ch ; In addition, according to equation (31), B s ≠0 will still slightly affect the Doppler gate width D L .

[0120] Alternatively, the influence caused by the linearly varying frequency position can also be taken into account by subtracting 2π·j·B in each case after a one-dimensional discrete Fourier transform of the I received values ​​of each frequency ramp k=0, ..., K-1. s / B ch k / K corrects the phase of the value of the range gate dimension j for all j and k (i.e., regardless of whether an object is present there, which may also be unknown at this point in time); the correction can be achieved by multiplying a complex pointer of length 1 and the corresponding phase.

[0121] As described above, in order to determine the range gate and Doppler gate of an object, the exact position of the power peak is obtained by interpolation; in particular, due to the signal window used in the discrete Fourier transform (DFT), the power peak has a level not only in one gate, but also in at least one adjacent gate, so that the actual position, which is generally non-integer, can be determined from the shape of the power peak, for example, by parabolic interpolation or by using the known shape of the power peak (resulting from the discrete Fourier transform of the window function itself). However, this interpolation is not arbitrarily accurate; for example, interpolation errors can occur due to superimposed noise (especially in the case of a poor signal-to-noise ratio) or due to extended, i.e. non-point-shaped, objects. This leads to inaccuracies in the determination of the distance and relative velocity of the object according to equations (29) and (30); for the relative velocity according to equation (30), it is particularly important that the range gate is there by a factor B. s / B ch is taken into account (in the above example, a factor of 4). Almost only the range gate is considered in equation (29) for determining the range (the Doppler gate has only a small weight T ch / T s), and thus almost only the interpolation error (originating from the range gate) is taken into account; however, this error is Lch =c / (2|B ch |) is taken into account, i.e. not with the usually significantly smaller door width R Ls =c / (2B s ) is taken into account, that is, for the accuracy of distance determination, no large modulation width B s , and therefore does not benefit from the change in frequency position on the frequency ramp (which has so far mainly only improved the range separation performance of objects with the same relative speed). Therefore, the inaccuracy, both for range and for relative speed, mainly comes from the error of the range gate.

[0122] However, inaccuracies in the determination of the distance and relative speed of the range gate error can be avoided by not always performing the same frequency ramps for the modulation bandwidth B. s The same sign is used, but the absolute value is kept constant and is varied over several radar cycles; that is, for example, +B is used alternately. s and -B s , so that the frequency position increases linearly on the frequency ramp every other radar cycle and decreases linearly in the other radar cycles. As a result, the sign of the range component changes in the Doppler gate l according to equation (27); if now from two with B s The sum of the Doppler gates of the object for the different sign of the radar cycle is obtained. Roughly speaking, the distance component is eliminated and the relative velocity is obtained. If the difference of the Doppler gates is formed, the opposite is true. In addition, it must be taken into account that, on the one hand, the distance varies slightly with the radar cycle for the relative velocity v≠0, and on the other hand, the average spacing T of the frequency ramps Dc varies over multiple radar cycles. After some intermediate calculation steps and simplifications, the range r averaged over two cycles can be obtained m and relative velocity v m :

[0123] v m =s ch ·D Ls+- ·(l + +l - ) / 2 / (1-D Ls+- ·t +- / (2|R Ls |)) (32)

[0124] r m =|R Ls |·(s ch ·(l + -l -) / 2-v m / 2·(1 / D Ls+ -1 / D Ls- )) (33)

[0125] Among them, l + is the Doppler gate within the first radar cycle, with positive modulation bandwidth +B s , and l - is time t +- The Doppler gate in the next radar cycle after has a negative modulation bandwidth -B s ; "Average" Doppler gate width D Ls+- From two radar cycles (at different average frequency ramp intervals), if necessary at different Doppler gate widths D Ls+ and D Ls- It is concluded that:

[0126] D Ls+- =2 / (1 / D Ls+ +1 / D Ls- ). (34)

[0127] Therefore, to determine the distance and relative speed of an object, only the Doppler gate of the object is required from two radar cycles, and no longer a range gate, which could lead to significant errors in previous methods. Large modulation bandwidth B s Small door width R Ls It is now also relevant for determining the distance that the interpolation error is taken into account in a correspondingly small manner.

[0128] Accurate distance measurement is important, for example, for avoiding collisions with obstacles (such as guardrails) or other vehicles located on the side of the vehicle, especially at close ranges. In this case, the distance is often less than the large door width R Lch =c / (2|B ch |), i.e., within the first range gate where the interpolation function is generally particularly poor (due to reflections from the bumper and / or the superposition of negative frequency components). s The distance is determined by the Doppler gate with a period of 1 / 4.5, so that the short distance can also be accurately determined.

[0129] In the above embodiment, the modulation bandwidth B s The sign of has changed within two radar cycles, while the absolute value remains constant. However, it is sufficient in principle to change B within two radar cycles. s The value of and / or the slope of the linear frequency position change are adjusted so that the influence of the range gate can be eliminated. Then, a weighting factor also appears in the sum and difference of the required Doppler gates, that is, the weights of the Doppler gate values ​​generated in the two cycles are different.

[0130] In order to stabilize the radar system with respect to interference from other radar systems, the modulation parameters are preferably varied, in particular analogously to the methods described in WO 2008 / 040341 ​​A1, DE 10 2009 016 480 A1 and EP 2 629 113 B1, for example:

[0131] - the average spacing of the frequency ramp over the period (as mentioned above, additionally also allows a simple resolution of velocity ambiguities);

[0132] - Modulation bandwidth B with period s and / or B ch (absolute value and / or sign);

[0133] - by the additional superposition of random or pseudo-random meanless components that vary over k, typically in the range of a few microseconds at most, the time interval T of the frequency ramp according to equations (19) and (20) D (k); For relatively moving objects, the received phase has a component that varies slightly on the frequency ramp, but it is still very small, so that the resulting effects (reduction in the level of noise and power peaks) after discrete Fourier transform (DFT) are negligible.

[0134] - by the additional superposition of random or pseudo-random meanless components varying in k, the frequency position F of the frequency ramp according to equation (4) c (k) (i.e., its center frequency); this change in the frequency position can also be achieved by always using the same frequency ramp, but changing the time at which the received signal sampling values ​​are started; the resulting phase change of the received signal, which is proportional to the range gate, can be compensated by a corresponding general phase correction after the first one-dimensional discrete Fourier transform (DFT);

[0135] The phase position of the individual transmission signals passes through an additional phase modulator in the transmission device, wherein the phase position varies randomly or pseudo-randomly over the frequency ramp, which is compensated again on the receiving side, preferably in a digital signal processing device.

[0136] In accordance with Figure 1The radar system under consideration has M = 4 receive antennas and associated receive channels m = 0, ..., M-1. After a two-dimensional discrete Fourier transform (DFT), digital beamforming is preferably calculated in each range-Doppler gate (j, l), for example, again in the form of a discrete Fourier transform (DFT) or a fast Fourier transform (FFT); thus, a three-dimensional Fourier transform is performed. The power peak is then determined in the three-dimensional spectrum. The azimuth angle of an object is derived from the position of its power peak in the third dimension, which is derived from the dimension m of the receive channel; the range and relative velocity are derived from the other two dimensions according to the aforementioned relationship. To provide more channels for angle formation, it is preferred to use not only multiple receive antennas but also multiple transmit antennas. The signals of all transmit and receive antenna combinations are evaluated to achieve as many virtual receive channels as possible. If all or some transmit and / or receive antennas are not operated simultaneously, multiple, preferably similar, frequency ramp sequences of the type described above are nested within each other.

[0137] In summary, the method presented here, as an example, allows for high-precision and high-resolution distance measurement by using a high modulation bandwidth. This approach, while not only preventing degradation of measurement and detection quality when relatively moving objects, but also eliminating the need for high computational performance in a digital signal processing device (as is required with conventional methods utilizing high modulation bandwidths). The fact that only moderate computational performance is required stems, on the one hand, from the fact that the discrete Fourier transform can be used as a fast Fourier transform (FFT) in its fast implementation for calculations, and, on the other hand, from the fact that the multidimensional fast Fourier transform (FFT) has fewer dimensions than conventional methods with high range resolution and high measurement accuracy, as the range measurement is partially shifted to the dimension where the relative velocity measurement is also performed. This exploits the fact that, in motor vehicle radar systems for ambient detection, high range resolution is primarily required for targets with the same radial relative velocity. Examples of radar systems for ambient detection of a vehicle include measuring the length and width of a traffic jam ahead, stopped vehicles under a bridge or near a guardrail, stationary road surroundings (guardrails, trees, buildings, etc.), and other vehicles, each of which typically has numerous reflection points. Therefore, good distance separation performance is also important, because the angular separation performance of radar systems is relatively poor due to the generally large beamwidth (caused by the limited size). This can, for example, result in reflections from the left and right guardrails not being separated and fused, so that the measured angle lies in the own lane and a stationary obstacle (e.g. a parked vehicle) is incorrectly identified.

[0138] It's also worth mentioning that this method only partially demonstrates its advantages in scenarios involving multiple targets with slightly different relative speeds and distances, because the total number of detection gates, specifically the range-Doppler gates, is not increased by increasing the modulation width over the linear variation in frequency position of the frequency ramp. However, these scenarios are generally rarely relevant for the aforementioned driver assistance functions.

[0139] It should be noted that the ideas and embodiments according to the present invention shown in the above application examples can be transferred to general measurements and parameter designs, that is, they can also be used for other values. Therefore, general parameters are also given in the equations and figures.

Claims

1. A method for detecting the surroundings of a radar system having a transmitting device for radiating a transmission signal containing a sequence of at least approximately identical individual signals, characterized in that Over a sequence of individual signals, the frequency positions of the individual signals and their time intervals, apart from an optional varying and at least approximately meanless component, vary at least approximately linearly, wherein the relative change in the time interval is at least approximately twice the relative change in the frequency position and the signs of these changes are opposite.

2. The method according to claim 1, wherein The frequency position of each signal is characterized by its center frequency.

3. The method according to claim 1, wherein Random or pseudo-random components are superimposed on the frequency position, time interval and / or phase position of the individual signals.

4. The method according to any one of claims 1 to 3, wherein The frequency of the individual signals is modulated linearly, and the slope of the frequency modulation is at least approximately the same for all individual signals, wherein the individual transmit signals are subsequently referred to as frequency ramps.

5. The method according to claim 4, wherein During the K frequency ramps numbered k=0, ..., K-1 below, I digital received values ​​numbered i=0, ..., I-1 below are obtained respectively, and a two-dimensional discrete Fourier transform is performed on the I·K received values ​​respectively, wherein the dimension generated by the received value index dimension i after transformation is called the range gate j=0, ..., J-1 below, and the dimension generated by the frequency ramp dimension is called the Doppler gate l=0, ..., L-1.

6. The method according to claim 5, wherein: The linear variation of the frequency position and the time intervals of the individual frequency ramps result in sharp power peaks in the received signal of the transmitted signal reflected at the object after a two-dimensional discrete Fourier transform, even if the object moves towards or away from the radar system, i.e. has a relative radial motion component.

7. The method according to claim 5 or 6, wherein: The linear change in the frequency position of the frequency ramp is taken into account in order to determine the radial relative velocity of the object by correcting the position of the power peak of the object after the two-dimensional discrete Fourier transform in the Doppler gate dimension l by a component that is linearly dependent on the range gate dimension j, wherein the linear factor is obtained from the quotient of the change in the frequency position on the frequency ramp and the change in the frequency within the reception time range during the respective frequency ramp.

8. The method according to claim 7, wherein: The positions of the power peaks are determined by interpolation, whereby non-integer values ​​are generated for the range gate dimension j and / or the Doppler gate dimension l.

9. The method according to claim 5 or 6, wherein: The linear change in the frequency position of the individual frequency ramps is taken into account by the fact that, after a one-dimensional discrete Fourier transform of the I received values ​​for each frequency ramp k=0, ..., K-1, the phase of the values ​​generated in the range gate dimension j is corrected by a phase component proportional to the product 2π·j·k / K, wherein the proportionality factor is the quotient of the change in the frequency position on the frequency ramp and the frequency change within the reception time range during the individual frequency ramps.

10. The method according to claim 9, wherein: Correction is achieved by multiplication with a complex vector of length 1 and with the corresponding phase.

11. The method according to any one of claims 1 to 3, wherein The sequence of K individual transmit signals is repeated cyclically, and in this process the slope of the linear frequency position change at the individual transmit signals is varied over the sequence at least in one of the sequences.

12. The method according to claim 11, wherein Each transmitted signal represents a frequency ramp, and two periods with opposite slopes are used for precise radial distance measurement and / or relative velocity measurement of the object, wherein the sum and difference of the positions of the power peaks of the object after a two-dimensional discrete Fourier transform, which are generated in the two periods, are used only in the Doppler gate dimension and not in the range gate dimension.

13. The method according to any one of claims 1 to 3, wherein The sequence of K individual frequency ramps is repeated cyclically, and in this case the slope of the frequency ramp itself is varied over the sequence at least in one of the sequences.

14. The method according to any one of claims 1 to 3, wherein The sequence of K individual transmission signals is repeated cyclically, and the average time interval of the transmission signals is varied over the sequence at least in one of the sequences.

15. The method according to any one of claims 1 to 3, wherein A plurality of transmit and / or receive antennas realizes a plurality of receive channels, and in addition to a two-dimensional discrete Fourier transformation respectively performed via I·K receive values, digital beamforming is also present on the receive channels or for generating the receive channels.

16. A radar system for detecting surroundings, comprising a transmitting device for radiating a transmission signal comprising a sequence of at least approximately identical individual signals, characterized in that Over a sequence of individual signals, the frequency positions of the individual signals and their time intervals, apart from an optional varying and at least approximately meanless component, vary at least approximately linearly, wherein the relative change in the time interval is at least approximately twice the relative change in the frequency position and the signs of these changes are opposite.

17. The radar system of claim 16, wherein: The frequency position of each signal is characterized by its center frequency.

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