Optoelectronic distance measuring module and surveying device

Abstract

Optoelectronic distance measuring module and surveying apparatus. The invention relates to an optoelectronic distance measuring module and to a surveying apparatus (1) comprising the optoelectronic distance measuring module. The optoelectronic distance measuring module comprises a laser diode or a low-power fiber laser as a transmitting unit of a distance measuring light beam (15), wherein a finite transmission code sequence (16, 16') with N chips (17) is transmitted and cross-correlated with a reference signal (30) for generating a compressed pulse (35, 36, 37). In order to provide sufficient pulse compression and accurate timing of the compressed pulse (35, 36, 37), the reference signal (30) is configured to have a number of chips (17) greater than N and is formed by reference signal components (33, 33') occupying a bipolar shape of the expected receiver output signal components assignable to the respective light pulses (18) of the transmission code sequence (16, 16').

Description

Technical Field

[0001] This invention relates to an optoelectronic distance measurement module configured for use in surveying equipment used in geodesy, construction, or topographic mapping. For simplicity, such surveying equipment will be referred to hereinafter as a so-called "surveying equipment." For example, such surveying equipment may be implemented as a velocimeter, total station, laser profilometer, or laser scanner. Background Technology

[0002] Coordinate measurement technology equipment used to measure target objects (e.g., surveying instruments) often operates based on electro-optic measurement systems. These devices typically emit light radiation (usually laser radiation in the visible or near-infrared spectral range) in the direction of the target object to be measured in order to determine the distance between the device and the target. The direction in which the target object is sampled can also be determined using angular measuring devices. By measuring the distance and angular position of the target, the target's 3D coordinates (e.g., given in polar coordinates) are determined and are typically subsequently processed further. For example, the 3D coordinates are stored as a point cloud, which is further processed to generate a digital 3D spatial model of the environment (e.g., a vector file model or a mesh model).

[0003] In surveying, construction, or mapping processes, the target object being measured reflects a portion of its emitted radiation back to the equipment, where it is received and converted into an electrical signal for distance determination. Besides naturally occurring targets, man-made targets can also be measured, such as so-called cooperative targets, which are configured to provide specific reflective properties to support precise coordinate measurements or target point identification. For example, cooperative targets are implemented as reflectors or retroreflectors, such as cubic corner reflectors, cat's-eye reflectors, or reflective strips; or as spherical tool spheres.

[0004] Especially for surveying equipment with scanning capabilities, the measured and processed data can be processed to represent a digital twin of the scanned environment or object surface. These surveying instruments are equipped with a laser ranging module (LiDAR), which preferably includes a laser source that provides high peak power in the form of pulses in the kilohertz or megahertz range to generate a highly sensitive measurement beam. Such lasers are typically fiber lasers, laser diode seed amplifiers (e.g., fiber amplifiers), or solid-state lasers (e.g., microchip lasers). These lasers are generally complex, require several watts of electrical power, and are expensive.

[0005] There is now a need to further miniaturize surveying equipment, for example, to provide improved mobility and increased application options, while reducing complexity and simplifying ease of use without sacrificing sensitivity and accuracy. Conventional laser devices often hinder the achievement of these goals. For instance, the packaging and overall setup of surveying equipment are often driven by the requirements of laser devices, which are typically bulky, heavy, and generate significant heat.

[0006] Conversely, laser diodes or low-power seeded single-mode fibers are smaller, offering excellent insertion efficiency and relatively low cost. Unfortunately, the peak power of such compact lasers is limited, with single-mode laser diodes, in particular, typically having a peak power of less than 10 watts. Such single-mode laser diodes are advantageous, for example, when measuring well-defined small spot sizes over long distances such as 100m or 1km. Therefore, most scanning survey equipment is equipped with multi-stage fiber lasers, which include spatial single-mode fiber on the output side, delivering high peak power of several kilowatts. However, when the measurement rate must be in the megahertz range, laser diodes are generally too weak to be used for accurately measuring distances over such long measurement ranges. Summary of the Invention

[0007] Therefore, the purpose of this invention is to provide a photoelectric distance measurement module that overcomes the above-mentioned defects of the prior art.

[0008] The specific objective is to provide an electro-optical distance measurement module, particularly for long measurement ranges, which offers further miniaturization, reduced complexity, and simplified usability of surveying equipment.

[0009] According to one aspect of the invention, these objectives are achieved by providing an optoelectronic distance measurement module, wherein the beneficial implementation and use of the laser diode is carried out in a manner that allows the laser diode to provide long-distance and high-speed distance measurement.

[0010] For example, emitting low-energy pulse groups—that is, a series of (sub)pulses with a few watts—provides the availability of spatial single-mode laser diodes for distance measurements over long measurement ranges, even when point measurement rates are at the level of millions of points per second. Furthermore, the lower peak power of the pulse group maintains a low average power, which better matches the properties of laser diodes, and their lifetime is increased due to lower stress.

[0011] However, this multi-pulse modulation scheme, which includes a pulse sequence of sub-pulses, has several drawbacks. For example, the signal and detector noise is greater than that of a single pulse transmission; the target echo of the pulse sequence is buried in the received and sampled waveform and cannot be identified; and two or more targets hit by the range-measuring beam cannot be easily separated if the distance between the two or more targets hit is less than the distance range associated with the transmission duration of the sub-pulse sequence.

[0012] This invention relates to an optoelectronic distance measurement module configured for use in surveying equipment, such as surveying equipment used for geodetic surveying, construction, or topographic mapping. For example, the surveying equipment may be implemented as a velocimeter, total station, laser profilometer, or laser scanner.

[0013] The distance measurement module includes an emitting unit with a laser diode (e.g., a spatial single-mode laser diode or a low-power fiber laser) and is configured to provide the emission of light pulses via the laser diode or low-power fiber laser, wherein the light pulses form a finite emission code sequence with N code chips. Specifically, the finite emission code sequence is a code sequence with minimal autocorrelation. The elements of the code sequence are called code chips. The distance measurement module also provides processing of a return signal corresponding to the emitted light pulses returned from a target in the environment to generate a measurement receiver output signal. For example, the return signal is provided by an optical receiver unit configured to detect reflected or scattered light pulses from a target, wherein, for example, a detection principle is performed incoherently.

[0014] For example, the receiver electronics of a distance measurement unit (e.g., the receiver electronics comprising a transimpedance amplifier, followed by a subsequent chain of signal amplifiers and filters) transforms the incoming pulse signal (e.g., having an initial Gaussian shape) detected by the photodetector into a corresponding sinusoidal electronic (analog and continuous) output signal ideally having two extrema. More generally, in the optimal case, such as in a noise-free environment and with a light pulse returning from the target with sufficient signal strength, the measuring receiver output signal has the shape of a desired receiver output signal formed by bipolar-shaped components, each of which can be (e.g., uniquely) assigned to the corresponding light pulse. In practice, the components of the electronic receiver output signal have the shape of a heavily attenuated bandpass waveform comprising a dominant maximum and a minimum with a short trailing edge, where the waveform typically has no DC offset. Thus, the dominant portion of the signal component can still be described as exhibiting a bipolar shape.

[0015] The distance measurement module is also configured to provide a reference signal similar to the desired receiver output signal and to encode reference pulses that form a finite reference code sequence with more than N chips. The reference signal is formed from reference signal components that occupy the bipolar shape of the desired receiver output signal components, and each of these components can be (e.g., uniquely) assigned to a corresponding reference pulse in the reference code sequence. For example, the number of chips in the reference code sequence is 3 times, 4 times, or 5 times N. Increasing the length of the reference code sequence has the benefit of increasing the number of coefficients and the emergence of new degrees of freedom for modeling and optimizing processing outputs such as cross-correlated signal outputs.

[0016] The distance measurement module uses a reference signal to perform cross-correlation between the measurement receiver output signal and the reference signal to generate a cross-correlation function, and identifies and times the compressed pulses in the cross-correlation function.

[0017] In other words, instead of emitting light energy as a single pulse, a series of P pulses or pulselets are emitted according to a finite-length transmission code, thus distributing the energy and reducing the peak power of the pulses. At the receiver, a so-called mismatch filter signal is applied to cross-correlate the received signals, effectively compressing the entire energy of the received pulse sequence into a single, dense, and narrow virtual peak with low residual sidelobes. Therefore, a laser pulse sequence arranged accordingly to a suitable digital code of finite length can reproduce the high resolution and low background originally provided by a short, high-power single pulse.

[0018] In one implementation, the transmitted code sequence is a unipolar pulse sequence of fewer than 32 pulses. Many known digital codes that produce minimal aperiodic autocorrelation (e.g., where the absolute value of the correlation is close to zero for all non-zero time offsets or timing clocks) are bipolar, i.e., the code includes zero code elements, positive code elements, and negative code elements. For example, the unipolar pulse sequence corresponds to a code with a bipolar pulse pattern, such as one of Barker codes, composite Barker codes, Ipatiov codes, complementary Golay codes, M-sequences, and Legendre sequences, where the negative or positive values ​​of the code with the bipolar pulse pattern are set to zero (no light pulse), and upon return, the light pulse corresponds to the positive or negative value of the code with the bipolar pulse pattern. For example, the code sequence {1,0,1,0,0,0} is equivalent to transmitting two sub-pulses or small pulses in a six-chip time sequence exactly when the element "1" appears.

[0019] Specifically, additional chips with zero values ​​(no light pulses) are added between adjacent chips of the code with the bipolar pulse pattern to form a transmit code sequence. This is also known as zero-padding. For example, an additional chip with a zero value is added to the right of each chip of the code with the bipolar pulse pattern to form a transmit code sequence. Therefore, the number of chips in the transmit code sequence is an integer multiple of the number of chips in the code with the bipolar pulse pattern.

[0020] In another embodiment, the reference signal is a continuous signal with multiple extrema, wherein each of the reference signal components is a continuous function segment of the reference signal with two extrema. For example, each reference signal component has a sinusoidal or quasi-sinusoidal shape and includes positive and negative values, wherein the amplitude of the reference signal component varies between components.

[0021] In another embodiment, the processing of the return signal is provided in such a way that all components of the desired receiver output signal have the same time signal width, wherein the reference signal is provided in such a way that each of the reference signal components has the same time signal width as the desired receiver output signal component.

[0022] In another embodiment, the reference signal is a discrete signal having multiple positive and negative values, wherein each of the reference signal components includes both positive and negative values. Here, the discrete signal occupies a bipolar (e.g., sinusoidal or sinusoidal) shape of the desired receiver output signal. For example, the positive and negative values ​​have the same absolute value and differ only in their signs.

[0023] In another embodiment, each of the reference signal components consists of two adjacent chips, one of which includes a positive value and the other of which includes a negative value of the corresponding reference signal component.

[0024] It is known that the receiver output signal can be in the form of a purely analog or continuous signal. Alternatively, the analog-to-digital converter samples the sinusoidal or bipolar receiver output signal, where the signal can be resampled to any point in time as long as the Nyquist theorem is satisfied (ignoring small information losses, such as those due to amplitude quantization). Furthermore, the reference function can be treated as a continuous analog signal, and pulse compression can be performed through pseudo-continuous cross-correlation between the two analog and continuous signals.

[0025] However, pulse compression by correlating analog continuous functions requires fairly intensive processing resources. Therefore, for example, in real-time signal processing, an analog-to-digital converter (ADC) is used to sample the receiver output signal at a rate higher than the Nyquist frequency of the signal. The continuous reference signal is also (e.g., mathematically) converted into a sequence of sampled signals, where sampling can be selectively performed from low rates (such as chip rates (fractions of the chip time interval)) to higher rates (such as the ADC rate or even higher (as long as it is synchronized with the ADC sampling rate)). To keep processing power low, it is advantageous to convert the reference signal into a discrete filter sequence comprising a discrete sequence of analog values. For example, the reference signal is chosen to be a discrete signal, where sinusoidal or sinusoidal reference signal components are mathematically sampled at intervals of chip length in the transmitted code sequence (and the reference code sequence) (e.g., in the phases of the maximum and minimum values ​​of the sinusoidal or sinusoidal reference signal components).

[0026] Such a discrete reference sequence simplifies the processing of cross-correlation. For example, for cross-correlation and therefore for pulse compression, time samples of the transmitted code sequence are acquired at times corresponding to the chip interval. This can be guaranteed by an encoding unit that controls the chip interval of the transmitting unit and the sampling point of the analog-to-digital converter at the receiver. Thus, the chip interval and the sampling time are synchronized. For example, the chip and sampling intervals are perfectly synchronized with each other, where the chip interval is typically two to eight times the sampling interval of the ADC.

[0027] In another embodiment, for cross-correlation, multiple time samples (e.g., four time samples) of the measurement receiver output signal are acquired at times corresponding to the chip interval of the reference signal. For example, the encoding unit of the distance measurement module is configured to control the chip interval of the transmitted finite transmission code sequence on the one hand, and control the sampling rate of the analog-to-digital converter of the distance measurement module used to analyze the measurement receiver output signal on the other hand.

[0028] In a simplified implementation, for cross-correlation, time samples of the measurement receiver output signal are acquired at times corresponding to the chip interval of the reference signal, for example, wherein the encoding unit of the distance measurement module is configured to control the chip interval of the transmitted finite transmission code sequence and the sampling points of the analog-to-digital converter of the distance measurement module for analyzing the measurement receiver output signal.

[0029] To derive an accurate distance, a two-step process can be used. The localization of the compressed pulse in the cross-correlation function provides a rough estimate of the target distance. In the second step, time is interpolated using existing techniques, such as those used in so-called waveform digitization (WFD) methods. For example, sampling interpolation algorithms are used to localize the peak of the compressed pulse with sub-picosecond accuracy. Alternatively or additionally, resampling methods are used to process the sampled compressed pulse as a continuous analog signal, producing full resolution with accuracy reduced to the Cramer Rao bound.

[0030] Therefore, in another embodiment, the distance measurement module is configured to identify compressed pulses in the digital representation of the cross-correlation function, and to provide interpolation of the compressed pulses between sampling points of the digital representation of the cross-correlation function when a compressed pulse is detected, so as to time the compressed pulses.

[0031] The so-called Waveform Digital Digitizer (WFD) principle is based on the well-known LiDAR technology, which uses pulse or phase difference coding methods. Its architecture typically includes an optical transmitter and a photodetector unit, where the photodetector unit includes photodiodes, preferably with intrinsic amplification, a low-noise broadband transimpedance amplifier, subsequent amplifier stages, several filters for signal shaping, and a fast sampling analog-to-digital converter (ADC) circuit in the range of several hundred MHz to several GHz, depending on the required time and amplitude resolution. Since the received signal is available in digital format, signal processing and calibration are performed by a processing unit, the core of which can be a field-programmable gate array (FPGA) or a digital signal processor (DSP). Using the WFD principle, echo detection, echo identification, and seamless cross-correlation between a finite-length reference signal and the arriving samples of the received signal are performed digitally and shifted sample by sample in real time. Each detected, identified, and compressed pulse is used to obtain relevant parameters for timing the compressed pulse, such as pulse identifier, signal strength, saturation ratio, signal-to-noise ratio (SNR), time of flight, pulse shape quality indicator, overlap contribution, etc.

[0032] A reference signal can be provided by optimizing a figure-of-fit function. For example, the figure-of-fit function is a scalar function representing the so-called peak-sidelobe ratio (SLR) of the cross-correlation between the received signal and a reference function, where the M amplitude coefficients R = {r1, r2, ..., rM} of the reference function are parameters that must be found. For the so-called dual-target case (where the time interval between two pulse sequences is less than the length of the reference function), an appropriate combination of the received and reference signals is applied to the figure-of-fit function. In principle, an optimized reference signal can be deduced for each pulse interval. However, this may contradict the purpose of simple real-time processing. A compromise is to establish a figure-of-fit function that includes the SLRs of different received dual-echo signals associated with different separated targets. In other words, the figure-of-fit function is a weighted sum of the SLRs of different compressed dual-echo signals. The final amplitude coefficients R are calculated using numerical optimization methods.

[0033] In another embodiment, a reference signal is provided by optimizing a figure-of-fit function, wherein the figure-of-fit function is a weighted sum of functions representing the sidelobe ratios of the cross-correlation between a common reference function and the output signals of different candidate receivers, the output signals of which are associated with desired receiver output signals for different double echoes of the received optical pulse, wherein the different double echoes differ from each other by different pulse intervals between the echoes of the received optical pulse, and the common reference function is associated with one of the different pulse intervals.

[0034] Especially in the case of scanning survey equipment, a moving laser beam can hit completely different distances from one emission code sequence to the next. Therefore, the sequence order of emission and return of sequentially emitted emission code sequences can be interchanged or overlapped, which requires a way to provide pulse discrimination and identification (the so-called ambiguity problem).

[0035] In existing technologies, ambiguity resolution can be achieved by transmitting different multi-pulse sequences. For example, interval or pulse-width modulation is known, where the spacing between and within pulse groups varies. In the simplest case, a single pulse is transmitted using interval modulation, or in more robust methods, double or triple pulses using distance modulation of sub-pulses are used. However, these methods generally still cannot separate multiple targets that are closer together than the length of the pulse sequence. The patterns of overlapping multi-pulse sequences are disrupted, and demodulation becomes erroneous. Furthermore, the signal-to-noise ratio (SNR) can be reduced by additive electronic and optical noise.

[0036] As described above, improved ambiguity resolution can be achieved by using unipolar transmit coding and bipolar sinusoidal receiver output signals, and by using a designed bipolar sinusoidal reference signal to compress reduced sidelobes beyond the main peak.

[0037] In another embodiment, the distance measurement module is configured to coordinate the emission of optical pulses to generate different finite emission code sequences, each having N chips. Each of the different finite emission code sequences is associated with a corresponding desired receiver output signal formed by a desired receiver output signal component of a bipolar shape (as described above, bipolar or sinusoidal), wherein each of the desired receiver output signal components can be (e.g., uniquely) assigned to a corresponding optical pulse. The distance measurement module is also configured to provide different reference signals, each of which is similar to a different one of the desired receiver output signals, and to encode reference pulses forming a finite reference code sequence having a number of chips greater than N.

[0038] Similar to the reference signal described above, each of the reference signals is formed by reference signal components, which occupy the bipolar shape of the desired receiver output signal components, and each of the reference signal components can be assigned to a corresponding reference pulse of the reference code sequence.

[0039] Cross-correlation is performed in parallel between the measurement receiver output signal and each of the different reference signals to generate different cross-correlation functions associated with each of the different reference signals (simultaneously). In each of the cross-correlation functions (which are created in parallel), compressed pulses are identified and timed, wherein the distance measurement module is configured to associate each of the compressed pulses with a corresponding one of a different finite transmission code sequence.

[0040] Therefore, for example, the distance measurement module can be configured to provide sequential transmission modes of different finite transmission code sequences, and to compare the sequential transmission modes with the timing of the compressed pulses associated with each cross-correlation function to provide range ambiguity correction.

[0041] The present invention also relates to a surveying device for performing three-dimensional spatial measurements of an environment using an optical distance measuring beam, wherein the surveying device is implemented as a velocimeter, total station, laser profilometer, or laser scanner, and wherein the surveying device includes an optoelectronic distance measuring module according to one of the above embodiments. Attached Figure Description

[0042] The photoelectric distance measurement module and surveying device according to different aspects of the invention will be described or explained in more detail below by way of example only, with reference to the working examples schematically shown in the accompanying drawings. In the drawings, the same elements are labeled with the same reference numerals. The described embodiments are generally not shown to scale and should not be construed as limiting the invention. Specifically,

[0043] Figure 1 Exemplary implementations of surveying equipment;

[0044] Figure 2 Exemplary applications of surveying equipment used to stake out different measurement locations on the ground;

[0045] Figure 3 Another exemplary application of surveying equipment for scanning the environment;

[0046] Figure 4 : Illustrative implementations of different transmission code sequences;

[0047] Figure 5 A block diagram of an exemplary implementation of a time-of-flight electronic distance measurement module based on the WFD principle;

[0048] Figure 6 A schematic implementation of the transmission code sequence, the corresponding measurement receiver output signal, and the reference signal;

[0049] Figure 7 : Figure 6 The amplification section of the reference signal;

[0050] Figure 8 A schematic diagram of the cross-correlation function between the output signal of the measurement receiver and the reference signal;

[0051] Figure 9 In the case of two targets, a schematic diagram of the cross-correlation function between the output signal of the measurement receiver and the reference signal is provided.

[0052] Figure 10 A schematic diagram of the discrete reference signal is derived.

[0053] Figure 11 A schematic diagram of the synchronization chip interval and sampling time;

[0054] Figure 12 Another schematic diagram for deriving the discrete reference signal. Detailed Implementation

[0055] Figure 1 An exemplary embodiment of a surveying device 1 (e.g., a velocimeter or total station) is illustrated. The surveying device includes a base 2, which can be mounted on a holding device, for example, in the form of a tripod bracket (not shown). A support structure 3 is mounted on the base 2 such that the support structure 3 can rotate about a vertical axis 4, wherein the support structure 3 holds an aiming component 5, which can rotate about a horizontal axis 6. Both the support structure 3 and the aiming component 5 can be rotated electrically, for example by means of an electric shaft 7, wherein the orientation of the support structure 3 and the aiming component 5 can be determined by a corresponding angle encoder (not shown).

[0056] The aiming component 5 is configured to emit a distance measuring beam toward a target object along the aiming axis 8. For example, the objective lens 9 is identical for both the transmitting and receiving channels of the distance measuring beam. The aiming component 5 houses an optoelectronic distance meter configured to determine the distance to a single target point aimed at by the aiming axis 8 based on at least a portion of the distance measuring beam returning from the target. For example, certain portions of the optoelectronic distance meter (e.g., the beam source) may also be arranged in the support structure 3, wherein an optical fiber-based waveguide system connects the elements integrated in the support structure to the aiming component 5 via a shaft 7.

[0057] Typically, surveying equipment also includes additional sensors (such as a so-called overview camera 10) to provide the user with images of the scene to be measured and for an automatic coarse target search function. For example, the coarse target search function provides the identification of cooperating targets (such as retroreflectors), allowing the surveying equipment to automatically move the aiming component 5 to point at the cooperating target. Typically, another so-called fine aiming camera (e.g., coaxially arranged with the aiming axis 8) is used to precisely point the aiming component 5, and thus the aiming axis 8, at the cooperating target (so-called locking the beam onto the cooperating target), and to provide automatic tracking of the cooperating target. For example, the objective lens 9 is a multifunctional optical component, such as imaging optics for the fine aiming camera, transmitting and receiving optics for the distance measurement unit, and optionally, a front lens for a visual telescope used to manually aim the aiming axis at the target to be measured.

[0058] The surveying device 1 may also include an additional sensor 11 (e.g., a distance imaging sensor) to provide depth images of the scene or to provide gesture control of the surveying device 1.

[0059] Typically, surveying equipment is configured to move a distance measuring beam around two axes (e.g., horizontal axis 6 and vertical axis 4) in a given example, which are usually orthogonal to each other (so-called dual-axis device for moving the distance measuring beam within a polar coordinate system frame).

[0060] For example, similar to the setup described above, industrial laser scanners typically include a base and a support, wherein the support is mounted on the base such that it can rotate about a first axis of rotation (often referred to as the slow axis of rotation). A rotating body is mounted on the support such that it can rotate about a second axis of rotation substantially orthogonal to the first axis of rotation, often referred to as the fast axis of rotation because the rotation of the rotating body is typically faster than that of the support. For example, an optical distance measuring device arranged in the support is configured to acquire distance measurement data. A transmitter is configured to transmit a distance measuring beam toward the target along the target axis, and a receiver is configured to detect at least a portion of the distance measuring radiation returning from the target as an echo, for example by means of an objective lens and a photosensitive sensor configured to measure the time-of-flight of the laser pulse.

[0061] Distance measuring radiation is emitted from a support (e.g., parallel or coaxial with the axis of rapid rotation) and deflected by a mirror surface (e.g., a plane mirror or a parabolic mirror) of the rotating body, causing the distance measuring radiation to rotate about the axis of rapid rotation. The amount of rotation of the support and the rotating body can be determined by means of an angle encoder configured to acquire angle data, wherein a laser scanner is configured to correlate the angle data with corresponding distance measurement data of the distance measuring device. Thus, the entire environment around the laser scanner can be measured essentially in space. Distance measuring units typically stand out due to their high ranging accuracy (millimeter or sub-millimeter range) and their high speed of at least one million points per second. In the prior art, such coordinate measuring instruments are typically equipped with complex and expensive laser modules. Utilizing the modulation and signal processing scheme of the present invention, relatively high performance can be achieved, for example, using a low-power single-mode laser diode or a simple single-stage fiber laser.

[0062] Figure 2 An exemplary use case of surveying equipment 1 is shown, wherein surveying equipment 1 is configured as a total station, wherein surveying equipment 1 and surveying rod 12 are used to stake out different measurement locations 13 on the ground.

[0063] The surveying pole 12 has a rigid pole-shaped body with an indicator tip for contacting the measurement point 13 on the ground. The pole 12 includes a cooperative target 14 (e.g., a retroreflector device) as a cooperative location-given device for determining the available position of a reference location at the pole 12, wherein the cooperative target 14 is positioned at a known location relative to the tip of the pole. The total station 1 is used to determine the coordinates of the cooperative target 14, which then provides the coordinates of the measurement point 13 given the known position relative to the tip and the known orientation of the pole.

[0064] The total station 1 can also be used for scanning applications, such as recording point clouds of construction sites, streets, or bridges. During the scanning process, the distance measuring beam sweeps across the object surface by moving the total station's aiming component about two axes. This rapid rotational movement is achieved through the aiming component, which is significantly lighter than the supporting structure, compared to laser scanners. The photoelectric distance meter is configured to record distances and surface point clouds of natural or man-made objects with very high ranging accuracy (millimeters or sub-millimeters), along with angular information related to the orientation of the aiming component.

[0065] Because tachometers or total stations are mobile (e.g., used in the field), battery-powered, and small in size, integrated laser sources (e.g., laser diodes) with low power and high efficiency are advantageous. In the past, achievable measurement rates were typically in the range of 10,000 to 100,000 points per second. Using the modulation and signal processing scheme of this invention, a measurement point rate of at least one million points per second can be achieved, for example, by utilizing a low-power single-mode laser diode.

[0066] Figure 3 An exemplary use case of surveying device 1 is illustrated, wherein surveying device 1 is configured as a scanning total station, wherein a street with adjacent buildings is measured. Here, surveying device 1 causes distance measuring beam 15 to scan around two axes 4, 6 to scan the environment to generate a 3D point cloud representing the environment. Surveying device 1 includes a distance measuring module configured for distance measurement on natural (diffuse) targets, such as arbitrary points on the walls of buildings.

[0067] Figure 4 The middle and bottom figures schematically illustrate different emission code sequences 16 and 16' according to time t when using a laser diode as the radiation source in a distance measurement module. In the example given in the middle figure, emission code sequence 16 includes 12 chips 17 and 4 emitted laser pulses 18. The emission code sequence 16 at the bottom is an extended version of the emission code sequence in the middle figure, where so-called zero-padding has been applied. Adding additional chips with zero values ​​(no light pulses) between adjacent chips of the initial code 16 results in an emission code sequence 16' with 23 chips. For example, zero-padding provides the advantage of encoding laser pulses by a code sequence consisting of multiple equal values ​​(e.g., multiple logic 1s) one after another (see below).

[0068] For example, the transmit code sequence corresponds to a known finite code sequence with minimal autocorrelation, where the negative value 19 of the known binary code sequence of finite length is set to zero (no light pulse). This is illustrated by the top and middle of the figure, where the initial bipolar code sequence 20 with the value 1-11-10010-1001 shown at the top of the figure is used to generate the transmit code sequence 16 in the middle of the figure by ignoring the negative value 19.

[0069] Many schemes are known to convert bipolar pulse codes (e.g., via Manchester encoding) into sequences of monopolar pulses. Since LiDAR devices preferably include laser sources with modulation schemes featuring energy pulses, the modulation is of an incoherent type. Therefore, the applied modulation is monopolar, where the emitted pulses or sub-pulses represent logic 1 and no pulses within the timing chip represent logic 0. Thus, when using known binary code sequences of finite length with minimal autocorrelation, it is necessary to convert the bipolar sequence into a monopolar sequence.

[0070] Laser diodes or miniature seeded fiber amplifiers can also emit pulses of varying energies or amplitudes, allowing the use of monopole or unipolar code sequences with any number of poles. However, it generally becomes easier to achieve the highest quality ranging accuracy when keeping the laser pulse energy or peak power constant. Therefore, using a digital binary encoding scheme with minimal autocorrelation can be considered beneficial, where the original bipolar code is converted into a unipolar inverse return-to-zero (RZ) encoding mode.

[0071] Laser diodes are miniaturized and inexpensive radiation sources for distance measurement units, but they have limited peak power. To avoid this limitation, instead of single pulses, pulse groups consisting of uniformly or spaced encoded pulse sequences are transmitted. This ensures higher average transmission power. For example, burst sequences are known to include regular pulse trains with interruptions between pulse trains, as well as random pulse trains encoded based on m-sequences or King's codes.

[0072] Compared to single-pulse measurements, long pulse sequences add additive noise to each pulse, which reduces the signal-to-noise ratio (SNR). Therefore, it is advantageous to use short pulse sequences to maintain a relatively high SNR. Another benefit of short pulse sequences appears in the case of scanning survey equipment, where the laser spot is scanned (moved) across a target surface. The longer the coded sequence, the more blurred the measurement point becomes. For example, pulse sequences between 5 pulses (P=5) and 32 pulses (P≤32) in the code sequence are used. Short pulse sequences are also advantageous for multi-return scenarios, such as when the laser beam hits the edge of an object, resulting in radiation being received from two target surfaces (the so-called mixed-pixel case). Undamaged target separation is achieved when the target spacing is longer than the length of the emitted pulse sequence. Short pulse sequences are conceivable, for example, using Barker codes, Walsh codes, or Ipatov codes.

[0073] On-off keying (OOK) is a type of amplitude shift keying (ASK). Different methods are known for designing the switch {1, 0} encoded envelope of the transmitted carrier, whose echo can be intensity or envelope detected, digitized, and incoherently processed to extract and calculate the distance to a target surface. Examples are simply positive elements of the transmitted code (+1→1, -1→0) or Manchester encoding (+1→1 0, -1→0 1). Manchester encoding doubles the code length, which adds additional noise compared to using only positive (or negative) elements.

[0074] For example, using binary Barker codes of length 13: {1,1,1,1,1,-1,-1,1,1,-1,1,-1,1,-1,1}, {1,-1,1,-1,1,1,-1,-1,1,1,1,1,1}, {-1,1,-1,1,-1,-1,1,1,-1,-1,-1,-1}, {-1,-1,-1,-1,-1,-1,1,1,-1,-1,1,-1,1,-1,1,-1}. When only the positive elements of the code (+1→1, -1→0) are used, the transmission code sequence becomes: {1,1,1,1,1,0,0,1,1,0,1,0,1},{1,0,1,0,1,1,0,0,1,1,1,1,1},{0,1,0,1,0,0,1,1,0,0,0,0,0},{0,0,0,0,0,1,1,0,0,1,0,1,0,1,0}, that is, a code sequence with 9 or 4 pulses.

[0075] As another example, to provide a more uniform energy distribution, a shorter binary Barker code of length 11 can be used: {1,1,1,-1,-1,-1,1,-1,-1,1,-1},{1,-1,1,1,-1,1,1,-1,-1,-1},{-1,1,-1,-1,1,1,-1,-1,-1,1,1,1},{-1,-1,-1,-1,1,1,1,-1,1,-1,1,1}. When only positive elements (+1→1, -1→0) are used again, these codes produce the following emission code sequences: {1,1,1,0,0,0,1,0,0,1,0},{1,0,1,1,0,1,1,1,0,0,0},{0,1,0,0,1,0,0,0,1,1,1},{0,0,0,1,1,1,0,1,1,0,1}.

[0076] The simple unipolar coding of the Barker 11 code shows a fairly uniform energy distribution for all four codes, and the sequence consists of five or six pulses, which is a good trade-off when more than one code is needed, such as to handle distance ambiguity.

[0077] Barker codes allow for short pulse sequences, which have the advantage of maintaining a high signal-to-noise ratio. However, with pulse compression, the peak-to-sidelobe ratio quality decreases. Sidelobe peaks become larger, and crosstalk between overlapping pulse sequences also increases. Nevertheless, short pulse sequences can represent a beneficial trade-off.

[0078] Other codes, unlike Barker codes, can also be short-length, such as so-called complementary single-pole code pairs. In this case, two complementary codes are transmitted one after the other to the target. Unless at time shift zero, the cross-correlation between this pair of transmitted pulses and the pair of reference functions is sign-complementary. The sum of this cross-correlation produces a strongly compressed pulse and a high peak-to-sidelobe ratio. However, when the two targets are separated by a distance smaller than the length of the transmitted sequence, the echoes partially overlap, and the advantage of a high peak-to-sidelobe ratio is lost.

[0079] Dividing the main pulse into a sequence of sub-pulses increases the signal-to-noise ratio without increasing the peak power of the laser diode (e.g., an AlGaInP / GaInP semiconductor laser diode emitting at a wavelength of 660 nm). Pulse compression methods using reference functions primarily transmit the received sub-pulses into a single pulse with the width of a single sub-pulse, thus simplifying further processing of the compressed signal. Reference functions can also account for so-called mixed-pixel cases or distance ambiguity problems, where dual-target discrimination is better compared to simple reference functions or burst modulation schemes.

[0080] By applying zero-padding to the coding sequence, all ON-chips have the same length, and thus each laser pulse has the same short length of a few hundred picoseconds. This is beneficial for precision LiDAR (optical detection and ranging) systems, for example, in terms of time constraints of photons, spatial uniformity of the beam in the far field, and constant polarization distribution.

[0081] For example, the Barker 13 code is transformed in a monopole and zero-fill sequence. Zero-fill provides that all laser pulses with the indicator value {1} have the same length, typically a few hundred ps. The monopole and zero-fill Barker 13 code takes the form: {1 0 10 1 0 1 0 1 0 0 0 0 1 0 1 0 0 0 1 0 0 0 1 0}. This code consists of 26 chips, i.e., the length of the code is N=26, and 9 pulses, also called subpulses P=9.

[0082] Although the nine emitted and reflected laser pulses are sharp peaks in the time domain, and the electrical bandpass filter is used for noise suppression and to achieve the Nyquist frequency associated with the sampler (ADC), the waveforms of the reflected subpulses typically have a bipolar shape (see below).

[0083] Based on the analog bandwidth of the receiver of the waveform digitizer and analog-to-digital converter (ADC), the preferred chip length is designed such that its time interval is longer than the receiver's dominant bipolar response. The chip length is given by the transmitter's modulation interval and corresponds to the time interval between two code elements (e.g., two laser pulses). For example, for an ADC with 1 GS / s, the receiver bandwidth (BW) is chosen to be 300 MHz. Therefore, a chip length of 3 / (2*BW) = 5 nsec is used. In this case, the ADC acquires five samples per received subpulse. The time length of the encoded signal is 5 ns * N = 130 ns, so the pulse train in the air will have a length of 39 m. For scanning survey equipment, the lateral spatial resolution is reduced by 130 ns of transmission duration. When the laser beam is deflected by a 100 Hz rotation and the target distance is 100 m, the lateral spatial resolution widens to 8 mm, which is acceptable for most survey applications. Therefore, modulation based on Barker codes or so-called composite Barker codes with shorter lengths is a suitable modulation scheme. Composite Barker codes are created by nesting inner Barker codes within outer Barker codes.

[0084] Figure 5 A block diagram of an exemplary implementation of a time-of-flight electronic distance measurement module based on the so-called waveform digitizer (WFD) principle is shown.

[0085] The functional processing of all modules is controlled by a control unit 21 with a power supply 22, wherein time processing is defined by a central crystal oscillator of type TCXO (temperature compensated crystal oscillator) or OCXO (oven controlled crystal oscillator). The central crystal oscillator is also used as a ppm-accuracy time scale for measuring time of flight. A frequency generator 23 generates a corresponding signal that, on the one hand, synchronizes the photoelectric transmitter 24, and on the other hand, synchronizes the photoelectric receiver module 25 and the WFD unit 26.

[0086] The emitted laser signal operates continuously or simultaneously through the internal optical path 27 and the external optical path 28. The external optical path is pointed at the target object to be measured (the pulses of the external optical path are also referred to as stop pulses). The internal optical path 27 is used in a known manner to transmit a so-called start pulse sequence to calibrate the absolute distance. For example, in the case of simultaneously measuring the internal optical path 27 and the external optical path 28, the combined received signal is irradiated onto the receiver of the photodetector unit 25. At short target distances, the start pulse and the stop pulse may partially overlap. However, by using pulse compression as described above, the two signal sequences can be separated, and the time of flight from the start sequence and the stop sequence can be measured accurately.

[0087] The opto-receiver unit 25 includes, for example, a low-noise broadband transimpedance amplifier and a subsequent amplifier stage to amplify the signal as linearly as possible, wherein the amplified signal is then fed to a filter bank and a fast analog-to-digital converter (ADC) circuit with suitable time and amplitude resolution. High-precision (e.g., sub-millimeter) WFD-based time-of-flight measurements require typical ADC sampling rates of at least 0.5 GHz up to several GHz. In the case of a 1 GHz sampling ADC, the received signal has a nanosecond sampling interval, which is one-fifth of the 5 nanosecond coded chip length at the transmitter.

[0088] WFD processing unit 26 is configured to perform pulse compression by digitally cross-correlating the received and sampled return signal from the target with a corresponding reference function of the transmitted code sequence. This reference function is digitally stored in the memory of WFD processing unit 26, while, for example, a pseudo-continuous reference function is stored in a lookup table containing amplitude sequences of short time intervals corresponding to the sampling intervals of the analog-to-digital converter. Therefore, the pseudo-continuous reference function has a higher quantity density than the chip length when encoded on the transmitter side. Because the sampling or quantity density of the reference function is higher than the chip length, it is called pseudo-continuous. The cross-correlation between the finite-length sequence of the reference function and the arriving samples of the received signal is performed digitally in real time, sample by sample, for example, by a field-programmable gate array (FPGA) or digital signal processing unit (DSP) as part of WFD processing unit 26. The time series of the cross-correlation result is a pseudo-continuous output signal containing compressed pulse information. In other words, compressed pulses are provided in the digital representation of the cross-correlation function. If the returned pulse sequence is included in this so-called "compressed output signal," then an isolated pulse, the width of a single small pulse of the received encoded pulse sequence, will appear.

[0089] For example, the main part of the real-time signal processing module resides on an FPGA or DSP. A peak detection and peak recognition module follows the signal compression module. This module is configured to stream the compressed output signal. If a compressed peak is detected, an recognition step is performed, for example, by examining the desired pulse shape. Simultaneously, a coarse distance is derived using the resolution of a pseudo-continuous cross-correlation function. In the next step, the region of interest around the compressed pulse is fed to a fine distance estimation module. It provides an interpolation of the compressed pulse between time samples of the cross-correlation function, typically with an interpolation density of 10,000. Thus, the time resolution is in the range down to femtoseconds.

[0090] WFD processing unit 26 is configured to perform time interpolation of the compressed output signal with accuracy in the micrometer range, for example, by applying a sampling interpolation algorithm to locate the peak of the received signal with sub-picosecond accuracy. Alternatively or additionally, WFD unit 26 may implement a resampling method to process the sampled received signal as a continuous signal, which produces full resolution and accuracy down to the Cramero bound.

[0091] The results and other data are output via the interface of the control unit 21, in which the control unit calculates the final distance and / or takes into account any calibrations and corrections, such as range wander, temperature or air pressure effects.

[0092] Figure 6The diagram illustrates how the transmit code sequence 16, the corresponding measurement receiver output signal 29, and the reference signal 30 are used to perform a cross-correlation between the measurement receiver output signal 29 and the reference signal 30 to generate a cross-correlation function.

[0093] Although the transmitted signal 16 is unipolar, the photodetector module (e.g., including the one referenced above) Figure 5 The described amplifier and filter stages generate a bipolar (e.g., sinusoidal or sinusoidal) receiver output signal 29. The waveforms of the sub-pulses have a shape close to a critically damped oscillation, for example, including two or three peaks. Therefore, even in an ideal case, the receiver output signal 29 comprises a combination of multiple waveforms, each having a primary main oscillation component 31 and a secondary main oscillation component 32, and one or more additional oscillation components 34. The sum of the primary main oscillation component 31 and the secondary main oscillation component 32 is associated with almost 100% of the energy of the incoming pulse. The primary main oscillation component 31 and the secondary main oscillation component 32 represent the binary pulse response of the photodetector channel. Additionally, electronic noise 41 degrades the signal-to-noise ratio of the received pulses in the receiver output signal 29.

[0094] The signal shape differs from previously known signals used for applying pulse compression methods. Pulse compression between the unipolar transmission sequence and the received bipolar waveform remains unknown. Therefore, a reference signal 30 for cross-correlation is needed, which takes into account the specific shape of the receiver output signal 29.

[0095] Figure 7 It shows Figure 6 The amplification section of the reference signal 30 (this section is composed of) Figure 6 (Indicated by the dashed rectangle in the figure). The reference signal is noise-free, where, for example, its oscillations are generated entirely by a bipolar waveform (also called the reference signal component). For example, each chip contains such a bipolar oscillation. Alternatively, as shown, each bipolar waveform covers two chips, for example, such that the waveform component begins at the beginning of all odd-numbered chips.

[0096] Each bipolar waveform (reference signal component) has its own well-defined amplitude. These so-called principal amplitudes are real numbers and can be positive or negative. In the accompanying figure, two bipolar waveforms 33 with positive amplitudes and one waveform 33' with negative amplitudes are highlighted by thick lines. The waveform 33' with negative amplitude is a bipolar oscillation (reference signal component) that begins with a negative peak, while the waveform 33 with positive amplitude is a bipolar oscillation (reference signal component) that begins with a positive peak. The oscillating component associated with the initial peak of the waveform is also referred to as the (positive or negative) principal oscillation component, which is similar to the receiver output signal 29 ( Figure 6The oscillation component disappears before the next (positive or negative) main oscillation component appears in the next waveform (reference signal component) 33, 33'. The oscillation component associated with the peak after the initial peak of the waveform is also referred to as the (positive or negative) secondary oscillation component. According to the invention, the shape of the bipolar waveform 33, 33' of the reference signal 30, which includes the (positive or negative) main oscillation component and the secondary oscillation component, corresponds to the main oscillation component 31 and the secondary main oscillation component 32. Figure 6 The shape of the waveform of the receiver output signal 29.

[0097] In summary, the two corresponding primary and secondary oscillation components form the so-called reference signal components 33 and 33', which occupy the bipolar shape of the desired receiver output signal component associated with a single emitted optical pulse. For example, the three reference signal components of reference signal 30 are highlighted in bold in the figure, namely, two reference signal components 33 including (i.e., starting from) the positive primary oscillation component and one reference signal component 33' including (i.e., starting from) the negative primary oscillation component.

[0098] For a transmit code sequence with N chips, the length of the reference signal is greater than N because the fidelity of pulse compression increases. In signal processing terminology, the reference function can be considered as a mismatch filter. The mismatch filter is preferably designed to have a length of 3N or 5N, but other lengths are also possible, such as 1.8N or 2.2N.

[0099] By measuring the output signal 29 of the receiver Figure 6 Cross-correlation is performed between the signal and the reference signal 30, and good pulse compression is also achieved for the output signal of the bipolar desired receiver. Therefore, the cross-correlation yields the following: Figure 8 The cross-correlation function shown schematically includes compressed pulses 35 that can be identified and timed by the distance measurement module.

[0100] Typically, better sidelobe suppression is achieved by using a mismatch filter (=reference signal) of length greater than N (e.g., 3N or 5N). N is the length of the transmitted coded pulse sequence or equivalently the number of chips. Typically, the mismatch filter is many times the length of the original code (of length N), and they include precise analog amplitudes to produce sufficient sidelobe suppression.

[0101] Orthogonal codes, such as Walsh codes, can also be used. These codes are designed to have little or no cross-correlation (meaning the correlation value is close to zero for all time shifts other than zero), making them distinguishable from each other. This can be advantageous, for example, when two or more surveying devices are operating at the same location. When a range measurement module receives a modulated pulse from another range measurement module, the demodulator is able to distinguish its own pulse code from the other pulse code while simultaneously receiving the pulse reflected at the target from its own laser diode.

[0102] For example, using a single-pole zero-filled Barker code 13a with a length of N = 26 chips. k ={1 0 1 0 1 0 1 0 1 00 0 0 0 1 0 1 0 0 0 1 0 0 0 1 0} gives the value at t TOF The flight time of the expected analog signal reflected from the target sig (t):

[0103]

[0104] The received analog and continuous signals include those encoded at time intervals t. c The emitted laser pulse is a series of bipolar pulselets, where t c This indicates the chip length. At the sampling time t given by the ADC... n At this point, the received analog and continuous signals are sampled at a high sampling rate.

[0105] Reference function s3N ref (t) has a length M = 3 * N and has the following form:

[0106]

[0107] In addition, the reference signal s3N ref (t) is represented in the manner of sampling, for example, with a sampling time t corresponding to the time interval of the ADC. n Its generation series s3N ref (t n However, any other pseudo-continuous representation is possible.

[0108] As mentioned above, the reference function s3N ref (t) is configured to have the same waveform per chip as the desired sub-pulse of the electronic receiver output signal.

[0109] The reflected and received analog signal S sig The cross-correlation between (t), or more precisely, the pseudo-continuous signal s sig (t n) and pseudo-continuous reference function s3N ref (t n The cross-correlation between them generates a pulse compression signal Cor3N(τ):

[0110]

[0111] Where τ represents the time delay or time shift between the two signals, which is the (new) time axis of the received compressed signal. N o It is the number of samples along the time axis that at least covers filter s3N. ref (t n The length of the compression peak is determined. During the measurement operation, if a compression peak occurs, the compressed output signal Cor3N(τ) is continuously checked along the time axis τ. If a compression peak is detected, an identification process is performed, and the time τ of the sample closest to the compression peak is stored until its offset corresponds to the time of flight (TOF). After this coarse TOF estimation, a fine TOF estimation is performed, for example by interpolating the waveform between time samples to extract the true maximum value of the compression peak.

[0112] The width of the main peak of the compressed pulse 35 corresponds to the chip time, where the compressed pulse 35 behaves similarly to when only a single strong laser pulse is received. As a result, the sensitivity and time resolution become closer to those of a range measurement module that uses a single strong laser pulse emitted from a medium to a high-power fiber laser, instead of using a laser pulse from a laser diode with weaker pulse emission. This full benefit is effective for single echo cases or when two targets are separated by more than two chips. Other mismatch filters exist that also work with closely spaced targets, where such filters produce sufficient quality in terms of peak-to-sidelobe ratio for multiple target returns or echoes separated by time intervals shorter than the length of the reference function. However, to utilize a good peak-to-sidelobe ratio, the design process for establishing the reference function advantageously includes multiple target returns.

[0113] The search for an optimized mismatch filter for pulse compression is illustrated by the following example of a mismatch filter for a length of M = 3 * N chips. The filter elements of the reference function are R = {r1, r2, ..., rM}, and the filter elements of the transmitted code sequence are S = {0, 0, ..., 0, R, 0, 0, ..., 0,}, where the length of S is increased by padding with 0s to match the length M = 3N. Encoding R and decoding S primarily begin with the chip sequence, where pseudo-continuous waveforms are applied in later design steps.

[0114] The filter coefficients can be found by maximizing the peak-to-sidelobe ratio (PSLR) of the cross-correlation function Cor(τ) over all delays within the length of the reference function R, where for τpeak + / -0.5t c τ outside of

[0115]

[0116] Among them, the cross-correlation Cor(τ) will be delayed by τ peak The peak value is reached at 0.

[0117] It can be done in integer step size t c The search for the maximum value of the peak-to-sidelobe ratio (PSLR) is completed in the middle, where t c This represents the chip length, therefore the search range is from k = -1N to +1N. Similarly, Cor(τ) peak The calculation of the filter coefficients R = {r1, r2, ..., rM} can be completed at integer points, which speeds up the search for the filter coefficients. Optimization of filter elements is a cumbersome process, and stochastic optimization techniques such as Monte Carlo simulations can be used with a sufficient number of selected starting vectors Ro.

[0118] For example, when applying Barker code 11, the initial vector of Ro is only bipolar Barker code 11. The initial values ​​of the reference signal become R = {r1 = 0, 0, ..., rN = 0, Barker code 11, r2N+1 = 0, 0, ..., r3N = 0}. According to the equation for PSLR above, Cor(τ) peak The signal can be calculated by employing the unipolar encoding described above and correlating it with the bipolar Barker code 11. In the next step, the signal is shifted or stepped to the left piece by piece, and the correlation function Cor(τ) is set for each step 0. k At this point, an initial guess is obtained for the coefficients rN, rN-1, ..., r1. Then, the coefficients r2N+1, r2N+2, ..., r3N are evaluated by stepping to the right side chip by chip until all coefficients of R are determined. The amplitude value rk can take any positive or negative value. This intermediate reference function may not be optimal because sidelobes are generated for shifts beyond the considered delay. Another optimization step can be applied, for example by a stochastic optimization technique, where all coefficients are deconstrained and varied in this loop to achieve the amplitude of the optimal reference function. After this optimization step, the final continuous reference function is established by taking the bipolar waveform of the desired receiver output signal component and multiplying it by the amplitude R calculated above. The method (recipe) given here indicates that the longer the reference function, the better the PSLR of the pulse compression.

[0119] Another way to define a suitable mismatch filter (e.g., a weaker but sometimes preferred, simpler approach) is to minimize the integrated sidelobe energy ISLR:

[0120]

[0121] Among them, the cross-correlation Cor(τ) will be delayed by τ peak The peak value is reached again at 0.

[0122] Further simplification can be achieved by minimizing the merit function ISLRr:

[0123]

[0124] The minimum value of ISLRr can be searched again in integer steps t. c Completed, therefore the search range is from k = -1N to +1N. Similarly, Cor(τ) peak The calculation of ) can be completed at integer points, which speeds up the search for filter coefficients R = {r1, r2, ..., rM}.

[0125] For example, a short monopole pulse sequence comprising a maximum of ten pulses is used, wherein the length of the pulse sequence is no longer than 4 * 10 = N chips. All laser pulses have the same length, typically covering less than one chip, to keep photons confined within well-defined short time intervals. The received pulses are converted into bipolar electrical pulses, and a mismatch filter of length 3N is used as a reference, wherein pulse compression is applied to the bipolar received pulse shape (waveform). Optionally, at least two different transmit codes are applied to resolve range ambiguity.

[0126] In the so-called dual-target case with strong pulse overlap, the received signal consists of two echoes (e.g., separated by fewer than three chips). For example, for a chip length of N = 26 chips and a target separated by 3.3N - 3N = 0.1N = 2.6 chips, mathematically, the expected continuous and bipolar signal S can be... sig (t) and reference function s3 ref (t) is given as a continuous waveform:

[0127]

[0128] The function “wav” represents the bipolar or sinusoidal impulse response of the receiver in front of the ADC, wherein the same impulse response is included in the reference signal used for pulse compression.

[0129] Pulse compression Cor3(τ) is accomplished through convolution between two signals and can be written as:

[0130]

[0131] Its generation Figure 9The cross-correlation function illustrated schematically with respect to a time delay τ includes two compressed pulses 36, 37 that can be identified and timed by the distance measurement module. Therefore, although the two targets are within the length of the transmitted pulse sequence, they can be separated.

[0132] The reference function s3 used for the dual-objective case ref (t) can be the same as in the case of a single target. However, as mentioned above, in the case of overlapping dual-target returns (also known as the mixed-pixel case), an improved reference function s2Tgt with increased peak-to-sidelobe ratio (SLR) and compressed signal can also be provided. ref (t). To derive the optimized reference function s2Tgt ref (t) , necessitates a new merit function. One approach is to maximize the merit function, which incorporates the SLRs of different received dual-echo signals associated with different separated targets. For example, this merit function could be a common reference function s²Tgt for different dual-echo signals and unknown parameters R = {r1, r2, ..., rM}. ref The weighted sum of the cross-correlation between (t) and the SLR. The output amplitude coefficient R is optimized by maximizing the merit function.

[0133] Pulse compression, performed in real time by correlating a continuous reference function, requires fairly intensive processing resources, even though the sampling corresponds to the rate of the ADC (so-called pseudo-continuous reference function). One possibility that allows for efficient, high-speed real-time data processing (e.g., enabling onboard real-time data processing on FPGAs with moderate processing power) is to transform the bipolar continuous reference function into a discrete filter sequence with spare samples or coefficients.

[0134] For each chip, a continuous bipolar reference function occupies (“matches”) the pulse waveform of the receiver electronics, where the waveform is a response to the detected laser pulse. For example, to generate a discrete reference signal, the chip-based waveform is converted into two digital values, one at a maximum of 38 and close to the minimum of 39 of the analog continuous waveform, such as... Figure 10 As shown.

[0135] For example, a discrete reference sequence adjusted to a sinusoidal waveform is obtained by sampling at intervals of chip length in the phases of the maximum and minimum values ​​of the sinus, where the primary oscillation component 31 is sampled at its extreme points, and the secondary primary oscillation component 32 is sampled near its extreme points. This method generates a discrete reference sequence, which simplifies the processing of cross-correlation. For cross-correlation and therefore for pulse compression, time samples of the transmitted code sequence are obtained at times corresponding to chip intervals. Through this transformation, the number density of elements of the reference function can be diluted down to a single amplitude value within a chip interval.

[0136] Because the coding unit controls the chip spacing of the transmitter and the sampling points of the ADC at the receiver, the time samples of the transmitted code sequence and the discrete reference sequence are synchronized, where the sample intervals can differ by integers or rational numbers. Since the Nyquist-Shannon sampling theorem must be satisfied, the sampling density of the receiver output signal 29 is typically integers or rational numbers higher than the chip density. For cross-correlation, a diluted subset of the sampled signal data can be obtained, matching the chip spacing of the reference sequence. When calculating cross-correlation, this subset is shifted to a finer time interval of the sampled receiver output signal 29, ensuring no information loss.

[0137] Figure 11 An example of a diluted subset of the sampled receiver output signal 29 is schematically illustrated, wherein the chip intervals and time samples of this subset are fully synchronized, thus providing two samples per chip for both the receiver output signal 29 and the reference signal 30. When calculating the cross-correlation coefficient, only one of the diluted sampling modes of the receiver output signal 29 is acquired. By calculating the cross-correlation coefficient, the discrete reference sequence is shifted along the original fine-sampled receiver output signal, where samples of the matching subset are acquired.

[0138] For example, by using a discrete reference, the discrete pulse compression PCompr of the received signal samples is performed. kdsc It can be formalized as:

[0139]

[0140] Figure 12 This schematically illustrates another way of deriving the discrete reference function, where, instead of as... Figure 10 The diagram illustrates sampling the maximum and minimum values ​​of a simulated continuous waveform, where continuous bipolar pulses are represented by opposite amplitude pairs. For example, for each waveform, the sign of the extremum 38 is changed to form a corresponding "mirror" value 40. When the waveform begins with a positive amplitude, the positive maximum value is used as the coefficient of the reference function, and subsequent coefficients are the same value but negative. When the waveform begins with a negative amplitude, the negative minimum value is used as the coefficient of the reference function, and the next coefficient is the same value but positive. Thus, a discrete reference signal is generated from the simulated continuous reference signal 30, including the values ​​of the waveform extrema 38 and the mirror values ​​40 of the waveform extrema.

[0141] Although the invention has been illustrated above with reference to some preferred embodiments, it must be understood that many modifications and combinations of different features of the embodiments can be made. All such modifications fall within the scope of the appended claims.

Claims

1. An optoelectronic distance measuring module, configured for use in a surveying apparatus (1), wherein, The distance measuring module comprises a transmitting unit (24) with a laser diode or a low power fiber laser and is configured for emitting light pulses (18) by the laser diode or the low power fiber laser, wherein the light pulses (18) form a finite transmit code sequence (16, 16') with N chips (17), and processing a return signal corresponding to the emitted light pulses returned from a target in the environment to generate a measurement receiver output signal (29), wherein, in an ideal case, the measurement receiver output signal (29) has a shape of a desired receiver output signal formed by a bi-polar shaped desired receiver output signal component, wherein each of the desired receiver output signal components is assignable to a respective light pulse (18), characterized in that the distance measuring module is configured for providing a reference signal (30) similar to the desired receiver output signal and encoding reference pulses forming a finite reference code sequence with a number of chips (17) greater than N, wherein the reference signal (30) is formed by reference signal components (33, 33') occupying the bi-polar shape of the desired receiver output signal components, and wherein each of the reference signal components (33, 33') is assignable to a respective reference pulse in the reference code sequence, performing a cross-correlation between the measurement receiver output signal (29) and the reference signal (30) to generate a cross-correlation function, and identifying and timing a compressed pulse (35, 36, 37) in the cross-correlation function.

2. The distance measuring module according to claim 1, wherein, The light pulses (18) form a code sequence with minimal autocorrelation.

3. The distance measuring module of claim 1, wherein, The number of chips (17) of the reference code sequence is 3 times N, 4 times N or 5 times N.

4. The distance measuring module of claim 1, wherein, The transmit code sequence (16, 16') is a unipolar pulse sequence with less than 32 pulses (18).

5. The distance measuring module of claim 4, wherein, The unipolar pulse sequence corresponds to a code with a bipolar pulse pattern (20), wherein a negative value (19) or a positive value of the code with the bipolar pulse pattern (20) is set to zero and, upon return, the light pulse corresponds to the positive value or the negative value of the code with the bipolar pulse pattern (20).

6. The distance measuring module of claim 5, wherein, The unipolar pulse sequence corresponds to one of a Barker code, an Ipatiov code, an M-sequence and a Legendre sequence.

7. The distance measuring module of claim 5, wherein, Additional chips (17) with zero values are added between adjacent chips of the code with the bipolar pulse pattern (20) to form the transmit code sequence.

8. The distance measuring module of claim 1, wherein, The reference signal (30) is a continuous signal with multiple extrema, wherein each of the reference signal components (33, 33') is a continuous function part of the reference signal (30) with two extrema.

9. The distance measuring module of claim 1, wherein, The processing of the return signal is provided in a manner that all of the desired receiver output signal components have the same temporal signal width, wherein the reference signal (30) is provided in a manner that each of the reference signal components (33, 33') has the same temporal signal width as the desired receiver output signal components.

10. The distance measuring module of claim 1, wherein, The reference signal (30) is a discrete signal having a plurality of positive and negative values, wherein each of the reference signal components comprises a positive value (38) and a negative value (39).

11. The distance measuring module of claim 10, wherein, The positive and negative values have the same absolute value.

12. The distance measuring module of claim 10, wherein, Each of the reference signal components consists of two adjacent chips (17), wherein one of the two adjacent chips comprises the positive value (38) and the other of the two adjacent chips comprises the negative value (39).

13. The distance measuring module according to any one of claims 10 to 12, wherein, For the cross-correlation, the temporal samples of the measurement receiver output signal (29) are taken at a time corresponding to a chip interval of the reference signal (30).

14. The distance measuring module of claim 13, wherein, The encoding unit of the distance measurement module is configured to control the chip interval at which the finite transmit code sequence (16, 16') is transmitted and the sampling points of an analog-to-digital converter of the distance measurement module used for analyzing the measurement receiver output signal (29).

15. The distance measurement module according to claim 1, the distance measurement module being configured to identify the compressed pulse (35, 36, 37) in a digital representation of the cross-correlation function and, upon detection of the compressed pulse (35, 36, 37), to interpolate the compressed pulse between sampling points of the digital representation of the cross-correlation function to time the compressed pulse.

16. The distance measuring module of claim 15, wherein, using a sampling interpolation algorithm configured to locate the peak of the compressed pulse with sub-picosecond accuracy, or wherein a resampling method processing the sampled compressed pulse as an analog continuous signal is used, resulting in a precision down to the Cramer-Rao limit.

17. The distance measuring module of claim 1, wherein, The reference signal (30) is provided by an optimization using a merit function, wherein the merit function is a weighted sum of functions representing a side lobe ratio of a cross-correlation between a common reference function and different candidate receiver output signals associated with desired receiver output signals for different double echoes of a received optical pulse, wherein the different double echoes differ from each other by different inter-pulse spacings between echoes of the received optical pulse and the common reference function is associated with one of the different inter-pulse spacings.

18. The distance measuring module of claim 1, wherein, The distance measurement module is configured to coordinate the transmission of the optical pulses (18) to generate different finite transmit code sequences (16, 16') each having N chips, wherein each of the different finite transmit code sequences is associated with a respective desired receiver output signal formed by a desired receiver output signal component having a bipolar shape, wherein each of the desired receiver output signal components is assignable to a respective optical pulse (18), and different reference signals (30) are provided, wherein each of the different reference signals is similar to a different one of the desired receiver output signals and encodes reference pulses forming a limited reference code sequence of a number of chips greater than N, cross-correlations are performed in parallel between the measured receiver output signal (29) and each of the different reference signals to generate a cross-correlation function associated with each of the different reference signals, and compressed pulses (35, 36, 37) in each of the cross-correlation functions are identified and timed and each of the compressed pulses is associated with a respective one of different limited transmit code sequences (16, 16’).

19. The distance measurement module according to claim 18, configured to provide a sequential transmission pattern of different limited transmit code sequences (16, 16’) and to compare the sequential transmission pattern with the timing of the compressed pulses associated with each of the cross-correlation functions to provide a distance ambiguity correction.

20. A surveying apparatus (1) for three-dimensional space measurement of an environment by means of an optical distance measuring light beam (15), wherein The surveying apparatus comprises an optoelectronic distance measurement module according to one of claims 1 to 19.

21. Surveying apparatus (1) according to claim 20, wherein The surveying apparatus (1) is implemented as a tachymeter, a total station, a laser profilometer or a laser scanner. The surveying apparatus (1) is implemented as a tachymeter, a total station, a laser profilometer or a laser scanner.

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