A control method, control device, control chip and switching power supply

Abstract

The application discloses a control method, a control device, a control chip and a switching power supply. The switching power supply adopts an LLC resonant converter topology, and a resonant capacitor is located at a near-ground end. The control method comprises the following steps which are performed in real time: obtaining a first voltage signal representing the size of an output voltage of the switching power supply; obtaining a second voltage signal representing the size of a resonant cavity current by detecting the voltage across the resonant capacitor; comparing the first voltage signal with a set threshold value; when the first voltage signal is greater than or equal to the set threshold value, determining that the switching power supply is in a low-voltage input stage; otherwise, determining that the switching power supply is in a high-voltage input stage, then selecting a corresponding working mode, controlling the conduction and turn-off of each switch tube, and realizing the operation of the switching power supply; comparing the second voltage signal with a resonant cavity peak current protection threshold value, and when the second voltage signal is greater than or equal to the resonant cavity peak current protection threshold value, performing resonant cavity peak current protection of the switching power supply. The application can reduce the design difficulty of the control device.

Description

Technical Field

[0001] This invention relates to the field of switching converter technology, specifically to a control method, control device, control chip, and switching power supply. Background Technology

[0002] With the rapid development of modern power electronics technology and the increasing demand for new energy sources, power electronic converters are gradually moving towards higher power density. LLC resonant converters, due to their advantages such as soft switching, low power device losses, and simple drive circuits, have become a widely studied object. For stable systems (operating environments with small input voltage and load variations), conventional LLC resonant converters often use either variable frequency PFM or fixed frequency PWM control methods. However, in actual applications, the operating environment is complex and variable. For example, when the input voltage and load vary significantly, if only PFM control is used, the LLC resonant converter requires different output voltage gains for different input voltages and loads, causing the converter's operating frequency to vary over a wide range. This makes the design of the magnetic device parameters in the resonant circuit more difficult.

[0003] Furthermore, in high power density applications, the size of passive components is generally subject to stricter constraints, necessitating an increase in the converter's operating frequency. However, this results in higher switching losses for the power devices, and the efficiency of traditional PFM-controlled LLC resonant converters decreases significantly when the output voltage gain is wide. Under the same operating conditions, if only PWM control is used, the LLC resonant converter, being a fixed-frequency phase-shift control, has a relatively fixed operating frequency, facilitating the design of magnetic components in the resonant circuit. However, to maintain a relatively constant output voltage under different operating conditions, the converter's drive circuit needs to generate a wide-range phase-shift signal for different output voltage gains. This makes it difficult for the hysteresis bridge arm in the phase-shift circuit to achieve soft switching, failing to leverage the advantages of the resonant converter. Therefore, to ensure that all power switches can achieve soft switching at the maximum phase-shift angle, the maximum phase-shift angle of the power switches must be limited, thus limiting the output gain range of the PWM-controlled LLC resonant converter under a wide operating range.

[0004] In summary, in wide-range input and output applications, using a single control strategy or a conventional LLC resonant converter topology, the converter cannot simultaneously achieve the soft-switching and high-gain characteristics of a resonant converter.

[0005] Currently, the industry has researched relevant topologies and control methods to address the gain limitation problem of LLC resonant converters with a wide input voltage range. For wide voltage input, variable topology and control strategies are employed to solve the gain problem under wide input conditions. Chinese patent document CN110768535A, entitled "A Wide Gain Control Method for a Variable Topology LLC Resonant Converter," proposes a variable-mode control method based on a full-bridge LLC converter topology. This method achieves a wider voltage gain by detecting the input voltage and changing the topology and control method. However, this topology requires six switches in total, including the full-bridge circuit and auxiliary branches, necessitating strict control of the logic timing of the main and auxiliary switches, making the control method relatively complex.

[0006] Chinese patent document CN112087147A, entitled "A Wide Gain Control Method for Converters and Its Application," proposes a dual-mode control method based on a half-bridge LLC converter, adding an auxiliary branch to achieve a wider voltage gain by detecting the input voltage and changing the control mode. For LLC resonant converters, the operating frequency is the core parameter enabling soft switching, directly affecting the converter's efficiency and the stability of the entire system; therefore, monitoring the resonant cavity's operating state is crucial. In this half-bridge LLC resonant converter topology, the auxiliary branch's switching transistor is located near ground. To avoid using additional isolated power supplies and drivers to control the auxiliary branch's switching transistor, its resonant capacitor must be placed far from ground. Figure 1 As shown. For the half-bridge LLC converter topology of this patent, in order to monitor the operating status of the resonant cavity under wide input conditions, conventional methods can use sampling of the resonant capacitor at the far end, sampling of the milliohm resistor at the near end, and sampling of the Hall current sensor at the near end. Monitoring the resonant voltage at the far end requires the use of a high-voltage operational amplifier, which increases the cost and makes the monitoring circuit more complex. In addition, the common-mode voltage of the detected signal is relatively high, which can easily cause switch malfunctions, leading to a decrease in the stability of the entire converter circuit system. Milliohm resistor sampling is low in cost, but if a high-power precision sampling resistor is used in wide-gain operating conditions, its temperature drift is relatively serious, which is not suitable for monitoring the operating status of the resonant cavity, and it will increase the power loss of the converter. Sampling with a Hall current sensor can monitor a large operating current with low loss, but it is sensitive to changes in external magnetic fields, is expensive, and has a large size, making it unsuitable for high power density applications. Summary of the Invention

[0007] In view of this, the technical problem to be solved by the present invention is to provide a control method, control device, control chip and switching power supply, which can at least to some extent solve the shortcomings of the prior art.

[0008] As a first aspect of the present invention, the technical solution of the provided control method is as follows:

[0009] A control method is applied to a switching power supply employing an LLC resonant converter topology. The primary-side circuit includes a switching transistor S1, a switching transistor S2, a resonant capacitor Cr, a resonant inductor Lr, a magnetizing inductor Lm, a clamping switching transistor S3, and a clamping diode D1. The resonant capacitor Cr, the resonant inductor Lr, and the magnetizing inductor Lm constitute a resonant cavity. The drain of the switching transistor S1 is connected to the positive input terminal of the switching power supply. The source of the switching transistor S1 is simultaneously connected to the drain of the switching transistor S2, the source of the clamping switching transistor S3, and one end of the resonant inductor Lr. The other end of the resonant inductor Lr is connected to one end of the magnetizing inductor Lm. The drain of the clamping switching transistor S3 is connected to the cathode of the diode D1. The other end of the magnetizing inductor Lm is simultaneously connected to the anode of the diode D1 and one end of the resonant capacitor Cr. The other end of the resonant capacitor Cr is connected to the input ground terminal of the switching power supply and the source of the switching transistor S2. The control method includes the following real-time steps:

[0010] A first voltage signal characterizing the magnitude of the output voltage of the switching power supply is obtained by the output voltage sampling circuit;

[0011] The resonant capacitor sampling circuit obtains a second voltage signal characterizing the magnitude of the resonant cavity current by detecting the voltage across the resonant capacitor Cr.

[0012] The controller compares the first voltage signal with a set threshold. When the first voltage signal is greater than or equal to the set threshold, the switching power supply is determined to be in the low-voltage input stage. When the first voltage signal is less than the set threshold, the switching power supply is determined to be in the high-voltage input stage. Then, the corresponding operating mode is selected according to the stage of the switching power supply, and a first control signal, a second control signal, and a third control signal are generated.

[0013] The drive control circuit isolates and transforms the first control signal, the second control signal, and the third control signal into a first drive signal, a second drive signal, and a third drive signal, respectively, which are used to control the conduction and cutoff of the switching transistor S1, the switching transistor S2, and the clamping switch transistor S3, thereby realizing the operation of the switching power supply.

[0014] The controller compares the second voltage signal with the resonant cavity peak current protection threshold, and performs resonant cavity peak current protection of the switching power supply when the second voltage signal is greater than or equal to the resonant cavity peak current protection threshold.

[0015] Furthermore, when the switching power supply is in the low-voltage input stage, the gain value of the switching power supply can be changed by changing the frequency of the first drive signal and the second drive signal.

[0016] Preferably, when the switching power supply is in the low-voltage input stage, the third drive signal is continuously at a low level; within the same switching cycle, the first drive signal and the second drive signal alternately go to a high level, and after ignoring the dead time, the duty cycle of the first drive signal and the duty cycle of the second drive signal are both fixed at 50%.

[0017] Furthermore, when the switching power supply is in the high-voltage input stage, the gain value of the switching power supply can be changed by altering the duty cycle of the first drive signal, the second drive signal, and the third drive signal.

[0018] Preferably, when the switching power supply is in the high-voltage input stage, within the same switching cycle, the first drive signal, the third drive signal, and the second drive signal are sequentially high-level, and after ignoring the dead time, the sum of the duty cycles of the first drive signal, the second drive signal, and the third drive signal is 100%, and the duty cycles of the first drive signal and the second drive signal are equal.

[0019] As a second aspect of the present invention, the technical solution of the provided control device embodiment is as follows:

[0020] A control device is applied to a switching power supply. The switching power supply adopts an LLC resonant converter topology. Its primary-side circuit includes a switching transistor S1, a switching transistor S2, a resonant capacitor Cr, a resonant inductor Lr, a magnetizing inductor Lm, a clamping switching transistor S3, and a clamping diode D1. The resonant capacitor Cr, the resonant inductor Lr, and the magnetizing inductor Lm constitute a resonant cavity. The drain of the switching transistor S1 is connected to the positive input terminal of the switching power supply. The source of the switching transistor S1 is simultaneously connected to the drain of the switching transistor S2, the source of the clamping switching transistor S3, and one end of the resonant inductor Lr. The other end of the resonant inductor Lr is connected to one end of the magnetizing inductor Lm. The drain of the clamping switching transistor S3 is connected to the cathode of the diode D1. The other end of the magnetizing inductor Lm is simultaneously connected to the anode of the diode D1 and one end of the resonant capacitor Cr. The other end of the resonant capacitor Cr is connected to the input ground terminal of the switching power supply and the source of the switching transistor S2. The control device includes the following circuit:

[0021] The output voltage sampling circuit is used to acquire a first voltage signal that characterizes the magnitude of the output voltage of the switching power supply in real time.

[0022] The resonant capacitor sampling circuit is used to obtain a second voltage signal characterizing the magnitude of the resonant cavity current in real time by detecting the voltage across the resonant capacitor Cr.

[0023] The controller is used to compare the first voltage signal with a set threshold. When the first voltage signal is greater than or equal to the set threshold, it determines that the switching power supply is in a low-voltage input stage. When the first voltage signal is less than the set threshold, it determines that the switching power supply is in a high-voltage input stage. Then, according to the stage in which the switching power supply is located, the controller selects the corresponding working mode and generates a first control signal, a second control signal, and a third control signal.

[0024] The drive control circuit is used to isolate and transform the first control signal, the second control signal, and the third control signal into a first drive signal, a second drive signal, and a third drive signal in real time, respectively, so as to control the conduction and cutoff of the switching transistor S1, the switching transistor S2, and the clamping switch S3, thereby realizing the operation of the switching power supply.

[0025] The controller is further configured to compare the second voltage signal with the resonant cavity peak current protection threshold, and to perform resonant cavity peak current protection of the switching power supply when the second voltage signal is ≥ the resonant cavity peak current protection threshold.

[0026] Furthermore, when the switching power supply is in the low-voltage input stage, the controller changes the gain value of the switching power supply by changing the frequency of the first drive signal and the second drive signal.

[0027] Preferably, when the switching power supply is in the low-voltage input stage, the third control signal generated by the controller is continuously at a low level; within the same switching cycle, the first drive signal and the second drive signal generated by the drive control circuit alternately go to a high level, and after ignoring the dead time, the duty cycle of the first drive signal and the duty cycle of the second drive signal are both fixed at 50%.

[0028] Furthermore, when the switching power supply is in the high-voltage input stage, the controller changes the gain value of the switching power supply by changing the duty cycle of the first drive signal, the second drive signal, and the third drive signal.

[0029] Preferably, when the switching power supply is in the high-voltage input stage, within the same switching cycle, the first drive signal, the third drive signal, and the second drive signal generated by the drive control circuit are sequentially at high level, and after ignoring the dead time, the sum of the duty cycles of the first drive signal, the second drive signal, and the third drive signal is 100%, and the duty cycles of the first drive signal and the second drive signal are equal.

[0030] As a third aspect of the present invention, the technical solution of the provided control chip embodiment is as follows:

[0031] A control chip that integrates the control device described in any of the second aspects above.

[0032] As a fourth aspect of the present invention, the technical solution of the provided switching power supply embodiment is as follows:

[0033] A switching power supply employs an LLC resonant converter topology. Its primary-side circuit includes a switching transistor S1, a switching transistor S2, a resonant capacitor Cr, a resonant inductor Lr, a magnetizing inductor Lm, a clamping switching transistor S3, and a clamping diode D1. The drain of the switching transistor S1 is connected to the positive input terminal of the switching power supply. The source of the switching transistor S1 is simultaneously connected to the drain of the switching transistor S2, the source of the clamping switching transistor S3, and one end of the resonant inductor Lr. The other end of the resonant inductor Lr is connected to one end of the magnetizing inductor Lm. The drain of the clamping switching transistor S3 is connected to the cathode of the diode D1. The other end of the magnetizing inductor Lm is simultaneously connected to the anode of the diode D1 and one end of the resonant capacitor Cr. The other end of the resonant capacitor Cr is connected to the input ground terminal of the switching power supply and the source of the switching transistor S2. The switching power supply further includes the control device described in any of the second aspects above.

[0034] The beneficial effects of this invention are as follows:

[0035] (1) In the LLC resonant converter of the present invention, the resonant capacitor is located near the ground end, which makes it easy to sample the voltage of the resonant capacitor. It is only necessary to monitor the voltage of the resonant capacitor near the ground end. For example, a simple RC voltage divider circuit can be used to sample the voltage of the resonant capacitor in real time. The sampled voltage can characterize the magnitude of the resonant cavity current and be directly transmitted to the controller. It can be used for converter working state analysis and converter protection control, which reduces the design difficulty of the control device.

[0036] (2) In this embodiment of the invention, the operating conditions of the switching power supply are divided into low-voltage input stage and high-voltage input stage. The corresponding working mode is selected according to the stage of the switching power supply, and then the switching transistor is turned on and off. The control mode of the converter for different input voltage segments can be smoothly switched.

[0037] (3) When the embodiment of the present invention adopts two control modes, variable pulse frequency PFM and variable pulse width PWM, the operating frequency of the converter changes within a small range, which reduces the difficulty of designing the resonant cavity parameters of the converter under wide input and output conditions, and effectively improves the efficiency and power density of the LLC resonant converter under wide input and output conditions.

[0038] (4) The LLC resonant converter in this embodiment of the invention adds a clamping branch. During the high voltage input stage, when the clamping switch in the clamping branch is turned on, the energy of the resonant current during the circulating current stage is stored in the loop composed of the transformer excitation inductance Lm, the resonant inductance Lr and the clamping branch. The resonant current will not flow through the resonant capacitor Cr. This can avoid the loss caused by the parasitic resistance of the resonant capacitor Cr, and also reduce the average value of the resonant current during high voltage input, thereby improving the overall efficiency of the LLC resonant converter. Attached Figure Description

[0039] Figure 1 A schematic diagram of the conventional sampling points for the operating state of a half-bridge LLC converter circuit and its resonant cavity;

[0040] Figure 2 This is a schematic diagram of the half-bridge LLC resonant converter circuit and its control principle according to the present invention.

[0041] Figure 3 The main operating waveforms of the half-bridge LLC resonant converter of the present invention under low-voltage PFM control are shown in the figure.

[0042] Figures 4-9 are equivalent circuit diagrams of each switching mode of the half-bridge LLC resonant converter of the present invention under low-voltage PFM control;

[0043] Figure 10 The main operating waveforms of the half-bridge LLC resonant converter of the present invention under high-voltage PWM control are shown below.

[0044] Figures 11-18 This is the equivalent circuit diagram of each switching mode of the half-bridge LLC resonant converter of the present invention under high-voltage PWM control. Detailed Implementation

[0045] It should be noted that, unless otherwise specified, the embodiments and features described in this application can be combined with each other. This application will now be described in detail with reference to the accompanying drawings and embodiments.

[0046] To enable those skilled in the art to better understand the present application, the technical solutions in the embodiments of the present application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present application, and not all embodiments. Based on the embodiments in the present application, all other embodiments obtained by those of ordinary skill in the art without creative effort should fall within the scope of protection of the present application.

[0047] It should be noted that the terms "first," "second," etc., in the specification, claims, and accompanying drawings of this application are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence. It should be understood that such data can be used interchangeably where appropriate for the purposes of describing embodiments of this application herein. Furthermore, the terms "comprising" and "having," and any variations thereof, are intended to cover non-exclusive inclusion; for example, a process, method, system, product, or apparatus that comprises a series of steps or units is not necessarily limited to those steps or units explicitly listed, but may include other steps or units not explicitly listed or inherent to such processes, methods, products, or apparatus.

[0048] It should be understood that in the specification, claims, and drawings, when a step is described as continuing into another step, the step may directly continue into that other step or be continued into that other step through a third step; when an element / unit is described as "continuing" into another element / unit, the element / unit may be "directly connected" to that other element / unit or "connected" to that other element / unit through a third element / unit.

[0049] Furthermore, the accompanying drawings are merely illustrative of this disclosure and are not necessarily drawn to scale. The same reference numerals in the drawings denote the same or similar parts, and therefore repeated descriptions thereof will be omitted. Some block diagrams shown in the drawings are functional entities and do not necessarily correspond to physically or logically independent entities. These functional entities can be implemented in software, in one or more hardware modules or integrated circuits, or in different network and / or processor devices and / or microcontroller devices.

[0050] First Embodiment

[0051] This embodiment provides a control method applied to a switching power supply. Please refer to the schematic diagram of the switching power supply. Figure 2The LLC resonant converter topology is adopted. Its primary-side circuit includes switching transistors S1 and S2, a resonant capacitor Cr, a resonant inductor Lr, a magnetizing inductor Lm, a clamping switch S3, and a clamping diode D1. The resonant capacitor Cr, resonant inductor Lr, and magnetizing inductor Lm constitute a resonant cavity. The drain of switching transistor S1 is connected to the positive input terminal of the switching power supply. The source of switching transistor S1 is simultaneously connected to the drain of switching transistor S2, the source of clamping switch S3, and one end of the resonant inductor Lr. The other end of the resonant inductor Lr is connected to one end of the magnetizing inductor Lm. The drain of clamping switch S3 is connected to the cathode of diode D1. The other end of the magnetizing inductor Lm is simultaneously connected to the anode of diode D1 and one end of the resonant capacitor Cr. The other end of the resonant capacitor Cr is connected to the input ground terminal of the switching power supply and the source of switching transistor S2. The control method of this embodiment includes the following real-time steps:

[0052] The first voltage signal characterizing the magnitude of the output voltage of the switching power supply is obtained by the output voltage sampling circuit;

[0053] The second voltage signal, which characterizes the magnitude of the resonant cavity current, is obtained by the resonant capacitor sampling circuit by detecting the voltage across the resonant capacitor Cr.

[0054] The controller compares the first voltage signal with a set threshold. When the first voltage signal is greater than or equal to the set threshold, it determines that the switching power supply is in the low-voltage input stage. When the first voltage signal is less than the set threshold, it determines that the switching power supply is in the high-voltage input stage. Then, according to the stage of the switching power supply, the corresponding working mode is selected, and the first control signal, the second control signal, and the third control signal are generated.

[0055] The drive control circuit isolates and transforms the first control signal, the second control signal, and the third control signal into the first drive signal, the second drive signal, and the third drive signal, respectively, which are used to control the conduction and cutoff of the switching transistor S1, the switching transistor S2, and the clamping switch S3, thereby realizing the operation of the switching power supply.

[0056] The controller compares the second voltage signal with the resonant cavity peak current protection threshold, and performs resonant cavity peak current protection of the switching power supply when the second voltage signal is greater than or equal to the resonant cavity peak current protection threshold.

[0057] In the LLC resonant converter described in the above embodiment, the resonant capacitor Cr is located near ground, which facilitates the sampling of the voltage of the resonant capacitor Cr. It is only necessary to monitor the voltage of the resonant capacitor Cr near ground. For example, a simple RC voltage divider circuit can be used to sample the voltage of the resonant capacitor Cr in real time. The sampled voltage can characterize the magnitude of the resonant cavity current and be directly transmitted to the controller. This can be used for converter operating status analysis and converter protection control, which reduces the design difficulty of the control device.

[0058] Furthermore, the above control method divides the operating conditions of the switching power supply into low-voltage input stage and high-voltage input stage, selects the corresponding operating mode according to the stage of the switching power supply, and then controls the switching transistor to switch on and off, so that the converter operating mode of different input voltage segments can be smoothly switched.

[0059] Furthermore, when the switching power supply is in the low-voltage input stage, the gain value of the switching power supply is changed by altering the frequencies of the first and second drive signals. Within the operating frequency range of the switching power supply, the lower the frequency, the higher the gain value, and vice versa. For a specific operating waveform under this condition, please refer to [link to waveform description]. Figure 3 The third control signal is always low, so the clamping switch S3 is always off and the clamping branch does not work. In the same switching cycle, the first drive signal and the second drive signal alternately go high, and the duty cycle of the first drive signal and the duty cycle of the second drive signal are both fixed at 50% after ignoring the dead time.

[0060] Furthermore, when the switching power supply is in the high-voltage input stage, the gain value of the switching power supply is changed by altering the duty cycles of the first, second, and third drive signals. A larger duty cycle results in a larger gain, and vice versa. For a specific operating waveform under this condition, please refer to [link to waveform description]. Figure 10 Within the same switching cycle, the first drive signal, the third drive signal, and the second drive signal are sequentially high-level. Ignoring the dead time, the sum of the duty cycles of the first drive signal, the second drive signal, and the third drive signal is 100%, and the duty cycles of the first drive signal and the second drive signal are equal. The clamping switch S3's on-time is located between switches S1 and S2, used to clamp the energy in the loop composed of the magnetizing inductor Lm and the resonant inductor Lr during the off-time of both switches S1 and S2, thus reducing losses.

[0061] When the switching power supply adopts both variable pulse frequency (PFM) and variable pulse width (PWM) control methods, the operating frequency variation range of the switching power supply is small, which can reduce the design difficulty of the resonant cavity parameters of the converter under wide input and output conditions, and effectively improve the efficiency and power density of the switching power supply under wide input and output conditions.

[0062] Second Embodiment

[0063] This embodiment provides a control device applied to a switching power supply. The switching power supply adopts an LLC resonant converter topology. Its primary-side circuit includes a switching transistor S1, a switching transistor S2, a resonant capacitor Cr, a resonant inductor Lr, a magnetizing inductor Lm, a clamping switching transistor S3, and a clamping diode D1. The resonant capacitor Cr, the resonant inductor Lr, and the magnetizing inductor Lm constitute a resonant cavity. The drain of the switching transistor S1 is connected to the positive input terminal of the switching power supply. The source of the switching transistor S1 is simultaneously connected to the drain of the switching transistor S2, the source of the clamping switching transistor S3, and one end of the resonant inductor Lr. The other end of the resonant inductor Lr is connected to one end of the magnetizing inductor Lm. The drain of the clamping switching transistor S3 is connected to the cathode of the diode D1. The other end of the magnetizing inductor Lm is simultaneously connected to the anode of the diode D1 and one end of the resonant capacitor Cr. The other end of the resonant capacitor Cr is connected to the input ground terminal of the switching power supply and the source of the switching transistor S2. The control device includes the following circuit:

[0064] Output voltage sampling circuit is used to acquire the first voltage signal that characterizes the magnitude of the output voltage of the switching power supply in real time;

[0065] The resonant capacitor sampling circuit is used to obtain a second voltage signal characterizing the magnitude of the resonant cavity current in real time by detecting the voltage across the resonant capacitor Cr.

[0066] The controller is used to compare the first voltage signal with a set threshold. When the first voltage signal is greater than or equal to the set threshold, it is determined that the switching power supply is in the low voltage input stage. When the first voltage signal is less than the set threshold, it is determined that the switching power supply is in the high voltage input stage. Then, according to the stage of the switching power supply, the corresponding working mode is selected, and the first control signal, the second control signal and the third control signal are generated.

[0067] The drive control circuit is used to isolate and transform the first control signal, the second control signal, and the third control signal into the first drive signal, the second drive signal, and the third drive signal in real time, respectively, which are used to control the conduction and cutoff of the switching transistor S1, the switching transistor S2, and the clamping switch S3, so as to realize the operation of the switching power supply.

[0068] The controller is also used to compare the second voltage signal with the resonant cavity peak current protection threshold, and to perform resonant cavity peak current protection of the switching power supply when the second voltage signal is greater than or equal to the resonant cavity peak current protection threshold.

[0069] Furthermore, when the switching power supply is in the low-voltage input stage, the drive control circuit changes the gain value of the switching power supply by changing the frequency of the first drive signal and the second drive signal. In this operating condition, preferably, the third control signal generated by the controller is continuously at a low level; within the same switching cycle, the first drive signal and the second drive signal generated by the controller alternately go to a high level, and ignoring the dead time, the duty cycle of both the first drive signal and the second drive signal is fixed at 50%.

[0070] Furthermore, when the switching power supply is in the high-voltage input stage, the drive control circuit changes the gain value of the switching power supply by changing the duty cycles of the first drive signal, the second drive signal, and the third drive signal. Preferably, in the same switching cycle, the first drive signal, the third drive signal, and the second drive signal generated by the controller are sequentially high-level, and the sum of the duty cycles of the first drive signal, the second drive signal, and the third drive signal, ignoring the dead time, is 100%. The duty cycles of the first drive signal and the second drive signal are equal. Moreover, the sum of the duty cycles of the first drive signal and the second drive signal is complementary to the conduction time of the third drive signal.

[0071] The various implementation schemes of the control device in this embodiment correspond to the various implementation schemes of the control method in the first embodiment, and the beneficial effects are the same, so they will not be described in detail.

[0072] Third Embodiment

[0073] This embodiment provides a control chip that integrates any of the control devices in the second embodiment. Designing the key circuits in the control device as integrated circuits enables the switching power supply of this invention to be smaller in size and more efficient in production.

[0074] Fourth embodiment

[0075] This embodiment provides a switching power supply; please refer to [link / reference]. Figure 2 The switching power supply adopts an LLC resonant converter topology. Its primary-side circuit includes switching transistors S1 and S2, resonant capacitor Cr, resonant inductor Lr, magnetizing inductor Lm, clamping switching transistor S3, and clamping diode D1. The drain of switching transistor S1 is connected to the positive input terminal of the switching power supply. The source of switching transistor S1 is simultaneously connected to the drain of switching transistor S2, the source of clamping switching transistor S3, and one end of resonant inductor Lr. The other end of resonant inductor Lr is connected to one end of magnetizing inductor Lm. The drain of clamping switching transistor S3 is connected to the cathode of diode D1. The other end of magnetizing inductor Lm is simultaneously connected to the anode of diode D1 and one end of resonant capacitor Cr. The other end of resonant capacitor Cr is connected to the input ground terminal of the switching power supply and the source of switching transistor S2. The switching power supply also includes any of the control devices in the second embodiment.

[0076] The following description, in conjunction with the accompanying drawings, details the operation of the half-bridge LLC resonant converter of the present invention under low and high voltage conditions.

[0077] When the first voltage signal is greater than or equal to the set threshold, the converter operates in PFM control mode. During this stage, the clamping switch S3 remains off, meaning the clamping branch 30 does not operate. Figure 3Figure 4 to 9 show the main operating waveforms of the converter in low-voltage PFM mode. Vgs1 is the first drive signal of primary-side switch S1, Vgs2 is the second drive signal of primary-side switch S2, iLr is the resonant cavity current, iLm is the magnetizing inductor current, iD1 is the current of secondary-side switch SR1, and iD2 is the current of secondary-side switch SR2. Based on the operating characteristics of the switches in different stages, six typical switching modes of the converter in one switching cycle are listed, as shown in Figures 4 to 9.

[0078] Switching mode 1 (t0, t1): such as Figure 4a As shown, at time t0, due to the inductive impedance characteristic of the converter's resonant cavity, the current phase lags behind the voltage. After the parasitic capacitance of switch S1 has discharged, the resonant current iLr freewheels through the body diode of switch S1, as follows. Figure 4b As shown, at this time, the switch S1 can achieve zero-voltage turn-on; during this stage, the secondary switch SR1 is in the on state, the switch SR2 is in the off state, the voltage across the magnetizing inductor Lm is clamped to nVo by the output (n is the turns ratio of transformer T, Vo is the output voltage of the converter), the magnetizing inductor current iLm decreases linearly in the reverse direction, and the resonant current iLr decreases rapidly to zero.

[0079] Switching mode 2 (t1, t2): such as Figure 5 As shown, at time t1, the resonant current iLr crosses zero. The resonant current iLr is a standard sine wave and increases in the positive direction. The primary-side switch S1 and the secondary-side switch SR1 continue to conduct.

[0080] Switching mode 3 (t2, t3): such as Figure 6 As shown, at time t2, the primary-side switch S1 and the secondary-side switch SR1 continue to conduct, and the excitation current iLm increases linearly after crossing zero.

[0081] Switching mode 4 (t3, t4): such as Figure 7 As shown, at time t3, the resonant current iLr is equal to the magnetizing current iLm, the current ID1 of the secondary switch SR1 is 0, the magnetizing inductor is no longer clamped (the transformer has no energy transfer), the switch SR1 is turned off, and zero current turn-off (ZCS) is achieved. During this stage, the resonant capacitor Cr, the resonant inductor Lr, and the magnetizing inductor Lm all participate in the resonance.

[0082] Switching mode 5 (t4, t5): such as Figure 8 As shown, at time t4, the primary-side switch S1 is turned off. At this time, the resonant current iLr begins to discharge the parasitic capacitance of the primary-side switch S2, providing conditions for its zero-voltage turn-on, and at the same time charging the parasitic capacitance of the primary-side switch S1.

[0083] Switching mode 6 (t5, t6): such as Figure 9As shown, at time t5, the resonant current freewheels through the body diode of the primary-side switch S2, and the secondary-side switch SR2 is turned on, causing the resonant current iLr and the excitation current iLm to decrease in the positive direction; at time t6, the primary-side switch S2 achieves ZVS, and the circuit enters the second half of the cycle.

[0084] When the first voltage signal is less than the set threshold, the converter switches to PWM control mode. Figure 10 This diagram shows the main operating waveforms of the converter in high-voltage input PWM control mode. Vgs1 is the first drive signal for primary-side switch S1, Vgs2 is the second drive signal for primary-side switch S2, Vgs3 is the third drive signal for clamping switch S3, iLr is the resonant cavity current, iLm is the magnetizing inductor current, iD1 is the current of secondary-side switch SR1, and iD2 is the current of secondary-side switch SR2. Based on the operating characteristics of the switches in different stages, seven switching modes of the converter within one switching cycle are listed, and are respectively... Figures 11-18 As shown.

[0085] Switching mode 1 (t0, t1): such as Figure 11 As shown, at time t0, the voltage across the magnetizing inductor Lm is clamped to nVo by the output, the magnetizing inductor current iLm increases linearly in the reverse direction, the resonant current iLr decreases in the reverse direction, the primary-side switch S1 turns on with zero voltage, and the secondary-side switch SR2 continues to conduct.

[0086] Switching mode 2 (t1, t2): such as Figure 12 As shown, at time t1, the resonant current iLr is equal to the magnetizing inductor current iLm, which is equivalent to no energy being transferred to the secondary side. At this time, the secondary side switch SR2 is turned off with zero current, while the secondary side switch SR1 is turned on. During this stage, the magnetizing inductor current iLr first decreases linearly in the reverse direction, and then increases linearly in the positive direction after crossing zero. The resonant current iLm first decreases rapidly in the reverse direction, and then increases rapidly after crossing zero, always being greater than the magnetizing inductor current.

[0087] Switching mode 3 (t2, t3): such as Figure 13 As shown, at time t2, the resonant current iLr reaches the maximum set value, and the controller sends the S1 turn-off drive signal. At this time, the primary-side switch SR1 is turned off, and the resonant current iLr begins to decrease linearly. The voltage across the magnetizing inductor Lm is clamped to nVo by the output, and the magnetizing inductor current iLm continues to increase linearly in the positive direction. At the same time, the secondary-side switch SR1 continues to conduct.

[0088] Switching mode 4 (t3, t4): such as Figure 14 As shown, at time t3, the clamping switch S3 is turned on, the primary winding of the transformer is short-circuited, and the resonant capacitor no longer participates in the resonance. However, since the voltage across the magnetizing inductor Lm is still clamped by the output, the magnetizing inductor current iLm continues to increase linearly in the positive direction.

[0089] Switching mode 5 (t4, t5): such as Figure 15 As shown, at time t4, the resonant current iLr is equal to the magnetizing inductor current iLm, and no energy is transferred to the secondary side. Therefore, the secondary switch SR1 can achieve zero-current turn-off. For a period of time thereafter, the resonant current iLr circulates through the clamping branch and remains equal to the magnetizing inductor current iLm. The secondary switch is always in the off state, and the energy required by the load is entirely provided by the output filter capacitor Co.

[0090] Switching mode 6 (t5, t6): such as Figure 16 As shown, at time t5, the clamping switch S3 is turned off, and the resonant current iLr and the magnetizing inductor current iLm decrease linearly at the same time. The resonant current iLr begins to discharge the parasitic capacitance of the switch S2. When the voltage across the parasitic capacitance of the switch S2 drops to zero, the body diode of the switch S2 turns on, providing the conditions for the zero-voltage turn-on of the switch S2.

[0091] Switching mode 7 (t6, t7): such as Figure 17 As shown, at time t6, the primary-side switch S2 achieves zero-voltage conduction, and the secondary-side switch S2 is turned on. The current flowing through the secondary-side switch is proportional to the difference between the resonant current iLr and the magnetizing inductor current iLm. During this stage, the magnetizing inductor current iLm decreases linearly in the positive direction, while the resonant current iLr first decreases in the positive direction and then increases in the reverse direction after crossing zero.

[0092] Switching mode 8 (t7, t8): such as Figure 18 As shown, at time t7, the resonant current iLr reaches its reverse peak value, and at the same time, the magnetizing inductor current iLm reaches the zero-crossing point, and the primary-side switch S2 is turned off. The resonant current iLr charges the parasitic capacitance of switch S2 and discharges the parasitic capacitance of switch S1. When the voltage across the parasitic capacitance of switch S1 drops to zero, the body diode of switch S1 turns on, providing the conditions for the zero-voltage turn-on of switch S1.

[0093] Based on the above brief description of the converter's operation, it can be seen that this wide-gain half-bridge LLC resonant converter can achieve zero-voltage turn-on of the switching transistors and zero-current turn-off of the secondary side under a wide range of input and output conditions.

[0094] The switching power supply in this embodiment enables precise monitoring of the resonant cavity's operating state in wide-gain applications, unaffected by input voltage limitations. This facilitates stable control of the converter system, and the control circuit boasts a simple structure, high reliability, and ease of implementation. By adding a clamping branch to the conventional LLC topology, this solution reduces energy losses caused by oscillations between the resonant inductance, transformer leakage inductance, and junction capacitance during the turn-off periods of the two switching transistors in the inverter circuit, without increasing the complexity of the system control circuit, thereby improving converter efficiency.

[0095] Furthermore, by employing the PFM and PWM dual-mode control method, the switching power supply of this invention can achieve the advantages of soft switching of LLC resonant converters in a wide range of input and output applications, while ensuring high output voltage gain and achieving optimal converter efficiency under full-range load conditions.

[0096] The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. For those skilled in the art, any modifications, equivalent substitutions, improvements, etc., made without departing from the principle of the present invention should be included within the protection scope of the present invention.

Claims

1. A control method applied to a switching power supply, wherein the switching power supply adopts an LLC resonant converter topology, and its primary-side circuit includes a switching transistor S1, a switching transistor S2, a resonant capacitor Cr, a resonant inductor Lr, a magnetizing inductor Lm, a clamping switching transistor S3, and a clamping diode D1, wherein the resonant capacitor Cr, the resonant inductor Lr, and the magnetizing inductor Lm constitute a resonant cavity; the drain of the switching transistor S1 is connected to the positive input terminal of the switching power supply, the source of the switching transistor S1 is simultaneously connected to the drain of the switching transistor S2, the source of the clamping switching transistor S3, and one end of the resonant inductor Lr, the other end of the resonant inductor Lr is connected to one end of the magnetizing inductor Lm, the drain of the clamping switching transistor S3 is connected to the cathode of the diode D1, the other end of the magnetizing inductor Lm is simultaneously connected to the anode of the diode D1 and one end of the resonant capacitor Cr, and the other end of the resonant capacitor Cr is connected to the input ground terminal of the switching power supply and the source of the switching transistor S2; characterized in that, The control method includes the following steps performed in real time: A first voltage signal characterizing the magnitude of the output voltage of the switching power supply is obtained by the output voltage sampling circuit; The resonant capacitor sampling circuit obtains a second voltage signal characterizing the magnitude of the resonant cavity current by detecting the voltage across the resonant capacitor Cr. The controller compares the first voltage signal with a set threshold. When the first voltage signal is greater than or equal to the set threshold, the switching power supply is determined to be in the low-voltage input stage. When the first voltage signal is less than the set threshold, the switching power supply is determined to be in the high-voltage input stage. Then, the corresponding operating mode is selected according to the stage of the switching power supply, and a first control signal, a second control signal, and a third control signal are generated. The drive control circuit isolates and transforms the first control signal, the second control signal, and the third control signal into a first drive signal, a second drive signal, and a third drive signal, respectively, which are used to control the conduction and cutoff of the switching transistor S1, the switching transistor S2, and the clamping switch transistor S3, thereby realizing the operation of the switching power supply. The controller compares the second voltage signal with the resonant cavity peak current protection threshold, and performs resonant cavity peak current protection of the switching power supply when the second voltage signal is greater than or equal to the resonant cavity peak current protection threshold. When the switching power supply is in the low-voltage input stage, the gain value of the switching power supply is changed by changing the frequency of the first drive signal and the second drive signal. When the switching power supply is in the high-voltage input stage, the gain value of the switching power supply is changed by changing the duty cycle of the first drive signal, the second drive signal, and the third drive signal.

2. The control method according to claim 1, characterized in that: When the switching power supply is in the low-voltage input stage, the third drive signal is continuously low; within the same switching cycle, the first drive signal and the second drive signal alternately go high, and after ignoring the dead time, the duty cycle of the first drive signal and the duty cycle of the second drive signal are both fixed at 50%.

3. The control method according to claim 1, characterized in that: When the switching power supply is in the high-voltage input stage, within the same switching cycle, the first drive signal, the third drive signal, and the second drive signal are sequentially high level, and after ignoring the dead time, the sum of the duty cycles of the first drive signal, the second drive signal, and the third drive signal is 100%, and the duty cycles of the first drive signal and the second drive signal are equal.

4. A control device applied to a switching power supply, the switching power supply employing an LLC resonant converter topology, its primary-side circuit including a switching transistor S1, a switching transistor S2, a resonant capacitor Cr, a resonant inductor Lr, a magnetizing inductor Lm, a clamping switching transistor S3, and a clamping diode D1, wherein the resonant capacitor Cr, the resonant inductor Lr, and the magnetizing inductor Lm constitute a resonant cavity; the drain of the switching transistor S1 is connected to the positive input terminal of the switching power supply, the source of the switching transistor S1 is simultaneously connected to the drain of the switching transistor S2, the source of the clamping switching transistor S3, and one end of the resonant inductor Lr, the other end of the resonant inductor Lr is connected to one end of the magnetizing inductor Lm, the drain of the clamping switching transistor S3 is connected to the cathode of the diode D1, the other end of the magnetizing inductor Lm is simultaneously connected to the anode of the diode D1 and one end of the resonant capacitor Cr, and the other end of the resonant capacitor Cr is connected to the input ground terminal of the switching power supply and the source of the switching transistor S2; characterized in that, The control device includes the following circuitry: The output voltage sampling circuit is used to acquire a first voltage signal that characterizes the magnitude of the output voltage of the switching power supply in real time. The resonant capacitor sampling circuit is used to obtain a second voltage signal characterizing the magnitude of the resonant cavity current in real time by detecting the voltage across the resonant capacitor Cr. The controller is used to compare the first voltage signal with a set threshold. When the first voltage signal is greater than or equal to the set threshold, it determines that the switching power supply is in a low-voltage input stage. When the first voltage signal is less than the set threshold, it determines that the switching power supply is in a high-voltage input stage. Then, according to the stage in which the switching power supply is located, the controller selects the corresponding working mode and generates a first control signal, a second control signal, and a third control signal. The drive control circuit is used to isolate and transform the first control signal, the second control signal, and the third control signal into a first drive signal, a second drive signal, and a third drive signal in real time, respectively, so as to control the conduction and cutoff of the switching transistor S1, the switching transistor S2, and the clamping switch S3, thereby realizing the operation of the switching power supply. The controller is further configured to compare the second voltage signal with the resonant cavity peak current protection threshold, and to perform resonant cavity peak current protection of the switching power supply when the second voltage signal is ≥ the resonant cavity peak current protection threshold; When the switching power supply is in the low-voltage input stage, the controller changes the gain value of the switching power supply by changing the frequency of the first drive signal and the second drive signal. When the switching power supply is in the high-voltage input stage, the controller changes the gain value of the switching power supply by changing the duty cycle of the first drive signal, the second drive signal, and the third drive signal.

5. The control device according to claim 4, characterized in that: When the switching power supply is in the low-voltage input stage, the third control signal generated by the controller is continuously at a low level; within the same switching cycle, the first drive signal and the second drive signal generated by the drive control circuit alternately go to a high level, and after ignoring the dead time, the duty cycle of the first drive signal and the duty cycle of the second drive signal are both fixed at 50%.

6. The control device according to claim 4, characterized in that: When the switching power supply is in the high-voltage input stage, within the same switching cycle, the first drive signal, the third drive signal, and the second drive signal generated by the drive control circuit are sequentially at high level, and after ignoring the dead time, the sum of the duty cycles of the first drive signal, the second drive signal, and the third drive signal is 100%, and the duty cycles of the first drive signal and the second drive signal are equal.

7. A control chip, characterized in that: It integrates the control device described in any one of claims 4 to 6.

8. A switching power supply, wherein the switching power supply adopts an LLC resonant converter topology, and its primary-side circuit includes a switching transistor S1, a switching transistor S2, a resonant capacitor Cr, a resonant inductor Lr, a magnetizing inductor Lm, a clamping switching transistor S3, and a clamping diode D1; the drain of the switching transistor S1 is connected to the positive input terminal of the switching power supply, the source of the switching transistor S1 is simultaneously connected to the drain of the switching transistor S2, the source of the clamping switching transistor S3, and one end of the resonant inductor Lr, the other end of the resonant inductor Lr is connected to one end of the magnetizing inductor Lm, the drain of the clamping switching transistor S3 is connected to the cathode of the diode D1, the other end of the magnetizing inductor Lm is simultaneously connected to the anode of the diode D1 and one end of the resonant capacitor Cr, and the other end of the resonant capacitor Cr is connected to the input ground terminal of the switching power supply and the source of the switching transistor S2; characterized in that, The switching power supply further includes the control device described in any one of claims 4 to 6.

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