A Sensorless Control Method for Switched Reluctance Motor at Low Speed Based on High-Frequency Injection

Abstract

A sensorless control method for synchronous reluctance motors based on high-frequency injection, which relates to the technical field of motor control. The present invention is to solve the problems of the existing sensorless control method for synchronous reluctance motors, such as low angle tracking accuracy, limited dynamic performance, excessive preprocessing tests required, and poor robustness, which cannot meet the requirements of industrial production applications for high-performance motors and drive systems. In the operation process of the synchronous reluctance motor, the present invention performs online identification of inductance parameters, realizing the simultaneous observation of angular position and inductance parameters. In addition, a four-vector high-frequency square-wave voltage injection method is adopted, which minimizes the calculation amount and reduces the influence of the injected voltage pulse on the sensorless control system.

Description

Technical Field

[0001] The present invention belongs to the technical field of motor control, and particularly relates to the control of a synchronous reluctance motor without a sensor at low speed. Background Art

[0002] Synchronous reluctance motors have existed for nearly a century, but it is only in recent years that they have become a commercially viable application motor. Synchronous reluctance motors have gradually attracted wide attention due to their advantages such as simple structure, low cost, and high efficiency. Structurally, the rotor of a synchronous reluctance motor is tightly laminated by circular silicon steel sheets. The special rotor structure makes the dq-axis inductances of the synchronous reluctance motor differ greatly, so the synchronous reluctance motor has strong saliency and only has reluctance torque.

[0003] The drive of a high-performance synchronous reluctance motor and the realization of field-oriented control rely on accurate rotor position information. The control with a position sensor relies on installing a mechanical position sensor at the motor shaft end to obtain rotor position information, but it is vulnerable to electromagnetic interference, increases the system cost, and is not conducive to application in industrial production. The adoption of sensorless control technology has become an inevitable trend in fields such as transportation, aerospace, industrial manufacturing, and household appliances. It eliminates the hardware cost of position sensors such as incremental encoders or resolvers, improves the reliability of motor operation, and has broad prospects in industrial applications. Sensorless control technology is mainly divided into fundamental wave model algorithms and salient pole model algorithms. According to the fundamental wave model of the motor, the back electromotive force or magnetic flux containing rotor position information is estimated, and then the rotor position is obtained from it. Such methods are not applicable to the low-speed range because the back electromotive force amplitude is too small and the signal-to-noise ratio is too low at low speed to extract effective position information. The high-frequency injection method in the salient pole model can effectively solve the above problems and is suitable for the application of synchronous reluctance motors in the low-speed range. However, there are some inherent problems in the positionless control algorithm, such as low angle tracking accuracy, limited dynamic performance, excessive preprocessing tests required, and poor robustness. In order to meet the requirements of industrial production applications for high-performance motors and drive systems, in-depth research should be carried out on the control methods and problems generated by synchronous reluctance motors without a sensor. Summary of the Invention

[0004] The present invention aims to solve the problems of the existing sensorless control method for synchronous reluctance motors, such as low angle tracking accuracy, limited dynamic performance, excessive preprocessing tests required, and poor robustness, which cannot meet the requirements of industrial production applications for high-performance motors and drive systems. Now, a low-speed sensorless control method for a synchronous reluctance motor based on high-frequency injection is provided.

[0005] A low-speed sensorless control method for a synchronous reluctance motor based on high-frequency injection includes the following steps:

[0006] Step 1: During the operation of the synchronous reluctance motor, inject high-frequency square-wave voltage vectors into four consecutive PWM cycles of the synchronous reluctance motor in the estimated synchronous rotating coordinate system, so that the current of the synchronous reluctance motor changes after each injection of the high-frequency square-wave voltage vector. The difference between the high-frequency square-wave voltage vectors injected in two adjacent PWM cycles is parallel to the d-axis or q-axis of the estimated synchronous rotating coordinate system. The four high-frequency square-wave voltage vectors are symmetric and the synthetic vector is zero;

[0007] Step 2: During the operation of the synchronous reluctance motor, collect the ABC three-phase current signals, and transform the ABC three-phase current signals into the estimated synchronous rotating coordinate system to obtain the estimated current values of the d-axis and q-axis respectively;

[0008] Step 3: Assume that in two adjacent PWM cycles, inject any two high-frequency square-wave voltage vectors u dq,ix and u dq,iy into the dq-axis of the synchronous reluctance motor in the estimated synchronous rotating coordinate system, so that the current change amounts ξ dqx and ξ dqy generated by the synchronous reluctance motor are obtained, and a current increment model is constructed:

[0009]

[0010] where, Δξ xy,dq is the difference between ξ dqx and ξ dqy , Δξ xy,d and Δξ xy,q are the d-axis and q-axis components of Δξ xy,dq respectively, is the amplitude of the synthetic vector of u dq,ix and u dq,iy , γ xy is the phase angle of the synthetic vector of u dq,ix and u dq,iy , T s is the current sampling period, is the rotor angle estimation error, L Δ is the half-difference inductance of the dq-axis, L Σ is the average inductance of the dq-axis, L d is the d-axis inductance, L q is the q-axis inductance, L dq is the mutual inductance of the dq-axis;

[0011] Step 4: Substitute the high-frequency square-wave voltage vector described in Step 1 and the corresponding current change amount into the current increment model described in Step 3 to obtain Δξ xy,d and Δξ xy,q respectively. x and y represent the serial numbers of two adjacent PWM cycles, x, y = 1, 2, 3, 4, and use Δξxy,d and Δξ xy,q Construct intermediate signals A, B, and C as follows:

[0012]

[0013]

[0014]

[0015] Step Five: Obtain the identification equations of L q and L dq respectively through the intermediate signals A, B, and C:

[0016]

[0017]

[0018] where m = u in T s , u in is the amplitude of the high-frequency square-wave voltage vector in Step One, and are the estimated values of L q and L dq identified from the intermediate signals A, B, and C respectively;

[0019] Step Six: Obtain the d-axis inductance L d according to the self-tuning technology of the synchronous reluctance motor, and substitute the d-axis inductance L d into the identification equations of L q and L dq to obtain and

[0020] Step Seven: Construct the rotor angle error signal ε:

[0021]

[0022] Step Eight: Normalize the rotor angle error signal ε using the normalization coefficient K b to obtain the estimated rotor angle error

[0023] Step Nine: Perform PI regulation on the estimated rotor angle error to obtain the estimated rotor speed Integrate the estimated rotor speed to obtain the estimated rotor angle Use and as the FOC control signals of the synchronous reluctance motor to achieve sensorless control of the synchronous reluctance motor at low speeds.

[0024] Furthermore, the four high-frequency square-wave voltage vectors in the above step one are as follows:

[0025]

[0026] After injecting the four high-frequency square-wave voltage vectors, the current change amounts generated by the synchronous reluctance motor are respectively as follows:

[0027]

[0028] Among them, are respectively the currents generated by injecting u dq,i1 , u dq,i2 , u dq,i3 , in the k-th FOC control period, is the current generated by injecting u dq,i4 in the (k - 1)-th FOC control period.

[0029] Furthermore, in the above step two, the d-axis and q-axis estimated current values are obtained according to the following formula:

[0030]

[0031] Among them, i a , i b and i c are respectively the current signal sampling values of the ABC three phases of the synchronous reluctance motor, is the zero-sequence component, θ e is the angle between the d-axis in the estimated synchronous rotating coordinate system and the A-phase in the natural coordinate system, and are respectively the d-axis and q-axis estimated current values in the estimated synchronous rotating coordinate system.

[0032] Furthermore, in the above step three, it is assumed that in two adjacent PWM periods, any two high-frequency square-wave voltage vectors u dq,ix and u dq,iy are injected into the dq axes of the synchronous reluctance motor in the estimated synchronous rotating coordinate system. Then, the current change amounts generated by the synchronous reluctance motor after injecting u dq,ix and u dq,iy are as follows:

[0033]

[0034]

[0035] Among them, is the admittance equation of the estimated synchronous rotating coordinate system, R s is the stator resistance, is the dq-axis estimated current, and are the estimated current values on the d-axis and q-axis in the estimated synchronous rotating coordinates respectively, and ω is the rotor angular velocity.

[0036] Furthermore, the admittance equation of the above-mentioned estimated synchronous rotating coordinate system has the following expression:

[0037]

[0038] where

[0039] Furthermore, the per-unitization coefficient K mentioned in Step 8 above b has the following expression:

[0040]

[0041] Furthermore, the above-mentioned and has the following expression:

[0042]

[0043] where k p and k i are the proportional and integral controller gains respectively.

[0044] The synchronous reluctance motor sensorless control method based on high-frequency injection according to the present invention performs online identification of inductance parameters during the operation of the synchronous reluctance motor, realizing simultaneous observation of the angular position and inductance parameters. In addition, the four-vector high-frequency square-wave voltage injection method is adopted, which minimizes the computational load and reduces the influence of the injected voltage pulse on the sensorless control system. In summary, the present invention can perform vector control of the synchronous reluctance motor without using a position sensor, eliminating the hardware cost of position sensors such as incremental encoders or resolvers, reducing the complexity of motor installation, improving the reliability of motor operation, and further improving the accuracy of angular observation and the reliability of sensorless control. The present invention is applicable to fields such as electric forklifts, fans, and water pumps. Description of the Drawings

[0045] Figure 1 is the overall block diagram of the synchronous reluctance motor sensorless control method, where SVPWM is the space vector modulation module, SynRM is the synchronous reluctance motor, A, B, and C are intermediate signals, and LPF is the low-pass filter;

[0046] Figure 2 is the injection voltage pattern diagram of the four-vector high-frequency square-wave signal;

[0047] Figure 3 For the four-vector high-frequency square-wave signal and the vector synthesis signal, four symmetric voltage vectors are sequentially injected into the dq axes in the estimated axis system in four consecutive PWM cycles;

[0048] Figure 4 is a coordinate system, where uxy is the difference between the externally applied voltage vectors in two adjacent PWM cycles.

[0049] Figure 5 is the block diagram of the online inductance parameter identification control algorithm:

[0050] Figure 6 is the identification result of L from 50% load to 100% load q and L dq ;

[0051] Figure 7 is the simulation result of the true angle, estimated angle, and angle error of the synchronous reluctance motor rotor during the period from 50% load to 100% load. Specific embodiments

[0052] The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all the embodiments. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts shall fall within the protection scope of the present invention. It should be noted that, without conflict, the embodiments in the present invention and the features in the embodiments may be combined with each other.

[0053] The positionless control algorithm of the synchronous reluctance motor is affected by the inductance accuracy. During the operation of the motor, the motor parameters will change with the change of its operating conditions. Especially for the synchronous reluctance motor, its rotor is formed by laminating special-shaped silicon steel sheets to form a structure with a high salient pole ratio, and it requires the stator excitation current to magnetize the rotor, which is prone to magnetic saturation, resulting in the dq-axis inductance parameters becoming smaller as the current increases. And the control performance of the synchronous reluctance motor and some control algorithms such as rotor position estimation are dependent on the motor parameters. Therefore, it is very important to identify the inductance parameters of the synchronous reluctance motor. In the dq-axis system, the relationship between the inductance parameters of the synchronous reluctance motor and the current is very complex, which makes the control of the SynRM complex. In order to improve the estimation accuracy of the rotor position information of the synchronous reluctance motor and enhance the robustness of the positionless control algorithm to parameter changes, it is of great significance and application value to study the sensorless control method based on high-frequency injection with inductance parameter identification. For this reason, this embodiment refers to Figures 1 to 7, this embodiment uses a 3KW synchronous reluctance motor with a rated torque of 9.55 N.m, a rated speed of 3000 r / min, a pole pair number of 2, a rated current of 8.2 A, a bus voltage of 540V, and the amplitude u of the injected high-frequency square-wave voltage vector in is 80V.

[0054] The method for sensorless control of a synchronous reluctance motor at low speed based on high-frequency injection in this embodiment includes the following steps:

[0055] Step 1: During the operation of the synchronous reluctance motor, inject high-frequency square-wave voltage vectors into four consecutive PWM cycles of the synchronous reluctance motor in the estimated synchronous rotating coordinate system (hereinafter referred to as the estimated coordinate system), so that the current of the synchronous reluctance motor changes after each high-frequency square-wave voltage vector is injected. The difference between the high-frequency square-wave voltage vectors injected in two adjacent PWM cycles is parallel to the d-axis or q-axis of the estimated coordinate system, and the four high-frequency square-wave voltage vectors are symmetric and the synthesized vector is zero.

[0056] As Figure 2 shown, the four high-frequency square-wave voltage vectors are respectively expressed as:

[0057]

[0058] The synthesized vectors of two adjacent injected high-frequency square-wave voltage vectors are respectively expressed as:

[0059] u dq,i1 and u dq,i2 's synthesized vector u dq,i12 = [u in 0] T , u dq,i2 and u dq,i3 's synthesized vector u dq,i23 = [0 u in T ,

[0060] u dq,i3 and u dq,i4 's synthesized vector u dq,i34 = [-u in 0] T , u dq,i4 and u dq,i1 's synthesized vector u dq,i41 = [0 -u in T .

[0061] As Figure 3 shown, after injecting the four high-frequency square-wave voltage vectors, the current change amount generated by the synchronous reluctance motor is:

[0062] ​​

[0063] The current increment difference is:

[0064]

[0065] Wherein, are respectively the currents generated by injecting u dq,i1 , u dq,i2 , u dq,i3 , u dq,i4 in the k-th FOC (vector control) control period, is the current generated by injecting u dq,i4 in the (k - 1)-th FOC control period.

[0066] Step 2, during the operation of the synchronous reluctance motor, sample the ABC three-phase current signals of the synchronous reluctance motor and transform them into the estimated coordinate system to obtain the dq-axis estimated current signals in the estimated coordinate system.

[0067] The specific expression is:

[0068]

[0069] Wherein, i a , i b and i c are respectively the sampled values of the ABC three-phase current signals of the synchronous reluctance motor, is the zero-sequence component, θ e is the angle between the d-axis in the estimated coordinate system and the A-phase in the natural coordinate system, and are respectively the d-axis and q-axis estimated current values in the estimated coordinate system.

[0070] Step 3, combine the d-axis and q-axis estimated current values in the estimated coordinate system and four high-frequency square-wave voltage vectors, calculate the intermediate signals A, B, C through the current increment model in the estimated coordinate system, and finally obtain the relationships of the d-axis inductance L d respectively and the q-axis inductance L q and the mutual inductance L dq between the dq axes. Specifically as follows:

[0071] The voltage equation of the synchronous reluctance motor on the dq axes is:

[0072]

[0073] Wherein, u dq = [u d , u q T and i dq = [i d , i q T ​​are the dq-axis voltages and currents respectively, and ω is the rotor angular velocity. is an orthogonal rotation matrix, R s is the stator resistance,

[0074] The dq-axis voltage equations in the estimated coordinate system are:

[0075]

[0076] where, and are the estimated dq-axis voltages and estimated currents respectively, is the half-difference inductance matrix, L Δ =(L d -L q ) / 2 is the half-difference inductance, L Σ =(L d +L q ) / 2 is the average inductance, is the rotor angle estimation error, is the estimated rotor speed.

[0077] The position estimation method based on high-frequency square-wave voltage vector injection needs to know the current increment information. Therefore, the dq-axis voltage equations in the estimated coordinate system can also be rewritten as current difference expressions:

[0078]

[0079] where, is the admittance equation in the estimated coordinate system,

[0080] From Figure 4 it can be seen that assuming two high-frequency square-wave voltage vectors u dq,ix and u dq,iy are respectively injected into the dq axes of the synchronous reluctance motor in the estimated synchronous rotating coordinate system in two adjacent PWM periods, and the synthetic vector of u dq,ix and u dq,iy is u dq,ixy , the current change amount generated in two adjacent PWM periods can be written as:

[0081]

[0082]

[0083] where, T s is the current sampling period, ξ dqx and ξ dqy are respectively the injected udq,ix and u dq,iy The generated current change amount.

[0084] Subtracting Equation (5) from Equation (6) can eliminate the influence of the back electromotive force term and the resistance term on the calculation of the current increment, and obtain the current increment model, that is:

[0085]

[0086] where γ xy and are the phase angle and amplitude of u dq,ixy respectively, and Δξ xy,d and Δξ xy,q are the d-axis and q-axis components of Δξ xy,dq respectively.

[0087] Substituting the synthesis vector of the two adjacent injected high-frequency square-wave voltage vectors and the current increment difference described in Step 1 into Equation (7), x, y = 1, 2, 3, 4, a complete description of the increment current can be obtained:

[0088]

[0089]

[0090]

[0091]

[0092]

[0093]

[0094]

[0095]

[0096] Based on the above increment current, intermediate signals A, B, and C are obtained:

[0097]

[0098]

[0099]

[0100] Step 4. No matter what injection method and signal demodulation method are adopted, only the above three equations or variants of these three equations can be obtained, and the unknown parameters are the d-axis inductance L d , q-axis inductance L q , dq-axis mutual inductance L dqand the rotor angle error signal ε. Therefore, new information must be introduced to determine all parameters. If L is obtained by looking up a table d , L q or L dq is any one of them, then the other three parameters can be calculated. In this embodiment, an accurate magnetic saturation model representing the d-axis inductance L d is obtained according to the self-tuning technology of the motor, and the relationships between L d and L q and L dq are calculated through the intermediate signals A, B, and C respectively. Then, the identification equations for L dq and L q are as follows:

[0101]

[0102]

[0103] where m = u in T s , and are the estimated values of L q and L dq identified from the intermediate signals A, B, and C respectively.

[0104] Step 5, estimate the rotor angle when the inductance parameters are known. To avoid the influence of cross saturation on the rotor angle estimation error at steady state, a phase-locked loop structure is adopted to construct the rotor angle error term ε. The rotor angle error term ε is normalized by the normalization coefficient K b to obtain the rotor angle estimation error and then passes through a PI regulator to obtain the rotor estimated speed which is further integrated to obtain the rotor estimated angle The and are fed back to the speed closed-loop and current closed-loop controls of the synchronous reluctance motor to achieve sensorless control of the synchronous reluctance motor. Specifically as follows:

[0105] Construct the rotor angle error signal ε as:

[0106]

[0107] where

[0108] When the angle error is very small then the rotor angle error signal ε is approximately equal to:

[0109]

[0110] Select the per-unit coefficient K b It is:

[0111]

[0112] The rotor angle error signal ε passes through the per-unit coefficient K b The per-unit value gives the estimated rotor angle error Then, through the PI regulator, the estimated rotor speed is obtained Integrating it further gives the estimated rotor angle

[0113]

[0114] where, k p and k i are the proportional and integral controller gains respectively

[0115] So far, we have obtained and and directly feedback them into the FOC control as angle and speed feedback to achieve sensorless control of the synchronous reluctance motor

[0116] The following is the validation of the effectiveness of this embodiment

[0117] First, the amplitude of the four-vector voltage signal injected into the estimated axis system is 80V, and the frequency is 2.5kHz. The motor speed is 500r / min Figure 6 The L from 50% load to 100% load is respectively shown q and L dq Identification results. It can be seen that under different working conditions, the identified inductance values are always close to the standard values

[0118] Secondly Figure 7 The simulation results of the true angle, estimated angle and angle error of the synchronous reluctance motor rotor during the period from 50% load to 100% load are shown. From Figure 7 (b), it can be seen that the angle error meets the requirements of motor sensorless observation, and the results verify the effectiveness of the present invention

[0119] Based on the injection of a four-vector high-frequency square-wave voltage, this embodiment fully obtains the current increment information, calculates the complete description of the incremental current on the model of the current increment difference, thereby obtaining an effective equation for inductance parameter identification, and performs online identification of inductance parameters. The traditional sensorless algorithm for synchronous reluctance motors is affected by the inductance accuracy, and its magnetic saturation phenomenon will cause changes in inductance parameters. Most of the inductance parameters in the existing sensorless control algorithms for synchronous reluctance motors are constant values or obtained through offline self-learning, without considering the influence of magnetic saturation phenomenon on the position estimation accuracy, which will increase the angle estimation error to a certain extent. This embodiment performs online identification of inductance parameters in the sensorless control algorithm of a synchronous reluctance motor with four-vector injection, improving the rotor position estimation accuracy and the robustness of the sensorless control algorithm to parameter changes.

[0120] Although the present invention has been described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the present invention. Therefore, it should be understood that many modifications can be made to the exemplary embodiments, and other arrangements can be designed, as long as they do not depart from the spirit and scope of the present invention as defined by the appended claims. It should be understood that different dependent claims and the features described herein can be combined in a manner different from that described in the original claims. It should also be understood that the features described in connection with a single embodiment can be used in other described embodiments.

Claims

1. A sensorless control method for a synchronous reluctance motor at low speed based on high-frequency injection, characterized in that, Including the following steps: Step 1: During the operation of the synchronous reluctance motor, high-frequency square-wave voltage vectors are respectively injected into four consecutive PWM cycles of the synchronous reluctance motor in the estimated synchronous rotating coordinate system, so that the current of the synchronous reluctance motor changes after each high-frequency square-wave voltage vector injection. The difference between the high-frequency square-wave voltage vectors injected in two adjacent PWM cycles is parallel to the d-axis or q-axis of the estimated synchronous rotating coordinate system, and the four high-frequency square-wave voltage vectors are symmetric and the synthesized vector is zero; Step 2: During the operation of the synchronous reluctance motor, collect the ABC three-phase current signals, and transform the ABC three-phase current signals into the estimated synchronous rotating coordinate system to obtain the estimated current values of the d-axis and q-axis respectively; Step 3: Assume that in two adjacent PWM cycles, any two high-frequency square-wave voltage vectors u dq,ix and u dq,iy are injected into the dq axes of the synchronous reluctance motor in the estimated synchronous rotating coordinate system, so that the current change amounts ξ dqx and ξ dqy generated by the synchronous reluctance motor are obtained, and a current increment model is constructed: where, Δξ xy,dq is the difference between ξ dqx and ξ dqy ; Δξ xy,d and Δξ xy,q are the d-axis and q-axis components of Δξ xy,dq respectively; is the magnitude of the resultant vector of u dq,ix and u dq,iy ; γ xy is the phase angle of the resultant vector of u dq,ix and u dq,iy ; T s is the current sampling period; is the rotor angle estimation error; L Δ is the dq-axis half-difference inductance; L Σ is the dq-axis average inductance; L d is the d-axis inductance; L q is the q-axis inductance; L dq is the dq-axis mutual inductance; Step 4: Substitute the high-frequency square-wave voltage vector and the corresponding current change amount described in Step 1 into the current increment model described in Step 3 to obtain Δξ xy,d and Δξ xy,q , where x and y represent the serial numbers of two adjacent PWM cycles, x, y = 1, 2, 3, 4, and use Δξ xy,d and Δξ xy,q to construct intermediate signals A, B, and C: Step Five: Obtain the identification equations of L q and L dq respectively through the intermediate signals A, B, and C: where m = u in T s , u in is the amplitude of the high-frequency square-wave voltage vector in the first step, and are the estimated values of L q and L dq identified from the intermediate signals A, B, and C, respectively; Step 6: Obtain the d-axis inductance L according to the self-tuning technology of the synchronous reluctance motor d , and substitute the d-axis inductance L d into the identification equations of L q and L dq to obtain and Step 7: Construct the rotor angle error signal ε: Step VIII: Using the per-unit coefficient K b Perform per-unit conversion on the rotor angle error signal ε to obtain the estimated rotor angle error Step 9: Perform PI regulation on the rotor angle estimation error to obtain the estimated rotor speed Integrate the estimated rotor speed to obtain the estimated rotor angle Use and as the FOC control signals of the synchronous reluctance motor to achieve sensorless control of the synchronous reluctance motor at low speeds.

2. A sensorless control method for a synchronous reluctance motor based on high-frequency injection according to claim 1, characterized in that, The four high-frequency square-wave voltage vectors in Step 1 are as follows: After injecting the four high-frequency square-wave voltage vectors, the current change amounts generated by the synchronous reluctance motor are as follows: Among them, are the currents generated by injecting u dq,i1 , u dq,i2 , u dq,i3 , u dq,i4 respectively in the k-th FOC control period, is the current generated by injecting u dq,i4 in the (k - 1)-th FOC control period.

3. A sensorless control method for a synchronous reluctance motor based on high-frequency injection according to claim 1, characterized in that In Step 2, the estimated current values of the d-axis and q-axis are obtained according to the following formula: wherein, i a , i b and i c are respectively the sampled values of the current signals of the ABC three phases of the synchronous reluctance motor, is the zero-sequence component, θ e is the angle between the d-axis in the estimated synchronous rotating coordinate system and the A-phase in the natural coordinate system, and are respectively the estimated current values of the d- and q-axes in the estimated synchronous rotating coordinate system.

4. A sensorless control method for a synchronous reluctance motor based on high-frequency injection according to claim 1, characterized in that In step 3, assuming that in two adjacent PWM cycles, any two high-frequency square-wave voltage vectors u dq,ix and u dq,iy are injected into the dq axes of the synchronous reluctance motor in the estimated synchronous rotating coordinate system, then the current variation generated by the synchronous reluctance motor after injecting u dq,ix and u dq,iy is as follows: Among them, For estimating the admittance equation in the synchronous rotating coordinate system, R s is the stator resistance, is the estimated current in the dq axes, and are the estimated current values in the d and q axes under the estimated synchronous rotating coordinate respectively, ω is the rotor angular velocity, 5. A sensorless control method for a synchronous reluctance motor based on high-frequency injection according to claim 4, characterized in that The admittance equation of the estimated synchronous rotating coordinate system The expression is as follows: Among them, 6. A sensorless control method for a synchronous reluctance motor based on high-frequency injection according to claim 1, characterized in that The per-unit coefficient K described in Step 8 b The expression is as follows:

7. A sensorless control method for a synchronous reluctance motor based on high-frequency injection according to claim 1, characterized in that and The expressions are as follows: where k p and k i are the proportional and integral controller gains, respectively.

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