System for detecting optically chirped distance using a square-wave digital chirp signal

Abstract

The present invention relates to a system for detecting an optically chirped distance using a square-wave digital chirp signal. An optical detection and ranging lidar system includes: a laser source configured to generate a carrier wave; an equalizer configured to modulate the amplitude of a chirp signal; a modulator configured to modulate the frequency of the carrier wave using the chirp signal; one or more scanning optics configured to generate a transmitted signal using the modulated carrier wave and transmit the transmitted signal; and a processor configured to determine a Doppler shift of an object based on a return signal and a reference signal; wherein the return signal is received from the object in response to the transmitted signal.

Description

[0001] This application is a divisional application of the Chinese patent application "Method and System for Optical Chirp Distance Detection Using Square Wave Digital Chirp Signals" with Chinese application number 201880044454.5, international application date of July 27, 2018, PCT application number PCT / US2018 / 044007, which entered the Chinese national phase on December 31, 2019.

[0002] Cross - Reference to Related Applications

[0003] This application claims the benefit of U.S. Application No. 15 / 661,377, filed on July 27, 2017, which is hereby incorporated by reference in its entirety as if fully set forth herein. Technical Field

[0004] This application relates to a system for detecting an optical chirp distance using a square wave digital chirp signal. Background Art

[0005] Optical detection of distances for optical detection and ranging, commonly referred to by the mnemonic Light Detection and Ranging (LIDAR), is used in a variety of applications from altimetry to imaging to collision avoidance. Compared to conventional microwave ranging systems (e.g., Radio Detection and Ranging (RADAR)), LIDAR provides finer scale range resolution at smaller beam sizes. Optical detection of distances can be accomplished by several different techniques, including direct ranging based on the round - trip time of an optical pulse to a target, and chirp detection based on the frequency difference between a transmitted chirped optical signal and a return signal scattered from the target.

[0006] To achieve acceptable ranging accuracy and detection sensitivity, direct - ranging LIDAR systems use short - pulse lasers with low pulse repetition rates and extremely high pulse peak powers. The high pulse power can cause rapid degradation of optical components. Chirp LIDAR systems use long optical pulses with relatively low peak optical powers. In this configuration, the ranging accuracy depends on the chirp bandwidth rather than the pulse duration, and thus excellent ranging accuracy can still be obtained.

[0007] The modulation of an optical carrier with a broadband radio frequency (RF) electrical signal has achieved useful chirped optical bandwidths. Recent advances in chirped LIDAR include using the same modulated optical carrier as a reference signal that is combined with the return signal at an optical detector to produce a relatively low beat frequency in the resulting electrical signal that is proportional to the difference in frequency between the reference and return optical signals. This beat frequency detection of the frequency difference at the detector is known as heterodyne detection. It has several advantages known in the art, such as the advantage of using off-the-shelf and inexpensive RF components. Except for terms inconsistent with those used herein, recent work described in Patent 7,742,152 shows a novel and simpler arrangement of optical components using an optical signal separated from the transmitted optical signal as the reference optical signal. This arrangement is referred to as homodyne detection in that patent. SUMMARY OF THE INVENTION

[0008] In a conventional chirped LIDAR system, an RF source is provided to generate an RF electrical signal at a bandwidth (e.g., 500 megahertz (MHz, 1 MHz = 10 6 hertz) to 10 gigahertz (\GHz, 1 GHz = 10 9 Hz)) to modulate the optical carrier. The present inventors have recognized that RF electrical signals generated by conventional RF sources have significant drawbacks. For example, the bandwidth of RF electrical signals generated by a digital-to-analog converter (DAC) or a direct digital synthesis (DDS) device is typically maximum at about 4 GHz, which limits the ranging accuracy of the LIDAR system. In addition, DAC or DDS devices are bulky, consuming valuable space in packages with limited size and weight and being expensive (e.g., $200,000). Further, although the bandwidth of RF electrical signals generated by DAS or DDS devices can be increased to about 10 GHz, this requires additional steps (e.g., RF multiplication, optical multiplication) that require high power and introduce undesirable characteristics into the signal. A device and method are provided that address the drawbacks of conventional RF signals provided by conventional RF sources in chirped LIDAR systems. The device and method involve using a field-programmable gate array (FPGA) transceiver for digital chirp generation in a chirped LIDAR system.

[0009] In a first set of embodiments, an apparatus for using a square wave digital chirp signal for optical chirp ranging is provided. The apparatus includes a laser source for emitting an optical signal. The apparatus further includes an RF waveform generator for generating an input digital chirp signal based on the square wave digital chirp signal. The apparatus further includes a modulator for modulating the frequency of the optical signal based on the input digital chirp signal. Additionally, the apparatus includes a beam splitter that splits the optical signal into a transmitted optical signal and a reference optical signal. The apparatus further includes a detector that combines the reference optical signal and a return optical signal based on the transmitted optical signal backscattered by an object. The detector is configured to generate an electrical output signal based on the combined reference optical signal and return optical signal. The apparatus further includes a processor that determines the distance to the object based on a characteristic of performing a Fourier transform on the electrical output signal.

[0010] In a second set of embodiments, a method for using a square wave digital chirp signal for optical chirp ranging is provided. The method includes: emitting an optical signal from a laser source; and modulating the frequency of the optical signal using a modulator based on an input digital chirp signal from an RF waveform generator. The input digital chirp signal is based on the square wave digital chirp signal. Additionally, the method includes splitting the optical signal into a transmitted optical signal and a reference optical signal using a beam splitter. The method further includes combining the reference optical signal and a return optical signal at a detector. The return optical signal is based on the transmitted optical signal backscattered by an object. Additionally, the method includes generating an electrical output signal from the detector based on the combining step. The method further includes determining the distance to the object using a processor based on a characteristic of performing a Fourier transform on the output electrical output signal.

[0011] Other aspects, features, and advantages will be readily apparent from the following detailed description by simply illustrating a number of specific embodiments and implementations, including the best mode contemplated for carrying out the invention. Other embodiments can have other and different features and advantages, and can be modified in several obvious respects, all without departing from the spirit and scope of the invention. Accordingly, the drawings and description are to be regarded as illustrative in nature rather than restrictive. BRIEF DESCRIPTION OF THE DRAWINGS

[0012] In the illustrations of the drawings, various embodiments are shown by way of example and not by way of limitation, where like reference numerals indicate like elements, and where:

[0013] Figure 1A is a set of graphs showing an exemplary optical chirp measurement of distance according to an embodiment;

[0014] Figure 1B is a graph showing an exemplary measurement of a beat frequency indicating distance generated by de-chirping according to an embodiment.

[0015] Figure 2A and Figure 2B is a block diagram showing exemplary components of a high - resolution LIDAR system according to various embodiments;

[0016] Figure 3A is a block diagram showing an exemplary component of an RF waveform generator of a LIDAR system according to an embodiment of Figure 2A ;

[0017] Figure 3B is a block diagram showing an exemplary component of an RF conditioning of an RF waveform generator according to an embodiment of Figure 3A ;

[0018] Figure 3C is a block diagram showing an exemplary component of an RF waveform generator of a LIDAR system according to an embodiment of Figure 2B ;

[0019] Figure 4A is a graph showing an example of an input digital chirp signal used in a conventional LIDAR system according to an embodiment;

[0020] Figure 4B is a graph showing an example of the spectrum of the input digital chirp signal of Figure 4A according to an embodiment;

[0021] Figure 5A is a graph showing an example of a square - wave digital chirp signal used in a LIDAR system according to an embodiment in Figure 2A and Figure 2B ;

[0022] Figure 5B is a graph showing an example of the spectrum of a square - wave digital chirp signal of Figure 5A according to an embodiment;

[0023] Figure 6A is a graph showing an example of the transfer function of a square - wave digital chirp signal of an RF waveform generator of Figure 3A according to an embodiment;

[0024] Figure 6B is a graph showing an example of the transfer function of an equalizer of an RF conditioning component of Figure 3B according to an embodiment;

[0025] Figure 6C is a graph showing an example of a synthesis function of an input digital chirp signal output from an RF waveform generator of Figure 3A according to an embodiment;

[0026] FIG. 7 is a graph showing an exemplary measurement of the beat frequency generated by de-chirping in a conventional LIDAR system using the input digital chirp signal of FIG. 4A according to an embodiment;

[0027] Figure 8 is a graph showing an exemplary measurement of the beat frequency generated by de-chirping in a LIDAR system using Figure 5A a square wave digital chirp signal according to an embodiment;

[0028] Figure 9 is a flowchart showing an exemplary method for using a square wave digital chirp signal for optical chirp ranging according to an embodiment;

[0029] Figure 10 is a block diagram of a computer system on which embodiments of the present invention can be implemented; and

[0030] Figure 11 shows a chipset on which embodiments of the present invention can be implemented. DETAILED DESCRIPTION

[0031] Described is a method, apparatus, and system for using a square wave digital chirp signal for optical chirp ranging and a computer-readable medium. In the following description, for the purpose of explanation, numerous specific details are set forth in order to provide a thorough understanding of the present invention. However, it will be apparent to one of ordinary skill in the art that the present invention may be practiced without these specific details. In other instances, well-known structures and devices are shown in block diagram form in order to avoid unnecessarily obscuring the present invention.

[0032] While the numerical ranges and parameters setting forth a broad scope are approximations, the numerical values set forth in the specific non-limiting examples are reported as precisely as possible. Any numerical value, however, inherently contains certain errors necessarily resulting from the standard deviation found in their respective testing measurements at the time of writing this document. In addition, unless clearly stated otherwise from the context, the numerical values presented herein have an implied precision given by the least significant digit. Thus, the value 1.1 implies a value ranging from 1.05 to 1.15. The term "about" is used to indicate a broader range centered around a given value and, unless clearly known from the context, implies a broader range around the least significant digit, such as "about 1.1" implies a range from 1.0 to 1.2. If the least significant digit is not clear, the term "about" implies twice, for example, "about X" implies a value within the range of 0.5X to 2X. For example, about 100 implies a value within the range from 50 to 200. In addition, it should be understood that all ranges disclosed herein cover any and all sub-ranges subsumed therein. For example, the range "less than 10" may include any and all sub-ranges between (and including) the minimum value of zero and the maximum value of 10, that is, any and all sub-ranges with a minimum value equal to or greater than zero and a maximum value equal to or less than 10, such as 1 to 4.

[0033] Some embodiments of the present invention are described below in the context of using broadband RF electrical signals to modulate one or more optical carriers in a chirped LIDAR system. However, the present invention is not limited to this context. In other embodiments, the present invention may be used in the context of using a broadband waveform generator to generate waveforms for: ranging, including phase encoding for phase shift keying, on-off keying, frequency shift keying, or broadband noise.

[0034] 1. Overview of Chirped Detection

[0035] Figure 1A Are a set of curves 110, 120, 130, 140 showing exemplary optical chirped measurements of distance according to an embodiment. The horizontal axis 112 is the same for all four curves and indicates time in arbitrary units, approximately milliseconds (ms, 1 ms = 10- 3 seconds). Curve 110 indicates the power of the light beam used as the transmitted optical signal. The vertical axis 114 in Curve 110 represents the power of the transmitted signal in arbitrary units. Trace 116 indicates that the power supply is turned on for a finite pulse duration, starting from time 0, τ. Curve 120 indicates the frequency of the transmitted signal. The vertical axis 124 represents the frequency transmitted in arbitrary units. Trace 126 shows that the frequency of the pulse increases from f1 to f2 within the pulse duration τ, and thus the bandwidth is B = f2 - f1. The rate of frequency change is (f2 - f1) / τ.

[0036] As shown in graph 120, the return signal is depicted in graph 130 having a horizontal axis 112 indicating time and a vertical axis 124 indicating frequency. The chirp 126 of graph 120 is also plotted as a dashed line on graph 130. The first return signal is given by trace 136a, which is just the transmitted reference signal with reduced intensity (not shown) and delayed by Δt. When the return signal is received from an external object after traveling a distance of 2R, where R is the distance to the target, the return signal starts at the delay time Δt given by 2R / c, where c is the speed of light in the medium (about 3×10 8 meters per second, m / s). During this time, the amount of frequency change depends on the distance and is called f R and is given by multiplying the rate of frequency change by the delay time. This is given by Equation 1a.

[0037] f R =(f2 - f1) / τ * 2R / c = 2BR / cτ (1a)

[0038] f R The value is measured by the frequency difference between the transmitted signal 126 and the return signal 136a in a time-domain mixing operation called de-chirping. Thus, the distance R is given by Equation 1b.

[0039] R = f R cτ / 2B (1b)

[0040] Of course, if the return signal arrives after the pulse has been fully transmitted, that is, if 2R / c is greater than τ, then Equations 1a and 1b are invalid. In this case, the reference signal is delayed by a known or fixed amount to ensure that the return signal overlaps with the reference signal. The fixed or known delay time of the reference signal is multiplied by the speed of light c to give an additional distance that is added to the distance calculated from Equation 1b. Although the absolute distance may deviate due to the uncertainty of the speed of light in the medium, this is a nearly constant error, and the relative distance based on the frequency difference remains very accurate.

[0041] In some cases, the location irradiated by the transmitted beam encounters two or more different scatterers within different ranges, such as the front and back of a translucent object, or the closer and farther parts of an object at variable distances from the LIDAR, or two separate objects within the irradiated location. In this case, a second attenuated intensity and a signal with a different delay will also be received, which is indicated by trace 136b on graph 130. This will have different f R measurement values that give different ranges using Equation 1b. In some cases, multiple return signals are received.

[0042] Graph 140 depicts the difference frequency f between the first return signal 136a and the reference chirp 126.R 。The horizontal axis 112 indicates time as in all the other alignment graphs of Figure 1A and the vertical axis 134 indicates the frequency difference on a much expanded scale. The trace 146 depicts the constant frequency f measured during the chirp transmission R which indicates a specific range as given by Equation 1b. If there is a second return signal 136b, it will produce a different and larger f value (not shown) during the de-chirping and thus, a larger range is obtained using Equation 1b. R

[0043] A common method for de-chirping is to direct both the reference optical signal and the return optical signal to the same optical detector. The electrical output signal of the detector is dominated by a beat frequency equal to or otherwise depending on the frequency difference between the two signals converging on the detector. The Fourier transform of this electrical output signal will produce a peak at the beat frequency. This beat frequency is in the radio frequency (RF) range of megahertz (MHz, 1 MHz = 10 6 Hertz = 10 6 cycles / second), rather than in the optical frequency range of terahertz (THz, 1 THz = 10 12 Hertz). Such signals are easily processed by common and inexpensive RF components such as a fast Fourier transform (FFT) algorithm running on a microprocessor or a specially constructed FFT or other digital signal processing (DSP) integrated circuit. In other embodiments, the return signal is mixed with a continuous wave (CW) tone acting as a local oscillator (relative to the chirp as the local oscillator). This results in the detected signal itself being a chirp (or any waveform transmitted). In this case, except for terms inconsistent with the terminology used herein, the detected signal will be matched filtered in the digital domain as described in Kachelmyer 1990. The disadvantage is that the digitizer bandwidth requirements are usually high. Otherwise, the positive aspects of coherent detection are maintained.

[0044] Figure 1B is a graph showing an exemplary measurement of the beat frequency indicating distance generated by de-chirping according to an embodiment. The horizontal axis 152 represents the frequency in megahertz; and the vertical axis represents the return signal power density I T relative to the transmitted power density I R (in decibels) (dB, power in dB = 20 log(I R / Ι T )). The trace 156 is the Fourier transform of the electrical output signal by the optical detector, such as produced by an FFT circuit and based on data published by Adany et al. in 2009. The horizontal position of the peak gives f R, and is used to estimate the range using Equation 1b. Additionally, other characteristics of the peak can be used to describe the returned signal. For example, the power value at the peak is characterized by the maximum value of trace 156, or more generally, by the difference 157 between the peak ( Figure 1B which is approximately -31 dB in Figure 1B ) and the noise floor at the peak shoulder ( Figure 1B which is approximately -50 dB in TM ); and the width of the peak is characterized by the frequency width 158 at half maximum (FWHM) ( TM which is approximately 0.08 MHz in TM ). If there are multiple distinguishable returns, there will be multiple peaks in the FFT of the electrical output of the photodetector, possibly having multiple different power levels and widths. Any method can be used to automatically identify the peaks in the trace, and those peaks are characterized by their position, height, and width. For example, in some embodiments, FFTW or peak detection through the MATLAB - Signal Processing Toolbox available from MATHWORKS TM of Natick, Massachusetts, USA is used. One can also use a custom implementation that relies on FFTW in CUDA and custom peak detection available from NVIDIA

[0045] of Santa Clara, California, USA. The custom implementation has been programmed on a field - programmable gate array (FPGA). Commonly used algorithms are to threshold the range profile and run a centroid algorithm, a peak fitting algorithm (3 - point Gaussian fitting), or a non - linear fitting of the peak of a function (such as Gaussian) to more precisely determine the position of the peak. s If the detected object (source) moves with velocity v o and the LIDAR system (observer) moves with velocity v R on the vector connecting the two, the returned signal can undergo Doppler shift, and the detected beat frequency f

[0046] will also shift, which can lead to errors within the detection range. In many cases, the shape of the detected object is identified based on the relative positions of multiple returns. Thus, the shape of the object may be incorrect and may compromise the ability to identify the object.

[0047]

[0048] where c is the speed of light in the medium. It should be noted that if the observer and the source move in the same direction at the same speed along the vector between them, these two frequencies are the same. The difference △f = f' - f between the two frequencies is the Doppler shift D, which, if uncorrected, constitutes an error in distance measurement and is given by Equation 2b.

[0049]

[0050] It should be noted that the magnitude of the error increases with the increase in the signal frequency f. It should also be noted that for a fixed LIDAR system (ν o = 0), for a target moving at 10 meters per second (ν o = 10), and the frequency of visible light being approximately 500 THz, the magnitude of the error is approximately 16 MHz, which is 75% of the magnitude of f R and is approximately 22 MHz in Figure 1B , resulting in an error of 75% in f R and thus a distance error of 75%. In various embodiments, the Doppler shift error is detected and used to correct the distance.

[0051] 2. Overview of Chirp Detection Hardware

[0052] To depict how the chirp detection method is implemented, some general and specific hardware methods are described. Figure 2A and Figure 2B are block diagrams showing exemplary components of a high - resolution LIDAR system according to various embodiments. In Figure 2A , the laser source 212 emits a carrier 201, which is frequency - modulated in the modulator 214 based on an input from the RF waveform generator 215 to produce a pulse with a bandwidth B and a duration τ. In various embodiments described in more detail later, the electrical input from the RF waveform generator 215 is an input digital chirp signal based on a square - wave digital chirp signal. In one embodiment, the square - wave digital chirp signal is generated by the RF waveform generator 215 and further conditioned by the RF waveform generator 215 to produce the input digital chirp signal. In some of these embodiments, the input digital chirp signal is itself a square - wave digital chirp signal without such conditioning. In some embodiments, the RF waveform generator 215 generates the square - wave digital chirp signal based on square - wave instructions 217 uploaded to the memory of the RF waveform generator 215. In other embodiments, the distance determination module 270 of the processing system 250 controls the RF waveform generator 215, including generating the square - wave digital chirp signal. In an exemplary embodiment, the square - wave instructions are provided in the distance determination module 270.

[0053] The optical splitter 216 splits the modulated optical waveform into an emission signal 205 having most of the energy of the light beam 203 and a reference signal 207 having very little energy, which nonetheless is sufficient to produce a good heterodyne or homodyne interference with the returned light 291 scattered from a target (not shown). In some embodiments, a scanning optical device 218 is used to scan the transmitted light beam at multiple angles to profile any object in its path.

[0054] The reference beam is sufficiently delayed in the reference path 220 such that the scattered light reaches the detector array 230. In some embodiments, the optical splitter 216 is upstream of the modulator 214 and the reference beam 207 is not modulated. In some embodiments, the reference signal is independently generated using a new laser (not shown) and is modulated using a separate modulator (not shown) in the reference path 220 and the input digital chirp signal from the generator 215, respectively. In some embodiments, as described in the following reference Figure 2B , different modulators are used; however, only one laser source 212 is used for both the emission signal and the reference signal to ensure coherence. In various embodiments, from less flexible methods to more flexible methods, the reference is made to arrive with the scattered or reflected field by: 1) placing a mirror in the scene to reflect a portion of the emitted light beam back to the detector array such that the path lengths exactly match; 2) using fiber optic delays to closely match the path lengths and broadcasting the reference beam near the detector array with or without path length adjustment, as Figure 2A shown, to compensate for the phase differences observed or expected within a particular range; or 3) using a frequency shift device (acousto-optic modulator) or a time delay modulated by a local oscillator waveform to produce a separate modulation to compensate for path length mismatches; or some combination. In some embodiments, the target is close enough and the pulse duration is long enough such that the return overlaps sufficiently with the reference signal without delay. In some embodiments, the reference signal 207b is optically mixed with the returned signal 291 at one or more optical mixers 232.

[0055] In some embodiments, the reference path 220 includes a phase modulator that receives an input digital chirp signal from the RF waveform generator 215, where the input digital chirp signal is based on a square wave digital chirp signal. In these embodiments, the input digital chirp signal received at the reference path 220 is independent of the input digital chirp signal received at the modulator 214. In one embodiment, the phase of the reference signal 207 is modulated by the phase modulator based on the input digital chirp signal to advantageously reduce the intensity noise variation between the reference beam 207b and the returned signal 291 at the optical mixer 232.

[0056] In various embodiments, multiple portions of the target scatter corresponding return beams 291 back to the detector array 230 for each scan beam, thereby generating a point cloud based on multiple ranges of corresponding multiple portions of the target illuminated by the multiple beams and the multiple returns. The detector array is a single pair or balanced pair optical detector, or a 1D or 2D array of such optical detectors is arranged in a plane substantially perpendicular to the return beam 291 from the target. The acquisition system 240 records the phase or amplitude of the interface mode, or some combination, multiple times during the pulse duration τ for each detector. The number of time samples per pulse duration affects the lower range. This number is typically a practical consideration based on the pulse repetition rate and the available camera frame rate. The frame rate is the sampling bandwidth, often referred to as the "digitizer frequency". Basically, if X detector array frames are collected during a pulse in a resolution bin with a Y range width, a range of X * Y can be observed. The acquired data is made available to the processing system 250, such as the computer system described below with reference to Figure 10 or the chipset described below with reference to Figure 11 . In some embodiments, the acquired data is a point cloud based on multiple ranges of corresponding multiple portions of the target.

[0057] The distance determination module 270 calculates the distance to the target based on the electrical output signal from the detector array 230. In some embodiments, the distance determination module 270 performs a Fourier transform on the electrical output signal to obtain the spectrum of the output signal and calculates the distance to the target based on one or more characteristics of the spectrum. In one embodiment, the distance determination module 270 is based on Figure 1B one or more characteristics (e.g., frequency value or f R ) of the peak of the spectrum of the electrical output signal depicted to calculate the distance. In this embodiment, the distance determination module 270 uses Equation (1b) and the characteristics (e.g., f R ) of the peak of the spectrum of the output signal from the detector array 230 to calculate the distance. The distance determination module 270 calculates multiple distances using Equation (1b), where there are multiple peaks (e.g., multiple f R values) in the spectrum of the output signal from the detector array 230.

[0058] In other embodiments, the module 270 is a Doppler compensation module that determines the magnitude of the Doppler shift and the correction range thereon. The Doppler compensation module is discussed in U.S. Provisional Application No. 62 / 428,109, filed November 30, 2016.

[0059] Figure 2BDepicts an alternative hardware arrangement that allows an input digital chirp signal based on a square wave digital chirp signal to be introduced from an RF waveform generator 215 into a reference path that generates an LO signal. In some embodiments, the input digital chirp signal introduced into the reference path is independent of the input digital chirp signal introduced into the transmit path. The laser source 212, beam splitter 216, transmit signal 205, scanning optics 218, optical mixer 232, detector array 230, acquisition system 240, and processing system 250 are as described above with reference to Figure 2A as described. In Figure 2B , there are two separate optical modulators, 214a in the transmit path and 214b in the reference path 282, to apply the input digital chirp from the generator 215 to an optical carrier.

[0060] The beam splitter 216 is moved between the laser source 212 and the modulators 214a and 214b to produce an optical signal 283 that impinges on the modulator 214a and a lower amplitude reference path signal 287a that impinges on the modulator 214b in the modified reference path 282. In this embodiment, the light 201 is split into a transmit (TX) path beam 283 and a reference / local oscillator (LO) path beam 287a before modulation occurs; and separate modulators are used in each path. Using the dual modulator method, the input digital chirp signal with an offset start frequency and / or offset start time and / or offset start phase can be used to program either path. By shifting the delays used in each range gate, the system can still make high-resolution measurements without a doubt despite other system limitations (detector and digitizer bandwidth, measurement time, etc.). As described above, the software-controlled delayed reference signal 287b is then mixed with the return signal 291. In other embodiments, the software-controlled delay of the LO reference path 282 allows the adaptive scan method to also be adaptive in the down-range dimension.

[0061] In some embodiments, the RF waveform generator 215 includes a pair of channels, where the first channel outputs a first input digital chirp signal to the modulator 214a, and the second channel outputs a second input digital chirp signal to the modulator 214b. In some embodiments, the first input digital chirp signal is based on a first square wave digital chirp signal, and the second input digital chirp signal is based on a second square wave digital chirp signal.

[0062] In one embodiment, modulator 214a is a frequency modulator that receives a first input digital chirp signal to modulate the frequency of transmitted beam 283 and generate a transmitted signal 205 pulse having a bandwidth B and a duration τ. In another embodiment, modulator 214b is a phase modulator that receives a second input digital chirp signal from RF waveform generator 215. In these embodiments, the first input digital chirp signal received at modulator 214a is independent of the second input digital chirp signal received at modulator 214b. In one embodiment, the phase of reference signal 207 is modulated by the phase modulator based on the second input digital chirp signal to advantageously reduce the intensity noise variation between reference beam 207b and return signal 291 at optical mixer 232.

[0063] In some embodiments, modulation of the current applied to the drive laser is used to actively linearize the lasers used. Electro-optic modulators that provide modulation are also used in experiments. As described in more detail below for the various embodiments, the system is configured to generate a chirp having a bandwidth B and a duration τ suitable for the desired lower range resolution. For example, in some of the illustrated embodiments, a B value of approximately 90 GHz and a τ of approximately 200 milliseconds (ms, 1 ms = 10 -3 seconds) are selected to operate within the relatively low detector array frame rate of the experiments conducted. These selections are made to observe a reasonable large range window of approximately 30 cm, which is typically important for determining the shape of an object and object recognition. This technique will be applicable to chirp bandwidths from 10 MHz to 5 THz. However, for 3D imaging applications, the typical range is from approximately 300 MHz to approximately 20 GHz for the chirp bandwidth, from approximately 250 nanoseconds (ns, ns = 10 -9 seconds) to approximately 1 millisecond (ms, 1 ms = 10 -3 seconds) for the chirp duration, from approximately 0 meters to approximately 20 km for the distance to the target, from approximately 3 millimeters (mm, 1 mm = 10 -3 meters) to approximately 1 meter (m) for the target point size, and from approximately 7.5 mm to approximately 0.5 m for the target depth resolution. It should be noted that in these cases, the range window can be extended to several kilometers and the Doppler resolution can also be high (depending on the chirp duration).

[0064] Although for purposes of illustration in Figure 2A and Figure 2BThe processes, devices, and data structures are depicted as monolithic blocks in a particular arrangement, but in other embodiments, one or more processes or data structures or portions thereof are arranged differently on the same or different hosts, arranged in one or more databases, or omitted, or one or more different processes or data structures are included on the same or different hosts. For example, optical splitter 216 and reference path 220 include zero or more optical couplers.

[0065] Figure 3A is a block diagram showing exemplary components of an Figure 2A RF waveform generator 215 of LIDAR system 200 according to an embodiment. In one embodiment, RF waveform generator 215 includes transceiver 302 to generate a square wave digital chirp signal 306. In some embodiments, transceiver 302 is a transceiver of a field programmable gate array (FPGA). In one embodiment, LIDAR system 200 uses the FPGA to perform other functions, and RF waveform generator 215 accesses one or more otherwise unused transceivers 302 of the FPGA. In one embodiment, the transceiver 302 of the FPGA has a bandwidth of approximately 32 GHz. For the purposes of this description, when discussing the parameters of a digital signal, the unit of Hertz (Hz) is interchangeably used with the unit of samples per second (SPS). This embodiment advantageously employs existing components of LIDAR system 200 to generate a broadband digitized chirp signal to modulate an optical carrier in LIDAR system 200. In an exemplary embodiment, processing system 250 also uses the FPGA to calculate the distance to a target and / or generate image data based on the calculated distance.

[0066] In one embodiment, the digital chirp waveform is generated in software running on a processor core on the FPGA. In some embodiments on a conventional FPGA, these processors are soft-core processors implemented in the FPGA architecture. In other embodiments on a system-on-chip (SoC) FPGA, in addition to the programmable logic architecture, there are hard-core processors permanently implemented in silicon.

[0067] In one embodiment, a software algorithm takes multiple parameters as input, such as chirp repetition rate, chirp start frequency, chirp bandwidth, chirp direction, and output sampling rate, and produces chirp waveform data as output. The software-generated waveform is loaded into a dual-port block random access memory (RAM) element, which is implemented in the programmable logic architecture of the FPGA. One port of these RAM blocks is connected to the processor core to enable this waveform loading; the other port is connected to a finite state machine (FSM) in the programmable logic. The FSM is designed to read out the waveform and feed it to the multi-gigabit transceiver (MGT) of the FPGA under the strict timing requirements needed for seamless, repetitive chirp generation. The FSM itself can also be controlled by software running on the processor core, enabling the generation and control of a complete waveform from a software environment.

[0068] The square wave waveform constitutes at least a part of instruction 217, which is uploaded to the FPGA transceiver 302 and used as an input to generate a square wave digital chirp signal 306. In some embodiments, the square wave instruction 217 is a parameterized bitstream based on one or more parameters of the square wave digital chirp signal 306. In one embodiment, the parameters include one or more of start frequency, stop frequency, pulse duration, and pulse repetition frequency (PRF). In an exemplary embodiment, the start frequency is about 500 MHz, the stop frequency is about 14 GHz, the pulse duration is about 100 microseconds (μsec), and the PRF is about 10 kilohertz (kHz). In some embodiments, the square wave instruction 217 includes a bitstream that indicates an output of a fixed amplitude over a first number of clock cycles (e.g., 111000) of the transceiver 302 to output a first frequency, followed by an output of a fixed amplitude over a second number of clock cycles (e.g., 1100) less than the first number of clock cycles to output a second frequency greater than the first frequency. In other embodiments, the square wave instruction is included in the range determination module 270 and transmitted from the processing system 250 to the FPGA transceiver 302, rather than being uploaded to the FPGA transceiver 302.

[0069] In some embodiments, the square wave command 217 is determined by digitizing a cosine chirp waveform (e.g., the cosine chirp 406 signal in FIG. 4A). In one embodiment, the bit stream of the square wave command 217 for each clock cycle is determined based on the independent variable of the cosine chirp waveform at the time increment corresponding to each clock cycle. In one embodiment, the bit stream is set to 1 for those time increments where the independent variable of the cosine chirp waveform is on the right half of the unit circle. In this embodiment, the bit stream is set to 0 for those time increments where the independent variable of the cosine chirp waveform is on the left half of the unit circle. This advantageously avoids resource-intensive steps, such as using a look-up table or other methods to calculate the cosine function, thus providing an efficient way to calculate the square wave command 217.

[0070] FIG. 4A is a graph 400 showing an example of an input digital chirp signal used in a conventional LIDAR system according to an embodiment. The horizontal axis 402 is time, measured in arbitrary units. The vertical axis 404 is voltage, measured in arbitrary units. In one embodiment, the input digital chirp signal is a cosine chirp 406 signal that linearly increases in frequency from a first frequency to a second frequency greater than the first frequency over time. FIG. 4B is a graph 450 showing an example of the spectrum of the input digital chirp signal of FIG. 4A according to an embodiment. The horizontal axis 452 is frequency in gigahertz (GHz), and the vertical axis 454 is power in decibels (dB). In one embodiment, the curve 456 represents the spectrum of the cosine chirp 406 signal. The bandwidth 460 of the spectrum extends from the first frequency to the second frequency. In one embodiment, the bandwidth 460 is in the range from about 500 MHz to about 10 GHz. Additionally, the dynamic range 462 is defined as the difference between the amplitude of the curve 456 within the bandwidth 460 and the amplitude of the curve 456 outside the bandwidth 460. The dynamic range 462 is at least 50 dB.

[0071] Figure 5A is a graph showing an example of a square wave digital chirp signal 306 used in Figure 2A and Figure 2B the LIDAR systems 200, 280 according to an embodiment. The horizontal axis 502 is time, measured in arbitrary units. The vertical axis 504 is voltage, measured in arbitrary units. In one embodiment, the square wave digital chirp signal 306 linearly increases in frequency from a first frequency to a second frequency greater than the first frequency over time. Figure 5B is a graph showing an example of Figure 5A the spectrum of the square wave digital chirp signal 306 according to an embodiment. The horizontal axis 552 is frequency in gigahertz (GHz), and the vertical axis 554 is power in decibels (dB).

[0072] In some embodiments, Figure 3A The RF waveform generator includes an RF conditioning 304 component that performs one or more conditioning steps to convert a square wave digital chirp signal 306 into a Figure 3A The input digital linear frequency modulation signal 308 to the modulator 214 is input to the modulator 214. In some embodiments, the conditioning step modifies Figure 5B 4B , the spectrum of the square wave digital linear frequency modulation signal 306 is such that the resulting spectrum of the input digital linear frequency modulation signal 308 shares one or more characteristics with the spectrum of the input digital linear frequency modulation signal in FIG. 4B . In one embodiment, the characteristics of the spectrum of the input digital linear frequency modulation signal in FIG. 4B include bandwidth 460 and / or dynamic range 462.

[0073] Figure 3B It is to show the embodiment according to Figure 3A FIG. 2 is a block diagram of exemplary components of RF conditioning 304 of RF waveform generator 215. In some embodiments, RF conditioning 304 components include filter 310, which is used to Figure 5B 4B . In some embodiments, the filter 310 removes a range of frequencies from the spectrum of the square wave digital linear frequency modulation signal 306. In one embodiment, the filter 310 is a bandpass filter that passes a frequency range overlapping with the bandwidth 460 and removes a frequency range that falls outside the bandwidth 460 of the cosine linear frequency modulation signal 406 in FIG. 4B . In some embodiments, the high-order harmonics 555, 557 in the spectrum of the square wave digital linear frequency modulation signal 306 are removed by the filter 310. In this embodiment, the filter 310 removes the high-order harmonics 555, 557 from the spectrum of the square wave digital linear frequency modulation signal 306. Figure 5B The frequency range (e.g., above 5 GHz) is removed from the spectrum of the square wave digital linear frequency modulation signal 306. In an exemplary embodiment, a high order low pass filter is used to significantly attenuate the 3rd and higher order harmonics. In some embodiments, the filtering advantageously converts the digital waveform into a sinusoidal waveform for use in the RF circuit. In an exemplary embodiment, the frequency range is removed so that the excluded frequency range encompasses the high order harmonics 555, 557. As will be appreciated by one of ordinary skill in the art, the high order harmonics 555, 557 are Figure 5A The square wave linear frequency modulation 306 results in a frequency component.

[0074] After using filter 310 to remove a frequency range from the spectrum, the resulting bandwidth 560 is the spectrum of the input digital chirp signal 308. In some embodiments, bandwidth 560 overlaps with bandwidth 460 of the input digital chirp signal in FIG. 4B. In some embodiments, bandwidth 560 of input digital chirp signal 308 exceeds bandwidth 460 of the input digital chirp signal. In various embodiments, bandwidth 560 is in the range from about 2 Ghz to about 30 GHz. Preferably, bandwidth 560 is in the range from about 10 GHz to about 30 GHz. In one exemplary embodiment, bandwidth 560 is in the range from about 2 Ghz to about 3 GHz. Generally, the bandwidth of input digital chirp signal 308 extends from a first frequency to a second frequency. In an exemplary embodiment, the first frequency is in the range from about 500 Mhz to about 2 Ghz, and the second frequency is in the range from about 3 Ghz to about 32 GHz. Although Figure 5B bandwidth 560 in the range from 2 GHz to 3 GHz is depicted, this is merely an exemplary embodiment.

[0075] In other embodiments, filter 310 removes a frequency range from Figure 5B the frequency power spectrum of the square wave digital chirp signal 306 in order to establish a minimum dynamic range for input digital chirp signal 308. The dynamic range is defined as the difference between the magnitude within bandwidth 560 of the frequency power spectrum and the magnitude outside bandwidth 560 of the frequency power spectrum. As Figure 5B depicted, before using filter 310 to remove the frequency range (e.g., harmonics 555, 557), the dynamic range 563 is about 10 dB, e.g., the difference between about 0 dB within bandwidth 560 and about -10 dB at the higher order harmonic 555. After using filter 310 to remove the frequency range (e.g., harmonics 555, 557), the dynamic range 562 increases to about 20 dB, e.g., the difference between about 0 dB within bandwidth 560 and about -20 dB at the noise floor. In one embodiment, the minimum dynamic range of input digital chirp signal 308 is in the range from about 20 dB to about 50 dB.

[0076] Although Figure 5B the magnitude of the spectrum depicted in is relatively flat across bandwidth 560, in some embodiments, the magnitude is non-uniform across bandwidth 560. In one embodiment, the magnitude gradually decreases at higher frequencies such that the magnitude is larger at lower frequencies and smaller at higher frequencies within bandwidth 560. If the spectrum of input digital chirp signal 308 has equal magnitude across bandwidth 560, it is advantageous for mixing.

[0077] In some embodiments, Figure 3BThe RF adjustment 304 component includes an equalizer 312 that is used to flatten the amplitude of the spectrum of the input digital chirp signal 308 over the bandwidth 560. Figure 6A is a graph showing an example of a transfer function 606 depicting the amplitude of the spectrum over the bandwidth 560. The horizontal axis 602 is frequency in arbitrary units, and the vertical axis 604 is power in arbitrary units. After the filter 310 has removed a frequency range from the square wave digital chirp signal 306, the filtered signal 311 is received at the equalizer 312, and the amplitude of its spectrum is similar to the transfer function 606 over the bandwidth 560. In some embodiments, the transfer function 606 is a characteristic of the transceiver 302 and depicts the amplitude of the spectrum of the square wave digital chirp signal over the bandwidth 560. Figure 6B is a graph showing, according to another embodiment, for Figure 3B is a graph showing an example of a transfer function 608 of an equalizer 312 of an RF adjustment component inserted to compensate for any adverse shape and / or filtered transfer function 606 of a transceiver. The horizontal axis 602 is frequency in arbitrary units, and the vertical axis 604 is power in arbitrary units. In some embodiments, the equalizer 312 receives the filtered signal 311 from the filter 310 according to the transfer function 608 and amplifies the amplitude of the spectrum of the filtered signal over the bandwidth 560. Figure 6C is a graph showing, according to an embodiment, from Figure 3A is a graph showing an example of a composite function 610 of an input digital chirp signal 308 output from the RF waveform generator 215. After the equalizer 312 amplifies the spectrum of the filtered signal 311 over the bandwidth 560 using the transfer function 608, the composite function 610 depicts the amplitude of the spectrum of the input digital chirp signal 308 over the bandwidth 560. In some embodiments, the amplitude of the spectrum is approximately equal over the bandwidth 560. In one exemplary embodiment, the amplitude of the spectrum is within about 10% flatness over the bandwidth 560. In another exemplary embodiment, the amplitude of the spectrum is characterized by a 3 dB to 6 dB variation over the bandwidth 560. In some embodiments, an RF amplifier 314 is provided to amplify the input digital chirp signal 308 before it is input to the modulator 214.

[0078] Figure 3C is a graph showing, according to an embodiment, Figure 2B is a block diagram of an exemplary component of an RF waveform generator 215 of a LIDAR system 280. Except that the FPGA transceiver 302 includes multiple channels (e.g., two channels) to generate multiple square wave digital chirp signals 306a, 306b, Figure 3C the RF waveform generator 215 of Figure 3A is similar to the waveform generator 215 of Figure 3CThe RF waveform generator 215 includes RF conditioning components 304 that perform one or more conditioning steps on each square-wave chirp signal 306a, 306b to generate input chirp signals 308a, 308b. In one embodiment, the first input chirp signal 308a is input to the modulator 214a to modulate the frequency of the transmit path beam 283, and the second input chirp signal 308b is input to the modulator 214b to modulate the phase of the reference path signal 287a. Modulation of the phase of the reference path signal 287a advantageously reduces the noise at the detector array 230 that is caused by intensity variations between the reference path signal 287b and the return light 291. In some embodiments, the square-wave chirp signals 306a, 306b are independent such that they are characterized by one or more different parameters (e.g., start frequency, stop frequency, pulse duration, pulse repetition frequency). In one embodiment, the square-wave instructions 207 uploaded to Figure 3C the FPGA transceiver 302 include different instructions for each square-wave chirp signal 306a, 306b.

[0079] FIG. 7 is a graph showing an exemplary measurement of the beat frequency generated by de-chirping in a conventional LIDAR system using the input chirp signal 406 of FIG. 4A. The horizontal axis 702 is frequency in MHz. The vertical axis 704 represents the return signal power density I T relative to the transmit power density I R (in decibels) (dB, power in dB = 20 log(I R / Ι T ))). The curve 706 depicts the magnitude of the power spectrum over the beat frequency. Multiple range return peaks 708a, 708b, 708c are depicted, where each range return peak indicates a corresponding range measurement for, e.g., different parts of an object or different objects. In one embodiment, the horizontal position of each peak 708a, 708b, 708c gives a different value of f R which is used with Equation (1b) to calculate the corresponding range. The dynamic range 712 is measured between the magnitude of the curve 706 at the return peak 708 (e.g., -40 dB) and the magnitude of the curve 706 at the noise floor (e.g., -80 dB). In one embodiment, the dynamic range 712 is approximately 40 dB. In some embodiments, a minimum dynamic range between approximately 20 dB and approximately 40 dB ensures accurate range detection.

[0080] Figure 8 is a graph showing, according to an embodiment, in the use of Figure 5AGraph of an exemplary measurement of the beat frequency generated by de-chirping in a LIDAR system 200 with a square-wave digital chirp signal 306. The horizontal axis 802 is frequency in MHz. The vertical axis 804 represents the return signal power density I T relative to the transmitted power density I R (in decibels) (dB, power in dB = 20 log(I R / Ι T ))). Curve 806 depicts the magnitude of the power spectrum at the beat frequency. Multiple range return peaks 808a, 808b, 808c are depicted, where each range return peak indicates a corresponding range measurement for, e.g., different parts of an object or different objects. The horizontal position of each peak 808a, 808b, 808c gives a different value of f R which is used by the range determination module 270 together with formula (1b) to calculate the corresponding range. In addition, multiple spurious peaks 810a, 810b are depicted, which would be obtained if the square-wave digital chirp signal 306 were input to the modulator 214 without RF conditioning 304. The spurious peaks 810a, 810b do not have corresponding peaks in FIG. 7 and would thus result in incorrect range calculations unless RF conditioning 304 is performed on the square-wave digital chirp signal 306. RF conditioning 304 advantageously removes the spurious peaks 810a, 810b from the beat frequency curve 806, thus avoiding incorrect range calculations.

[0081] The dynamic range 812 is measured between the magnitude of curve 806 at the return peak 808 (e.g., -35 dB) and the magnitude of curve 806 at the noise floor (e.g., -80 dB). In one embodiment, the dynamic range 812 is approximately 35 dB, which is greater than the minimum dynamic range (e.g., 20 dB) for accurate range detection. By removing the spurious peaks 810a, 810b, RF conditioning also advantageously increases the dynamic range 812 above the minimum dynamic range for accurate range detection. In some embodiments, although RF conditioning 304 decreases the magnitude of the return peak 808 relative to the magnitude of the return peak 708, for example, the dynamic range 812 exceeds the minimum dynamic range for accurate range detection.

[0082] Figure 9 is a flow chart showing an exemplary method 900 for using a square-wave digital chirp signal 306 for optical chirp ranging according to an embodiment. Although for purposes of illustration Figure 9The flowchart depicts the overall steps in a specific order, but in other embodiments, one or more steps or portions thereof are performed in a different order, or overlap in time, are performed serially or in parallel, or are omitted, or one or more additional steps are added, or are combined in some way to alter the method.

[0083] In step 901, an optical signal is emitted from a laser source. In one embodiment, in step 901, a carrier 201 is emitted from the laser source 212 in the system 200 of Figure 2A In other embodiments, in step 901, a carrier 201 is emitted from the laser source 212, and in Figure 2B In the system 280 of

[0084] In step 903, a square-wave digital chirp signal is generated using a transceiver. In one embodiment, in step 903, a square-wave digital chirp signal 306 is generated within the transceiver 302 from the FPGA. In another embodiment, the square-wave digital chirp signal 306 is generated by any high-speed serial transceiver on the chip. In an exemplary embodiment, the square-wave digital chirp signal 306 is generated by a digital signal processor (DSP). In one exemplary embodiment, the DSP is implemented on an application-specific integrated circuit (ASIC). In some embodiments, in step 903, an upload square-wave instruction 207 in the FPGA transceiver 302 is executed to generate the square-wave digital chirp signal 306. In other embodiments, in step 903, the square-wave instruction is included in the distance determination module 270 and transmitted from the processing system 250 to the FPGA transceiver 302 to generate the square-wave digital chirp signal 306. In other embodiments, other digital signal sources are used.

[0085] In one embodiment, in step 903, a single square-wave digital chirp signal 306 is generated within the transceiver 302 and output from a single channel of the transceiver 302. In other embodiments, in step 903, multiple square-wave digital chirp signals 306a, 306b are generated within the transceiver and output from multiple channels of the transceiver 302.

[0086] In step 905, the spectrum of the square-wave digital chirp signal 306 generated in step 903 is determined. In one embodiment, in Figure 5B the spectrum of the square-wave digital chirp signal 306 is depicted and includes higher-order harmonics 555, 557. In some embodiments, step 905 is omitted in the case of predetermining the spectrum of the square-wave digital chirp signal 306.

[0087] In step 907, a frequency range is removed from the spectrum of the square wave digital chirp signal 306 determined in step 905. In some embodiments, in step 907, the frequency range is removed using filter 310. In some embodiments, in step 907, filter 310 is a bandpass filter that passes a range of frequencies within bandwidth 560 and removes one or more frequency ranges outside of bandwidth 560 of the spectrum. In one embodiment, bandwidth 560 is based on the bandwidth 460 of the spectrum of the cosine chirp signal 406. In some embodiments, in step 907, filter 310 removes the frequency range such that bandwidth 560 overlaps bandwidth 460 and bandwidth 560 is in the range from about 10 GHz to about 30 GHz. In other embodiments, in step 907, filter 310 removes the frequency range such that the dynamic range 563 (e.g., 10 dB) is increased to the dynamic range 562 (e.g., 20 dB). In some embodiments, the increased dynamic range 562 exceeds a minimum dynamic range (e.g., 20 dB) for accurate distance detection. In some embodiments, in step 907, filter 310 removes the frequency range that encompasses the higher order harmonics 555, 557. In an exemplary embodiment, filter 310 removes the frequency range above a threshold frequency (e.g., 5 GHz) to remove the higher order harmonics 555, 557. In some embodiments, in the case where the spectrum of the square wave digital chirp signal 306 is pre-determined, step 905 is omitted, and step 907 removes the frequency range based on a predetermined spectrum. In some embodiments, in step 907, filter 310 converts the square wave digital chirp signal 306 into a filtered signal 311 incident on the equalizer 312.

[0088] In step 909, the amplitudes of the spectrum from step 907 are equalized to generate the input digital chirp signal 308. In one embodiment, the amplitudes of the spectrum of the filtered signal 311 over bandwidth 560 are similar to the transfer function 606, e.g., decreasing gradually at higher frequencies within bandwidth 560. In one embodiment, in step 909, the equalizer 612 amplifies the amplitudes of the spectrum of the filtered signal 311 over bandwidth 560 using the transfer function 608. The resulting amplitudes of the spectrum of the input digital chirp signal 308 over bandwidth 560 are similar to the combined transfer function 610, e.g., approximately equal over bandwidth 560. In some embodiments, the input digital chirp signal 308 is amplified by the RF amplifier 314 and then input to the modulator 214.

[0089] In other embodiments, any digital-to-analog converter may be used in or instead of steps 903, 905, 907, 909 to generate an input digital chirp signal 308 based on the uploaded square wave command 207. As will be appreciated by those skilled in the art, the selection of the digital-to-analog converter involves a trade-off between bit depth and bandwidth.

[0090] In step 911, a modulator is utilized to modulate the frequency of the optical signal in step 901 based on the input digital chirp signal 308 from step 909. In one embodiment, in step 911, the frequency of the carrier 201 is modulated using the modulator 214 of FIG. 2 based on the input digital chirp signal 308 from the RF waveform generator 215 to generate a pulse having a bandwidth B and a duration τ. In other embodiments, in step 911, where the carrier 201 is emitted from the laser source 212, and Figure 2B a beam splitter 216 then divides the carrier 201 into an emission path beam 283 and a reference path signal 287a, and the frequency of the emission path beam 283 is modulated using the modulator 214a based on a first input digital chirp signal 308a from the RF waveform generator 215.

[0091] In step 913, the optical signal of step 911 is divided into an emission optical signal and a reference optical signal. In one embodiment, in step 913, the modulated beam 203 from the Figure 2A modulator 214 therein is divided into an emission signal 205 and a reference signal 207a using the beam splitter 216. In one embodiment, relative to the Figure 2B embodiment, step 913 is omitted because the beam splitter 216 has already separated the optical signal before modulation in step 911.

[0092] In step 914, a modulator is utilized to modulate the phase of the reference path signal 287a based on the input digital chirp signal from step 909. In one embodiment, in step 914, the phase of the reference path signal 287a is modulated using the Figure 2B modulator 214b based on the input digital chirp signal 308b from the RF waveform generator 215. In some embodiments, in step 911, the input digital chirp signal 308b is different from the input digital chirp signal 308a incident on the modulator 214a to modulate the frequency of the emission path beam 283. In some embodiments, step 914 is omitted.

[0093] In step 915, the reference path optical signal and the return signal 291 are combined at a detector. In one embodiment, in step 915, at the Figure 2AThe reference path signal 207b and the return signal 291 are combined at the detector array 230 in. In another embodiment, in step 915, at Figure 2B the reference path signal 287b and the return signal 291 are combined at the detector array 230 in.

[0094] In step 917, an electrical output signal is generated from the detector. In one embodiment, based on step 915, an electrical output signal is generated from the detector array 230. In one embodiment, the electrical output signal includes one or more beat frequencies based on the frequency difference between the reference signal and the return signal.

[0095] In step 919, the distance to the object is determined based on the characteristics of the Fourier transform of the electrical output signal in step 917. In one embodiment, based on the distance determination module 270, the processing system 250 performs a Fourier transform on the electrical output signal received from the detector array 230 to obtain the spectrum of the electrical output signal (e.g., Figure 8 curve 806 in). Additionally, the processing system 250 identifies one or more distance return peaks 808a, 808b, 808c in the spectrum of the electrical output signal. The processing system 250 further determines one or more characteristics of the peaks and zero or more Doppler shifts (e.g., frequency values or f R ). The processing system 250 then calculates the range of each peak based on the detected frequency and / or the Doppler-corrected frequency and formula (1b).

[0096] 3. Computational Hardware Overview

[0097] Figure 10 is a block diagram showing a computer system 1000 on which embodiments of the present invention can be implemented. The computer system 1000 includes a communication mechanism, such as a bus 1010, for transferring information between other internal and external components of the computer system 1000. The information is represented as physical signals of measurable phenomena, typically voltages, but in other embodiments includes phenomena such as magnetic, electromagnetic, pressure, chemical, molecular atomic, and quantum interactions. For example, north-south magnetic fields or zero and non-zero voltages represent the two states (0, 1) of binary digits (bits). Other phenomena can represent higher-radix numbers. The superposition of multiple simultaneous quantum states before measurement represents a qubit. A sequence of one or more digits constitutes digital data, which is used to represent digital or character codes. In some embodiments, information referred to as analog data is represented by nearly continuous measurable values within a specific range. The computer system 1000 or a part thereof constitutes means for performing one or more steps of one or more methods described herein.

[0098] A binary digit sequence constitutes digital data, which is used to represent numerical or character codes. Bus 1010 includes a number of parallel information conductors, enabling rapid transfer of information between devices coupled to bus 1010. One or more processors 1002 for processing information are coupled to bus 1010. Processor 1002 performs a set of operations on the information. The set of operations includes introducing information from bus 1010 and placing information on bus 1010. The set of operations typically also includes comparing two or more information units, shifting the positions of information units, and combining two or more information units, such as by addition or multiplication. The sequence of operations performed by processor 1002 constitutes computer instructions.

[0099] Computer system 1000 also includes a memory 1004 coupled to bus 1010. Memory 1004, such as random access memory (RAM) or other dynamic storage devices, stores information including computer instructions. Dynamic memory allows computer system 1000 to change the information stored therein. RAM allows information units stored at locations called memory addresses to be stored and retrieved independently of information at adjacent addresses. Memory 1004 is also used by processor 1002 to store temporary values during the execution of computer instructions. Computer system 1000 also includes a read-only memory (ROM) 1006 or other static storage devices coupled to bus 1010 for storing static information including instructions not changed by computer system 1000. Also coupled to bus 1010 is a non-volatile (permanent) storage device 1008 (such as a magnetic disk or optical disk) for storing information including instructions that remains present even if computer system 1000 is turned off or otherwise powered down.

[0100] Information including instructions is provided to bus 1010 from an external input device 1012 for use by the processor, such as a keyboard containing alphanumeric keys operated by a human user or a sensor. The sensor detects conditions in its vicinity and converts these detections into signals compatible with signals used to represent information in computer system 1000. Other external devices coupled to bus 1010 that are primarily used for human interaction include a display device 1014 for presenting images, such as a cathode ray tube (CRT) or liquid crystal display (LCD), and a pointing device 1016, such as a mouse or trackball or cursor direction keys, for controlling the position of a small cursor image presented on display 1014 and issuing commands associated with graphical elements presented on display 1014.

[0101] In the illustrated embodiment, special-purpose hardware, such as application specific integrated circuit (IC) 1020, is coupled to bus 1010. The special-purpose hardware is configured to perform operations not performed by processor 1002 fast enough for a particular purpose. Examples of special-purpose ICs include graphics accelerator cards for generating images for display 1014, cryptographic boards for encrypting and decrypting messages sent over a network, speech recognition, and interfaces to special external devices such as robotic arms and medical scanning devices that repeatedly execute certain complex sequences of operations that are more efficiently implemented in hardware.

[0102] Computer system 1000 also includes one or more instances of communication interface 1070 coupled to bus 1010. Communication interface 1070 enables two-way communication with various external devices that operate with their own processors, such as printers, scanners, and external disks. Typically, the coupling is made using network link 1078 connected to local network 1080, to which various external devices with their own processors are connected. For example, communication interface 1070 can be a parallel port or a serial port or a universal serial bus (USB) port on a personal computer. In some embodiments, communication interface 1070 is an integrated services digital network (ISDN) card or a digital subscriber line (DSL) card or a telephone modem that provides an information communication connection to a corresponding type of telephone line. In some embodiments, communication interface 1070 is a cable modem that converts signals on bus 1010 into signals for communication connection over coaxial cable or into optical signals for communication connection over fiber optic cable. As another example, communication interface 1070 can be a local area network (LAN) card to provide a data communication connection to a compatible LAN such as Ethernet. A wireless link can also be implemented. Carrier waves, such as acoustic and electromagnetic waves (including radio waves, light waves, and infrared waves), travel through space without wires or cables. Signals include artificial changes in the amplitude, frequency, phase, polarization, or other physical properties of a carrier wave. For a wireless link, communication interface 1070 transmits and receives electrical, acoustic, or electromagnetic signals, including infrared and optical signals, that carry information streams, such as digital data.

[0103] As used herein, the term computer-readable medium refers to any medium that participates in providing information to processor 1002, which includes instructions for execution. Such a medium may take many forms, including but not limited to non-volatile media, volatile media, and transmission media. Non-volatile media includes, for example, optical or magnetic disks, such as storage device 1008. Volatile media includes, for example, dynamic memory 1004. Transmission media includes, for example, coaxial cables, copper wire, fiber optic cables, and waves that travel through space without wires or cables, such as acoustic waves and electromagnetic waves, including radio waves, light waves, and infrared waves. As used herein, the term computer-readable storage medium refers to any medium that participates in providing information to processor 1002 other than a transmission medium.

[0104] Common forms of computer-readable media include, for example, floppy disks, flexible disks, hard disks, magnetic tape, or any other magnetic medium, CD-ROM (compact disc read-only memory), DVD (digital versatile disc), or any other optical medium, punch cards, paper tape, or any other physical medium with patterns of holes, RAM (random access memory), PROM (programmable read-only memory), EPROM (erasable programmable read-only memory), flash EEPROM (electrically erasable programmable read-only memory), or any other memory chip or cartridge, carrier waves, or any other medium from which a computer can read. The term "non-transitory computer-readable storage medium" is used herein to refer to any medium that participates in providing information to processor 1002 other than carrier waves and other signals.

[0105] Logic encoded in one or more tangible media includes one or both of processor instructions on a computer-readable storage medium and special-purpose hardware, such as ASIC 1020.

[0106] Network link 1078 generally provides information communication through one or more networks to other devices that use or process the information. For example, network link 1078 may provide a connection through local network 1080 to host computer 1082 or to a device 1084 operated by an Internet service provider (ISP). The ISP device 1084 in turn provides data communication services through the public global packet-switching communication network of the network now commonly referred to as the Internet 1090. A computer referred to as server 1092 connected to the Internet provides services in response to information received through the Internet. For example, server 1092 provides information representing video data for presentation at display 1014.

[0107] The present invention relates to the use of a computer system 1000 for implementing the techniques described herein. According to one embodiment of the invention, those techniques are performed by the computer system 1000 in response to one or more sequences of one or more instructions contained in a memory 1004 being executed by a processor 1002. Such instructions (also referred to as software and program code) may be read into the memory 1004 from another computer-readable medium, such as a storage device 1008. Execution of the instruction sequences contained in the memory 1004 causes the processor 1002 to perform the method steps described herein. In alternative embodiments, hardware, such as an application specific integrated circuit 1020, may be used in place of or in combination with software to implement the present invention. Accordingly, embodiments of the present invention are not limited to any specific combination of hardware and software.

[0108] Signals transmitted over the network link 1078 and other networks via the communication interface 1070 carry information to and from the computer system 1000. The computer system 1000 may send and receive information, including program code, via the networks 1080, 1090, in particular via the network link 1078 and the communication interface 1070. In an example using the Internet 1090, a server 1092 sends program code for a particular application requested by a message sent from the computer 1000 via the Internet 1090, an ISP device 1084, a local network 1080, and the communication interface 1070. The received code may be executed by the processor 1002 upon receipt, or may be stored in the storage device 1008 or other non-volatile storage for later execution, or both. In this manner, the computer system 1000 may obtain application program code in the form of a signal on a carrier wave.

[0109] Various forms of computer-readable media may be involved in carrying one or more instruction or data sequences, or both, to the processor 1002 for execution. For example, the instructions and data may initially be carried on a magnetic disk of a remote computer, such as a host 1082. The remote computer loads the instructions and data into its dynamic memory and sends the instructions and data over a telephone line using a modem. A modem local to the computer system 1000 receives the instructions and data over the telephone line and converts the instructions and data into a signal on an infrared light carrier wave used as the network link 1078 using an infrared light emitter. An infrared detector acting as the communication interface 1070 receives the instructions and data carried in the infrared signal and places information representing the instructions and data onto a bus 1010. The bus 1010 carries the information to the memory 1004, and the processor 1002 retrieves and executes the instructions from the memory using some of the data sent with the instructions. The instructions and data received in the memory 1004 may optionally be stored on the storage device 1008 before or after being executed by the processor 1002.

[0110] Figure 11Shows a chipset 1100 on which embodiments of the present invention can be implemented. The chipset 1100 is programmed to perform one or more steps of the methods described herein and includes, for example, the processor and memory components incorporated in one or more physical packages (e.g., chips) as described with respect to FIG. 1. By way of example, a physical package includes an arrangement of one or more materials, components, and / or wires on a structural component (e.g., a substrate) to provide one or more characteristics such as physical strength, size savings, and / or limitation of electrical interactions. It is contemplated that in some embodiments, the chipset may be implemented in a single chip. The chipset 1100 or a portion thereof constitutes means for performing one or more steps of the methods described herein.

[0111] In one embodiment, the chipset 1100 includes a communication mechanism such as a bus 1101 for transferring information between the components of the chipset 1100. The processor 1103 has connectivity with the bus 1101 to execute instructions and process information stored, for example, in the memory 1105. The processor 1103 may include one or more processing cores, each configured to execute independently. A multi-core processor enables multi-processing within a single physical package. Examples of multi-core processors include two, four, eight, or more processing cores. Alternatively or additionally, the processor 1103 may include one or more microprocessors configured in a cascade via the bus 1101 to enable independent execution of instructions, pipelining, and multi-threading. The processor 1103 may also be accompanied by one or more dedicated components to perform certain processing functions and tasks, such as one or more digital signal processors (DSPs) 1107 or one or more application specific integrated circuits (ASICs) 1109. The DSP 1107 is typically configured to process real-world signals (e.g., sound) in real time independent of the processor 1103. Similarly, the ASIC 1109 may be configured to perform specialized functions not easily performed by a general-purpose processor. Other dedicated components that assist in performing the inventive functions described herein include one or more field programmable gate arrays (FPGAs) (not shown), one or more controllers (not shown), or one or more other specialized computer chips.

[0112] The processor 1103 and the attached components have connectivity with the memory 1105 via the bus 1101. The memory 1105 includes dynamic memory (e.g., RAM, disk, writable optical disc, etc.) and static memory (e.g., ROM, CD-ROM, etc.) for storing executable instructions that, when executed, perform one or more steps of the methods described herein. The memory 1105 also stores data associated with or generated by the execution of one or more steps of the methods described herein.

[0113] 4. Changes, Extensions, and Modifications

[0114] In the foregoing specification, the present invention has been described with reference to specific embodiments of the invention. However, it will be apparent that various modifications and changes can be made thereto without departing from the broader spirit and scope of the invention. Accordingly, the specification and drawings are to be regarded as illustrative rather than restrictive. Throughout the specification and claims, unless the context requires otherwise, the word "comprise" and its variations such as "comprises" and "comprising" will be understood to imply the inclusion of the stated item, element or step or group of items, elements or steps but not the exclusion of any other item, element or step or group of items, elements or steps. Further, the indefinite articles "a" or "an" are intended to denote one or more of the item, element or step so modified. As used herein, unless otherwise clear from the context, a value is "about" another value if it is within a factor of two (twice or half) of the other value. Although exemplary ranges are given, it is also intended in various embodiments to include any included range unless otherwise clear from the context. Thus, in some embodiments, a range from 0 to 10 includes a range from 1 to 4.

[0115] 5. References

[0116] Adany, P., C. Allen, and R. Hui, "Chirped Lidar Using Simplified Homodyne Detection", Journal of Lightwave Technology, August 15, 2009, Vol. 27, No. 16.

[0117] Hui, R., C. Allen, and P. Adany, "Coherent detection scheme for FM Chirped laser RADAR", U.S. Patent 7,742,152, June 22, 2010.

[0118] Kachelmyer, A. L., "Range-Doppler Imaging with a Laser Radar", Lincoln Laboratory Journal, 1990, Vol. 3, No. 1.

Claims

1. A light detection and ranging system, comprising: A laser source configured to generate a carrier wave; An equalizer configured to modulate the amplitude of a chirp signal, wherein the chirp signal is a square wave digital chirp signal; A modulator configured to modulate the frequency of the carrier wave using the chirp signal; One or more scanning optical devices configured to generate a transmitted signal using the modulated carrier wave and transmit the transmitted signal; And A processor configured to determine the Doppler shift of an object based on a return signal and a reference signal; Wherein the return signal is received from the object in response to the transmitted signal.

2. The light detection and ranging system according to claim 1, further comprising a reference path configured to delay the reference signal.

3. The optical detection and ranging system according to claim 1, wherein, The reference signal is generated from the carrier wave by splitting the modulated carrier wave.

4. The optical detection and ranging system according to claim 1, wherein, The laser source is a first laser source, and the reference signal is generated by a second laser source separate from the first laser source.

5. The optical detection and ranging system according to claim 1, wherein, The frequency of the chirp signal changes between a first frequency and a second frequency.

6. The light detection and ranging system according to claim 1, further comprising a filter configured to increase the dynamic range of the chirp signal to from 20 dB to 50 dB.

7. The optical detection and ranging system according to claim 1, wherein, The bandwidth of the chirp signal is from 10 GHz to 30 GHz.

8. The optical detection and ranging system according to claim 1, wherein, The processor is configured to determine the distance to the object using the Doppler shift.

9. The light detection and ranging system according to claim 1, further comprising a beam splitter configured to divide the carrier wave into the transmitted signal and the reference signal.

10. The optical detection and ranging system according to claim 1, wherein, The processor is configured to determine the Doppler shift by performing a Fourier transform using the reference signal and the return signal.

11. The light detection and ranging system according to claim 1, further comprising a radio frequency (RF) waveform generator, the RF waveform generator including the equalizer, the RF waveform generator being configured to provide the chirp signal to the modulator as a digital signal.

12. A system for determining the Doppler shift of an object, comprising: Means configured to: Generate a carrier wave; Flatten the amplitude of the spectrum of a chirp signal over a specific bandwidth, wherein the chirp signal is a square wave digital chirp signal; Modulate the frequency of the carrier wave using the chirp signal; Generate a transmitted signal using the modulated carrier wave; And Transmit the transmitted signal; And A processor configured to determine the Doppler shift of the object based on a return signal and a reference signal, wherein the return signal is received from the object in response to the transmitted signal, and wherein the reference signal is generated from the carrier wave.

13. The system according to claim 12, wherein, The means is configured to delay the reference signal such that the reference signal is received together with the return signal.

14. The system according to claim 12, wherein, The frequency of the chirp signal changes between a first frequency and a second frequency.

15. The system according to claim 12, wherein The device is configured to filter the chirp signal to increase the dynamic range of the chirp signal from 20 dB to 50 dB.

16. The system according to claim 12, wherein, The bandwidth of the chirp signal ranges from 10 GHz to 30 GHz.

17. The system according to claim 12, wherein The processor is configured to determine the distance to the object using the Doppler shift.

18. A system for determining the Doppler shift of an object, comprising: An equalizer configured to modulate the amplitude of a chirp signal, wherein the chirp signal is a square-wave digital chirp signal; A modulator configured to modulate a carrier using the chirp signal to generate a transmitted signal; One or more scanning optics configured to transmit the transmitted signal; And One or more processors configured to determine the Doppler shift of the object based on a return signal and a reference signal, wherein the return signal is based on the scattering of the transmitted signal by the object, and wherein the reference signal is generated from the carrier.

19. The system according to claim 18, wherein, The one or more processors are configured to determine the distance to the object using the Doppler shift.

20. The system according to claim 18, wherein, The processor is configured to determine the Doppler shift by performing a Fourier transform using the reference signal and the return signal.

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