Power amplifier circuits and electronic equipment

Abstract

The present application discloses a power amplifier circuit and electronic device, which relate to the field of radio frequency and are used to reduce the insertion loss of the carrier branch when the carrier branch and the peak branch are coupled through a coupler. The power amplifier circuit includes: a first power amplifier, a second power amplifier, a matching network, a second phase-shifting network, a coupler, and a matching load. The input end of the first power amplifier is used to input a radio frequency signal. The output end of the first power amplifier is electrically connected to the input end of the coupler. The through end of the coupler and the first end of the matching network are electrically connected to a junction point. The second end of the matching network is used to output the radio frequency signal after power amplification. The isolation end of the coupler is grounded through the matching load. The coupling end of the coupler is electrically connected to the junction point through the second phase-shifting network and the second power amplifier. When the first power amplifier performs power back-off, the second power amplifier is open-circuited.

Description

Technical Field

[0001] The present application relates to the field of radio frequency, and in particular to a power amplifier circuit and electronic equipment. Background Art

[0002] Power amplifiers are used in the power amplifier circuits of electronic devices to amplify radio frequency signals. In the linear region, the output power of the power amplifier increases linearly with increasing input power. As the input power increases further, the amplifier enters the nonlinear region, where the output power no longer increases with input power and reaches saturation, achieving maximum efficiency. Typically, power amplifiers are designed so that the saturated output power corresponds to the peak power of the input RF signal. This means that when the input RF signal reaches peak power, the output power also reaches saturation, allowing the amplifier to operate in the linear region as much as possible. However, this means that the power amplifier cannot reach saturation most of the time, and therefore its efficiency is not optimal most of the time.

[0003] In the prior art, power back-off technology is used to improve power amplification efficiency by using two power amplifiers. Specifically, frequency components are used to load modulate the first power amplifier, allowing it to enter the saturation region earlier as input power increases. This improves the efficiency of the power amplifier in amplifying small signals (i.e., low-power RF signals). The branch containing this power amplifier is called the carrier branch. The second power amplifier is used to amplify large signals (i.e., high-power RF signals), thereby improving overall power amplification efficiency. The branch containing this power amplifier is called the peak branch.

[0004] When the carrier branch and the peak branch are coupled through a coupler, the carrier branch is electrically connected to the straight-through path of the coupler, and the peak branch is electrically connected to the coupling path of the coupler. In addition, a phase shift network is required to be added to the carrier branch to adjust the load impedance of the power amplifier, which increases the insertion loss of the carrier branch and reduces the efficiency of the amplifier circuit. Summary of the Invention

[0005] Embodiments of the present application provide a power amplifier circuit and an electronic device for reducing the insertion loss of a carrier branch when the carrier branch and the peak branch are coupled through a coupler.

[0006] To achieve the above objectives, the embodiments of the present application adopt the following technical solutions:

[0007] In a first aspect, a power amplifier circuit is provided, comprising: a first power amplifier, a second power amplifier, a matching network, a second phase-shifting network, a coupler, and a matching load, wherein the input end of the first power amplifier is used to input a radio frequency signal, the output end of the first power amplifier is electrically connected to the input end of the coupler, the through end of the coupler and the first end of the matching network are electrically connected to a combining point, the second end of the matching network is used to output the power-amplified radio frequency signal, the isolation end of the coupler is grounded through the matching load; the coupling end of the coupler is electrically connected to the combining point through the second phase-shifting network and the second power amplifier; when the first power amplifier performs power back-off, the second power amplifier is open-circuited.

[0008] The power amplifier circuit provided by the embodiment of the present application comprises a first power amplifier whose input end is used to input a radio frequency signal, an output end of the first power amplifier is electrically connected to the input end of the coupler, a straight-through end of the coupler and a first end of the matching network are electrically connected to a junction point, a second end of the matching network is used to output a radio frequency signal after power amplification, and the isolation end of the coupler is grounded through a matching load; the coupling end of the coupler is electrically connected to the junction point through a second phase-shifting network and a second power amplifier; when the first power amplifier performs power back-off, the second power amplifier is open-circuited. The branch where the first power amplifier is located is called the carrier branch, and the branch where the second power amplifier is located is called the peak branch. When the power is backed off, the peak branch is open-circuited, and the straight-through path of the coupler is equivalent to a quarter-wavelength line, which can achieve a 90-degree phase shift for the radio frequency signal at the center frequency, equivalent to a phase-shifting network, which plays the role of impedance transformation and can change the load impedance of the first power amplifier. Since the phase-shifting network electrically connected to the output end of the first power amplifier is cancelled, the insertion loss of the carrier branch is reduced.

[0009] In one possible implementation, a third power amplifier is further included. The coupling end of the coupler is electrically connected to the input end of the second power amplifier via the third power amplifier and a second phase-shifting network. Before reaching the power back-off point, the third power amplifier is deactivated, causing the peak branch to open. The coupler's through-path is equivalent to a quarter-wavelength line, which can replace the phase-shifting network electrically connected to the output end of the first power amplifier.

[0010] In one possible embodiment, the first power amplifier is a Class AB power amplifier, the second power amplifier is a Class B power amplifier or a Class C power amplifier, and the third power amplifier is a Class C power amplifier. Because the first power amplifier is a Class AB power amplifier, the first power amplifier begins operating when the input RF signal is a small signal. Because the second power amplifier is a Class B power amplifier or a Class C power amplifier, the second power amplifier begins operating when the input RF signal is a large signal. Because the third power amplifier is a Class C power amplifier, when the input RF signal is a small signal, the input RF signal power is low and insufficient to trigger the third power amplifier to operate, causing the third power amplifier to cut off and enter an open-circuit state. When the power of the input RF signal increases until it reaches the power back-off point, that is, when the RF signal transitions from a small signal to a large signal, the second and third power amplifiers are triggered to begin operating. Specifically, the first power amplifier is used to amplify small signals, while the second and third power amplifiers are used to amplify large signals. This allows the power amplifier circuit to maintain a high level of overall power amplification efficiency through power back-off. In addition, the third power amplifier adopts a Class C power amplifier, which requires a higher power of the input RF signal to be turned on. The peak branch is opened before the power back-off point is reached, thereby preventing the peak branch from being turned on prematurely before the power back-off point is reached.

[0011] In one possible implementation, the difference between the absolute value of the coupling coefficient of the coupler and the sum of the gains of the second power amplifier and the third power amplifier is within a preset range. Specifically, the absolute value of the coupling coefficient of the coupler is close to the sum of the gains of the second power amplifier and the third power amplifier. The increased power of the second and third power amplifiers just compensates for the power reduction at the coupling end of the coupler relative to the input end, and the power of the carrier branch and the peak branch at the combining point is balanced, achieving maximum coupler efficiency.

[0012] In one possible embodiment, the second phase-shifting network is a 180-degree phase-shifting network. A constant 90-degree phase difference is generated between the coupled end and the through-end of the coupler. The second power amplifier generates a phase shift of approximately 70 degrees, and the second phase-shifting network can generate a 180-degree phase shift. The phase difference between the peak branch and the carrier branch is then approximately 90 + 70 + 180 = 340 degrees, which is 20 degrees less than 360 degrees. The two branches merge nearly in phase at the junction, nearly maximizing the output power of the power amplifier circuit.

[0013] In one possible implementation, a third phase-shifting network is further included. The third phase-shifting network may be a 20-degree phase-shifting network, connected in series with the branch where the second power amplifier resides. This third phase-shifting network can produce a 20-degree phase shift. The phase difference between the peak branch and the carrier branch is approximately 90+70+180+20=360 degrees. The two branches merge nearly in phase at the junction, nearly maximizing the output power of the power amplifier circuit.

[0014] In one possible implementation, the first power amplifier is a class AB power amplifier, and the second power amplifier is a class C power amplifier. Because the second power amplifier is a class C power amplifier, it requires a higher input RF signal power to be turned on, and the peak branch is opened before the power back-off point is reached, thereby preventing the peak branch from being turned on prematurely before the power back-off point is reached.

[0015] In one possible implementation, the difference between the absolute value of the coupling coefficient of the coupler and the gain of the second power amplifier is within a preset range. That is, the absolute value of the coupling coefficient of the coupler is close to the gain of the second power amplifier. The increased power of the second power amplifier just compensates for the power reduction at the coupling end of the coupler relative to the input end. The power of the carrier branch and the peak branch at the combining point is balanced, and the efficiency of the coupler is maximized at this point.

[0016] In one possible embodiment, a fourth phase-shifting network is further included, wherein the first end of the fourth phase-shifting network is electrically connected to the combining point, and the second end of the fourth phase-shifting network is grounded. At the center frequency, the fourth phase-shifting network is grounded, and grounding is equivalent to 0 ohms. After the 90-degree phase shift of the fourth phase-shifting network, the equivalent impedance of the fourth phase-shifting network is infinite, which is equivalent to an open circuit and does not affect the equivalent impedance Ropt / 2 of the combining point. At frequencies other than the center frequency, the fourth phase-shifting network can achieve phase shifts at other angles. The equivalent impedance of the fourth phase-shifting network is not ideally infinite. The farther the frequency of the RF signal is from the center frequency, the lower the equivalent impedance of the fourth phase-shifting network. Since the fourth phase-shifting network is connected in parallel with the two branches, it will reduce the equivalent impedance of the combining point during power fallback, thereby increasing the output power of the power amplifier circuit, reducing the power loss caused by the misalignment of the combining point phase, and increasing the efficiency of the power amplifier circuit. Therefore, the bandwidth of the RF signal applicable to this power amplifier circuit is larger.

[0017] In a possible implementation, the fourth phase shift network is a quarter-wavelength line. At the center frequency, the quarter-wavelength line can achieve a 90-degree phase shift.

[0018] In one possible implementation, the second phase shift network is used to adjust the phase shift angle of the branch where the second power amplifier is located, so that at the center frequency of the RF signal, the branch where the first power amplifier is located and the branch where the second power amplifier is located are phase-aligned at the combining point, thereby improving the efficiency of the power amplifier circuit.

[0019] In a second aspect, an electronic device is provided, comprising an antenna and a power amplifier circuit as described in the first aspect and any embodiment thereof, wherein the power amplifier circuit is configured to output a power-amplified radio frequency signal to the antenna.

[0020] The technical effects of the second aspect refer to the technical effects of the first aspect and any of its embodiments and will not be repeated here. BRIEF DESCRIPTION OF THE DRAWINGS

[0021] Figure 1 A schematic diagram of the appearance of an electronic device provided in an embodiment of the present application;

[0022] Figure 2 A schematic diagram of the structure of an electronic device provided in an embodiment of the present application;

[0023] Figure 3 A schematic structural diagram of a mobile communication module provided in an embodiment of the present application;

[0024] Figure 4 A schematic structural diagram of a power amplifier circuit provided in an embodiment of the present application;

[0025] Figure 5 Schematic diagram of the structures of several quarter-wavelength lines provided in the embodiments of the present application;

[0026] Figure 6 A schematic diagram of a load impedance change of a power amplifier provided in an embodiment of the present application;

[0027] Figure 7 A schematic diagram of the relationship between power back-off and efficiency provided in an embodiment of the present application;

[0028] Figure 8 A schematic structural diagram of another power amplifier circuit provided in an embodiment of the present application;

[0029] Figure 9 A schematic structural diagram of another power amplifier circuit provided in an embodiment of the present application;

[0030] Figure 10 A schematic structural diagram of another power amplifier circuit provided in an embodiment of the present application;

[0031] Figure 11 A schematic diagram of a phase difference generated between a second power amplifier and a third power amplifier provided in an embodiment of the present application;

[0032] Figure 12 A schematic diagram of the structure of a 180-degree phase-shift network provided in an embodiment of the present application;

[0033] Figure 13A schematic structural diagram of another power amplifier circuit provided in an embodiment of the present application;

[0034] Figure 14 A schematic structural diagram of another power amplifier circuit provided in an embodiment of the present application;

[0035] Figure 15 A schematic structural diagram of another power amplifier circuit provided in an embodiment of the present application;

[0036] Figure 16 A schematic structural diagram of another power amplifier circuit provided in an embodiment of the present application;

[0037] Figure 17 A schematic diagram showing the variation of equivalent impedance with frequency provided in an embodiment of the present application;

[0038] Figure 18 A schematic diagram of impedance ratio variation with frequency provided in an embodiment of the present application. DETAILED DESCRIPTION

[0039] First, some concepts involved in this application are described.

[0040] The terms "first", "second", etc. involved in the embodiments of the present application are only used to distinguish features of the same type and cannot be understood as indicating relative importance, quantity, order, etc.

[0041] The terms "exemplary" or "for example" in the embodiments of this application are used to indicate examples, illustrations, or descriptions. Any embodiment or design described in this application as "exemplary" or "for example" should not be construed as being preferred or advantageous over other embodiments or designs. Rather, the use of words such as "exemplary" or "for example" is intended to present the relevant concepts in a concrete manner.

[0042] The terms "coupling" and "connection" involved in the embodiments of this application should be understood in a broad sense. For example, they may refer to a physical direct connection, or an indirect connection achieved through electronic devices, such as a connection achieved through resistors, inductors, capacitors or other electronic devices.

[0043] The saturation, linear, and nonlinear regions of a power amplifier. In the linear region, the output power of a power amplifier increases linearly with increasing input power. When the input power of the power amplifier increases further, the power amplifier enters the nonlinear region, where the output power of the power amplifier no longer increases with increasing input power, and the output power of the power amplifier reaches saturation.

[0044] Power amplifier classification: Power amplifiers can be divided into Class A, Class B, Class C, and Class AB based on the current conduction angle of the RF signal. Class A power amplifiers have a conduction angle of 360 degrees, meaning they are always on. Class A power amplifiers are suitable for scenarios with small RF signals. Class B power amplifiers have a conduction angle of 180 degrees. The quiescent bias point is the critical point between on and off. The power amplifier is on when the input RF signal power exceeds this quiescent bias point. Class C power amplifiers have a conduction angle less than 180 degrees and a higher quiescent bias point than Class B power amplifiers, requiring a higher input RF signal power to conduct. Class B and Class C power amplifiers are suitable for scenarios with large RF signals. Class AB power amplifiers have a conduction angle between 180 and 360 degrees. They are equivalent to Class A power amplifiers for small RF signals and Class B power amplifiers for large RF signals. The small signal refers to the power of the radio frequency signal input to the power amplifier being less than the threshold, and the large signal refers to the power of the radio frequency signal input to the power amplifier being greater than the threshold.

[0045] The efficiency of a power amplifier refers to the ratio of the output power to the input power of the power amplifier.

[0046] Power back-off: The 1dB compression point of the power amplifier (i.e., the critical point between the linear and nonlinear regions of the power amplifier) ​​is reduced by a certain amount (e.g., 6dB-10dB). This allows the power amplifier to enter the saturation region earlier as the input power increases, thereby improving the power amplifier's efficiency in amplifying small signals.

[0047] like Figure 1As shown, an embodiment of the present application provides an electronic device 101, which is an electronic device with wireless communication capabilities. The electronic device can be mobile or fixed. The electronic device can be deployed on land (e.g., indoors or outdoors, handheld or vehicle-mounted, etc.), on water (e.g., ships, etc.), or in the air (e.g., airplanes, balloons, and satellites, etc.). The electronic device can be referred to as user equipment (UE), access terminal, terminal unit, subscriber unit, terminal station, mobile station (MS), mobile station, terminal agent, or terminal device, etc. For example, the electronic device can be a mobile phone, tablet computer, laptop computer, smart bracelet, smart screen, smart watch, virtual reality (VR) device, augmented reality (AR) device, terminal in industrial control, terminal in self-driving, terminal in remote medical, terminal in smart grid, terminal in transportation safety, terminal in smart city, terminal in smart home, etc. The embodiments of the present application do not limit the specific type and structure of the electronic device. A possible structure of the electronic device is described below.

[0048] like Figure 1 As shown in Figure A, the electronic device 101 may include a front camera 2931 and a display screen 294. Figure 1 As shown in Figure B, the electronic device 101 may include a rear camera 2932. The front camera 2931 and the rear camera 2932 are used to capture static images or dynamic videos (collectively referred to as images). The display screen 294 is used to display images or receive user touch operations.

[0049] Taking mobile phones as an example, Figure 2The figure shows a possible structure of an electronic device 101. The electronic device 101 may include a processor 210, an external memory interface 220, an internal memory 221, a universal serial bus (USB) interface 230, a power management module 240, a battery 241, a wireless charging coil 242, antenna 1, antenna 2, a mobile communication module 250, a wireless communication module 260, an audio module 270, a speaker 270A, a receiver 270B, a microphone 270C, an earphone interface 270D, a sensor module 280, a button 290, a motor 291, an indicator 292, a camera 293, a display 294, and a subscriber identification module (SIM) card interface 295. Optionally, in some embodiments, an audio digital signal processor (ADSP) 243 is also included.

[0050] It should be understood that the structures illustrated in the embodiments of the present application do not constitute a specific limitation on the electronic device 101. In other embodiments of the present application, the electronic device 101 may include more or fewer components than shown, or may combine or separate certain components, or arrange the components differently. The illustrated components may be implemented in hardware, software, or a combination of software and hardware.

[0051] The processor 210 may include one or more processing units, such as a field programmable gate array (FPGA), an application specific integrated circuit (ASIC), a system on chip (SoC), a central processing unit (CPU), an application processor (AP), a network processor (NP), a digital signal processor (DSP), a microcontroller unit (MCU), a programmable logic device (PLD), a modem processor, a graphics processing unit (GPU), an image signal processor (ISP), a controller, a video codec, and a neural network processing unit (NPU). The different processing units may be independent devices or integrated into one or more processors. For example, the processor 210 may be an application processor (AP). Alternatively, the processor 210 may be integrated into a system on chip (SoC). Alternatively, the processor 210 may be integrated into an integrated circuit (IC) chip. The processor 210 may include an analog front end (AFE) and a micro-controller unit (MCU) in an IC chip.

[0052] The processor 210 executes the antenna feed control method provided in the embodiment of the present application by executing the program and computer instructions stored in the internal memory 221.

[0053] Processor 210 may also include a memory for storing computer instructions and data. In some embodiments, the memory in processor 210 is a cache memory. This memory can store computer instructions or data that have just been used or are being recycled by processor 210. If processor 210 needs to use the computer instructions or data again, it can directly access the memory. This avoids repeated accesses, reduces processor 210 latency, and thus improves system efficiency.

[0054] In some embodiments, the processor 210 may include one or more interfaces, including an inter-integrated circuit (I2C) interface, an inter-integrated circuit sound (I2S) interface, a pulse code modulation (PCM) interface, a universal asynchronous receiver / transmitter (UART) interface, a mobile industry processor interface (MIPI), a general-purpose input / output (GPIO) interface, a subscriber identity module (SIM) interface, and / or a USB interface.

[0055] The ADSP 243 can be coupled to the audio module 270 and the sensor module 280. The ADSP 243 can be used to process audio signals and sensor data. When the processor is in a dormant state, the ADSP 243 can still keep working, thereby reducing the power consumption of the electronic device.

[0056] It is understood that the interface connection relationship between the modules illustrated in the embodiments of the present application is merely an illustrative illustration and does not constitute a structural limitation on the electronic device 101. In other embodiments of the present application, the electronic device 101 may also adopt a different interface connection method from the above embodiments, or a combination of multiple interface connection methods.

[0057] External memory interface 220 can be used to connect an external memory card, such as a micro SanDisk (Micro SD) card, to expand the storage capacity of electronic device 101. The external memory card communicates with processor 210 via external memory interface 220 to implement data storage. For example, files such as music and videos can be stored on the external memory card.

[0058] The internal memory 221 can be used to store computer-executable program code, which includes computer instructions. The processor 210 executes the computer instructions stored in the internal memory 221 to perform various functional applications and data processing of the electronic device 101. In addition, the internal memory 221 can include high-speed random access memory and non-volatile memory, such as at least one disk storage device, flash memory device, universal flash storage (UFS), etc.

[0059] The memory involved in the embodiments of the present application may be a volatile memory or a non-volatile memory, or may include both volatile and non-volatile memories. The non-volatile memory may be a read-only memory (ROM), a programmable read-only memory (PROM), an erasable programmable read-only memory (EPROM), an electrically erasable programmable read-only memory (EEPROM), or a flash memory. The volatile memory may be a random access memory (RAM), which is used as an external cache. By way of example and not limitation, many forms of RAM are available, such as static RAM (SRAM), dynamic RAM (DRAM), synchronous DRAM (SDRAM), double data rate SDRAM (DDR SDRAM), enhanced SDRAM (ESDRAM), synchronous link DRAM (SLDRAM), and direct RAM bus RAM (DR RAM). It should be noted that the memory of the systems and methods described herein is intended to include, but is not limited to, these and any other suitable types of memory.

[0060] The electronic device 101 can implement audio functions such as music playback and recording through the audio module 270, the speaker 270A, the receiver 270B, the microphone 270C, the headphone jack 270D, and the application processor.

[0061] The audio module 270 is used to convert digital audio information into analog audio signal output, and is also used to convert analog audio input into digital audio signals. In some embodiments, the audio module 270 can be set in the processor 210, or some functional modules of the audio module 270 can be set in the processor 210. The speaker 270A, also known as the "speaker", is used to convert audio electrical signals into sound signals. The receiver 270B, also known as the "earpiece", is used to convert audio electrical signals into sound signals. The microphone 270C, also known as the "microphone" or "microphone", is used to convert sound signals into electrical signals. The electronic device 101 can be provided with at least one microphone 270C. The headphone jack 270D is used to connect wired headphones. The headphone jack 270D can be a USB interface 230, or it can be a 3.5mm open mobile terminal platform (OMTP) standard interface, or a cellular telecommunications industry association of the USA (CTIA) standard interface.

[0062] Keys 290 include a power button, volume button, and other buttons. Keys 290 can be mechanical or touch-sensitive. Electronic device 101 can receive key inputs and generate key signal inputs related to user settings and function control of electronic device 101. Motor 291 can generate vibration alerts. Motor 291 can be used for incoming call vibration alerts or for touch vibration feedback. Indicator 292 can be an indicator light that can indicate charging status, battery level changes, messages, missed calls, notifications, and more. SIM card interface 295 is used to connect a SIM card. A SIM card can be connected to and disconnected from electronic device 101 by inserting or removing it from SIM card interface 295. Electronic device 101 can support one or N SIM card interfaces, where N is a positive integer greater than one. SIM card interface 295 can support nano SIM cards, micro SIM cards, and SIM cards. In some embodiments, the electronic device 101 uses an embedded SIM (eSIM) card. The eSIM card can be embedded in the electronic device 101 and cannot be separated from the electronic device 101.

[0063] The electronic device 101 can implement a shooting function through an ISP, a camera 293, a video codec, a GPU, a display screen 294, and an application processor. The ISP is used to process data fed back by the camera 293. In some embodiments, the ISP can be set in the camera 293. The camera 293 is used to capture still images or videos. In some embodiments, the electronic device 101 may include 1 or N cameras 293, where N is a positive integer greater than 1, for example Figure 1 The front camera 2931 and the rear camera 2932 are shown.

[0064] Electronic device 101 can implement display functions through a GPU, display screen 294, and an application processor. A GPU is a microprocessor for image processing that connects display screen 294 and the application processor. The GPU is used to perform mathematical and geometric calculations for graphics rendering. Processor 210 may include one or more GPUs that execute computer instructions to generate or modify display information.

[0065] The sensor module 280 may include a pressure sensor, a gyroscope sensor, an air pressure sensor, a magnetic sensor, an acceleration sensor, a distance sensor, a proximity light sensor, a fingerprint sensor, a temperature sensor, a touch sensor, an ambient light sensor, an angle sensor, etc. When the display screen 294 is a foldable screen, the angle sensor can detect the folding angle of the display screen 294, and the folding angle range is 0-180 degrees.

[0066] The battery 241 may include one or more batteries to power the load.

[0067] The power management module 240 is configured to receive charging input from a charger. The charger can be a wireless charger, such as a wireless charging dock or another electronic device 101 with reverse wireless charging functionality. The power management module 240 can receive wireless charging input via the electronic device's wireless charging coil 242. Alternatively, the charger can be a wired charger, for example, via the USB port 230. The power management module 240 is also referred to as a charging chip.

[0068] The power management module 240 not only charges the battery 241 but also provides power to the electronic device. The power management module 240 receives input from the battery 241 and provides power to the processor 210, internal memory 221, external memory interface 220, display 294, camera 293, and wireless communication module 260. The power management module 240 can also monitor parameters such as the battery 241's capacity, voltage, battery cycle count, and battery health status (leakage, impedance). In other embodiments, the power management module 240 can also be located within the processor 210.

[0069] The display screen 294 is used to display images, videos, etc. The display screen 294 includes a display panel. In some embodiments, the electronic device 101 may include one or more display screens 294 .

[0070] The wireless communication function of the electronic device 101 can be implemented through the antenna 1, the antenna 2, the mobile communication module 250, the wireless communication module 260, the modulation and demodulation processor, etc.

[0071] Antenna 1 and Antenna 2 are used to transmit and receive electromagnetic wave signals. Each antenna in electronic device 101 can be used to cover a single or multiple communication frequency bands. Different antennas can also be reused to improve antenna utilization. For example, antenna 1 can be reused as a diversity antenna for a wireless local area network. In other embodiments, the antennas can be used in conjunction with a tuning switch.

[0072] The mobile communication module 250 can provide wireless communication solutions including 2G / 3G / 4G / 5G for application in the electronic device 101. The wireless communication module 260 can provide wireless communication solutions including wireless local area networks (WLAN) (such as wireless fidelity (Wi-Fi) networks), Bluetooth (BT), global navigation satellite system (GNSS), frequency modulation (FM), near field communication (NFC), infrared (IR), etc. for application in the electronic device 101. In some embodiments, the antenna 1 of the electronic device 101 is coupled to the mobile communication module 250, and the antenna 2 is coupled to the wireless communication module 260, so that the electronic device 101 can communicate with the network and other devices through wireless communication technology.

[0073] like Figure 3 As shown, the mobile communication module 250 may include a baseband chip 31 , a radio frequency integrated circuit (RFIC) 32 and a power amplifier circuit 33 .

[0074] The baseband chip 31 converts data from the processor 210 into baseband signals, performing modulation and demodulation, digital filtering, and equalization on the baseband signals. The RFIC 32 converts the baseband signals from the baseband chip 31 into RF signals, which are then transmitted via the power amplifier circuit 33 and antenna 34. The power amplifier circuit 33 improves the efficiency of amplifying the RF signals during power backoff. It should be noted that the baseband chip 31 and RFIC 32 can also be integrated with the processor 210 in a system-on-chip (SoC).

[0075] like Figure 4 As shown, the power amplifier circuit 33 includes: a first power amplifier (also called a carrier power amplifier) ​​3301, a second power amplifier (also called a peak power amplifier) ​​3302, a matching network 3303, a first phase shift network 3304, a power divider 3305 and a second phase shift network 3306.

[0076] The matching network 3303 is used to perform impedance transformation on the equivalent impedance (eg, 10 ohms) of the combining point CON to obtain an equivalent impedance RL (eg, 50 ohms), thereby achieving impedance matching.

[0077] The input end of the power divider 3305 is used to Figure 3 The RFIC 32 shown is electrically connected to input a radio frequency signal. The first output terminal of the power divider 3305 is electrically connected to the input terminal of the first power amplifier 3301, and the second output terminal of the power divider 3305 is electrically connected to the first terminal of the second phase shift network 3306. The second terminal of the second phase shift network 3306 is electrically connected to the input terminal of the second power amplifier 3302. The output terminal of the first power amplifier 3301 is electrically connected to the first terminal of the first phase shift network 3304. The second terminal of the first phase shift network 3304, the output terminal of the second power amplifier 3302, and the first terminal of the matching network 3303 are electrically connected to the junction point CON. The second terminal of the matching network 3303 is used to connect to the junction point CON. Figure 3 The antenna 34 is shown electrically connected, and the second end of the matching network 3303 is used to output the power-amplified radio frequency signal.

[0078] The power divider 3305 is configured to evenly divide the RF signal inputted at the input terminal into a first RF signal and a second RF signal, and output the first RF signal through the first output terminal and the second RF signal through the second output terminal. The power of the first RF signal and the second RF signal are both half the power of the input RF signal.

[0079] The first power amplifier 3301 is used to power amplify the first RF signal. The first power amplifier 3301 is a class AB power amplifier, that is, the first power amplifier 3301 is equivalent to a class A power amplifier when the first RF signal is a small signal, and is equivalent to a class B power amplifier when the first RF signal is a large signal. The second power amplifier 3302 is used to power amplify the second RF signal. The second power amplifier 3302 is a class B power amplifier or a class C power amplifier, that is, the second power amplifier 3302 begins to operate when the second RF signal is a large signal. The first power amplifier 3301 and the second power amplifier 3302 have the same load impedance when saturated, for example, both are pure resistors Ropt, which achieves the highest efficiency. The load impedance of a power amplifier refers to the output voltage / output current of the power amplifier.

[0080] Both the first phase-shift network 3304 and the second phase-shift network 3306 are frequency components, i.e., components whose load impedance and phase shift angle vary with the frequency of the RF signal. The first phase-shift network 3304 performs impedance transformation, for example, changing the load impedance of the first power amplifier 3301. The first phase-shift network 3304 or the second phase-shift network 3306 can be a quarter-wavelength line, where the wavelength refers to the wavelength corresponding to the center frequency of the RF signal. A quarter-wavelength line, also known as a 90-degree phase-shift line, can produce a 90-degree phase shift on the RF signal at the center frequency of the RF signal.

[0081] First phase-shifting network 3304 performs a phase shift on the first RF signal, with the phase shift angle being related to the frequency of the RF signal. At the center frequency of the RF signal, first phase-shifting network 3304 can produce a 90-degree phase shift on the first RF signal. The impedance of first phase-shifting network 3304 is equal to the load impedance of first power amplifier 3301 in saturation, thereby achieving impedance matching with a reflection coefficient of zero.

[0082] The second phase shift network 3306 performs a phase shift on the second RF signal, and the phase shift angle is related to the frequency of the RF signal. The second phase shift network 3306 is used to adjust the phase shift angle of the peak branch so that at the center frequency of the RF signal, the carrier branch and the peak branch are phase-aligned at the junction CON, thereby improving the efficiency of the power amplifier circuit 33. Specifically, at the center frequency of the RF signal, the second phase shift network 3306 can produce a 90-degree phase shift on the second RF signal, and the impedance of the second phase shift network 3306 is equal to the load impedance of the first power amplifier 3301 when saturated. Under ideal conditions, for any frequency of the RF signal, the phase shift angle of the second phase shift network 3306 on the second RF signal is the same as the phase shift angle of the first phase shift network 3304 on the first RF signal, so that the first RF signal and the second RF signal are combined in phase at the junction CON to maximize the output power.

[0083] In the embodiment of the present application, the branch where the first power amplifier 3301 is located is called a carrier branch, and the branch where the second power amplifier 3302 is located is called a peak branch.

[0084] For example, Figure 5 As shown, the embodiments of the present application provide several implementation methods of quarter-wavelength lines. Figure 5 As shown in Figure A, the quarter-wavelength line can be a microstrip line with an electrical length of 90 degrees. This quarter-wavelength line can achieve a -90-degree phase shift on the RF signal at the center frequency, and the phase shift angle changes linearly at other frequencies.

[0085] like Figure 5 As shown in Figure B, the quarter-wavelength line can be a series inductor π-type network, including capacitors C1, C2, and inductor L1. The first end of inductor L1 is electrically connected to the first end of capacitor C1 and serves as the input of the series inductor π-type network. The second end of capacitor C1 is grounded. The second end of inductor L1 is electrically connected to the first end of capacitor C2 and serves as the output of the series inductor π-type network. The second end of capacitor C2 is grounded. This quarter-wavelength line can achieve a -90-degree phase shift on the RF signal at the center frequency, and the phase shift angle varies linearly at other frequencies.

[0086] like Figure 5 As shown in Figure C, the quarter-wavelength line can be a π-type network of series capacitors, including capacitor C1, inductor L1, and inductor L2. The first end of capacitor C1 is electrically connected to the first end of inductor L1 and serves as the input of the π-type network. The second end of inductor L1 is grounded. The second end of capacitor C1 is electrically connected to the first end of inductor L2 and serves as the output of the π-type network. The second end of inductor L2 is grounded. This quarter-wavelength line can achieve a 90-degree phase shift on the RF signal at the center frequency, and the phase shift angle varies linearly at other frequencies.

[0087] like Figure 5 As shown in Figure D, the quarter-wavelength line can be a series inductor T-type network, including capacitor C1, inductor L1, and inductor L2. The first end of inductor L1 serves as the input end of the series inductor T-type network, the second end of inductor L1, the first end of inductor L2, and the first end of capacitor C1 are electrically connected, the second end of capacitor C1 is grounded, and the second end of inductor L2 serves as the output end of the series inductor T-type network. This quarter-wavelength line can achieve a -90-degree phase shift on the RF signal at the center frequency, and the phase shift angle changes linearly at other frequencies.

[0088] like Figure 5As shown in Figure E, the quarter-wavelength line can be a series capacitor T-type network, including capacitors C1, C2, and L1. The first end of capacitor C1 serves as the input of the series capacitor T-type network. The second end of capacitor C1, the first end of capacitor C2, and the first end of inductor L1 are electrically connected. The second end of inductor L1 is grounded. The second end of capacitor C2 serves as the output of the series capacitor T-type network. This quarter-wavelength line can achieve a 90-degree phase shift on the RF signal at the center frequency, and the phase shift angle changes linearly at other frequencies.

[0089] Figure 5 The inductors in each network shown satisfy L=Ropt / (2πf), and the capacitors satisfy C=1 / (2πfRopt), where f is the center frequency of the RF signal and Ropt is the load impedance when the first power amplifier 3301 is saturated.

[0090] The working principle of the power amplifier circuit 33 to achieve power amplifier power back-off is as follows:

[0091] The load impedance of the power amplifier is equal to the output voltage / output current of the power amplifier, such as Figure 6 As shown in Figure A, when the load impedance of the power amplifier remains unchanged (that is, the slope of the output voltage / output current remains unchanged), when the output current of the power amplifier is small, the output voltage is also small. At this time, the efficiency of the power amplifier is low. Figure 6 As shown in B, when the output current of the power amplifier is small, the efficiency of the power amplifier can be improved by increasing the load impedance of the power amplifier and then increasing the output voltage of the power amplifier, making it easier for the power amplifier to enter the saturation region earlier.

[0092] Therefore, if Figure 7As shown in the MN section of the "Power Amplifier Circuit" in FIG. 2 , because the second power amplifier 3302 is a Class B or Class C power amplifier, when the input RF signal is a small signal, the power of the input RF signal is low and insufficient to trigger the second power amplifier 3302 to operate. The second power amplifier 3302 is cut off and enters an open circuit state, and the load impedance of the second power amplifier 3302 is infinite (∞). Because the first power amplifier 3301 is a Class AB power amplifier, when the input RF signal is a small signal, the first power amplifier 3301 begins to operate. The first phase shift network 3304 transforms the load impedance of the first power amplifier 3301, increasing the load impedance of the first power amplifier 3301. This increases the output voltage of the first power amplifier 3301, causing the first power amplifier 3301 to enter the saturation state earlier. In other words, as the input power increases, the first power amplifier 3301 is more likely to enter the saturation region earlier, improving the efficiency of the first power amplifier 3301 in amplifying small signals, thereby improving the overall power amplification efficiency of the power amplifier circuit 33. The output voltage of the first power amplifier 3301 increases as the power of the input radio frequency signal increases, and the output power of the first power amplifier 3301 also increases as the power of the input radio frequency signal increases.

[0093] like Figure 7 As shown at the mid-power back-off point N, because second power amplifier 3302 is a Class B or Class C power amplifier, when the input RF signal is small, the input RF signal power is too low to trigger second power amplifier 3302. Second power amplifier 3302 is cut off and becomes open-circuited, and the load impedance of second power amplifier 3302 is infinite (∞). After impedance transformation by first phase-shifting network 3304, the load impedance of the first power amplifier 3301, Ropt / 2, at the combining point CON, becomes 2Ropt. The output power of power amplifier circuit 33 is backed off by 6 dB, meaning that the output power of power amplifier circuit 33 is one-quarter of the saturated output power.

[0094] like Figure 7As shown in the NP segment of the "Power Amplifier Circuit" in the figure, the power of the RF signal input to the power amplifier circuit 33 is relatively high, triggering the second power amplifier 3302 to start operating. As the power of the input RF signal continues to increase, the load impedance of the first power amplifier 3301 decreases from 2Ropt to Ropt, and the load impedance of the second power amplifier 3302 decreases from infinity (∞) to Ropt. Due to its high output voltage, the first power amplifier 3301 remains in a pre-saturated state, and the output voltage remains essentially constant. The output current increases with the increase in the power of the input RF signal. Therefore, the output power of the first power amplifier 3301 increases with the increase in the power of the input RF signal. The output voltage of the second power amplifier 3302 increases with the increase in the power of the input RF signal. Therefore, the output power of the second power amplifier 3302 increases with the increase in the power of the input RF signal. The overall power amplification efficiency of the power amplifier circuit 33 can still be maintained at a high level.

[0095] It should be noted that at the power back-off point N, the output power of the power amplifier circuit 33 is one-fourth of the saturated output power. This is because, compared with the first power amplifier 3301 and the second power amplifier 3302 both reaching saturation, at the power back-off point N, only the first power amplifier 3301 is working, so the output power of the entire power amplifier circuit 33 is halved. Since the load impedance of the first power amplifier 3301 is 2Ropt, which is twice the load impedance Ropt at saturation, the output power of the first power amplifier 3301 is half of the saturated output power. The corresponding ratio of the output power of the first power amplifier at the power back-off point N is 0.5, which further reduces the output power of the power amplifier circuit 33 by half.

[0096] like Figure 7 As shown in the power saturation point P of the "Power Amplifier Circuit" in Figure 3, when the power of the input RF signal continues to increase, causing both first power amplifier 3301 and second power amplifier 3302 to reach saturation, the load impedance of first power amplifier 3301 and the load impedance of second power amplifier 3302 are both Ropt. Because the carrier branch and the peak branch both have the same phase shift network, the RF signals of the two branches are phase-aligned at the combining point CON. The equivalent impedance of the combining point CON is Ropt / 2, and the overall power amplification efficiency of the power amplifier circuit 33 is maximized.

[0097] for Figure 7 Although the class B power amplifier or class AB power amplifier shown has the highest efficiency at the power saturation point P, when the power is backed off, the efficiency will drop significantly as the power back-off increases. Figure 4 Compared with the class B power amplifier and the class AB power amplifier, the power amplifier circuit shown not only has the highest efficiency at the power saturation point P, but also has higher efficiency in power amplification throughout the entire MP section.

[0098] like Figure 8 As shown, an embodiment of the present application provides another power amplifier circuit 33, including: a first power amplifier (also called a carrier power amplifier) ​​3301, a second power amplifier (also called a peak power amplifier) ​​3302, a matching network 3303, a first phase shift network 3304, a second phase shift network 3306, a coupler 3307, and a matching load R.

[0099] The input terminal of the first power amplifier 3301 is used to Figure 3 The RFIC 32 shown is electrically connected to input a radio frequency signal. The output end of the first power amplifier 3301 is electrically connected to the input end of the coupler 3307, the through end of the coupler 3307 is electrically connected to the first end of the first phase-shift network 3304, the coupling end of the coupler 3307 is electrically connected to the first end of the second phase-shift network 3306, and the isolation end of the coupler 3307 is grounded through the matching load R. The second end of the second phase-shift network 3306 is electrically connected to the input end of the second power amplifier 3302. The second end of the first phase-shift network 3304, the output end of the second power amplifier 3302, and the first end of the matching network 3303 are electrically connected to the junction CON. The second end of the matching network 3303 is used to connect to the Figure 3 The antenna 34 is electrically connected, and the second end of the matching network 3303 is used to output the power-amplified RF signal. Figure 4 The relevant description will not be repeated here.

[0100] During the design phase, the coupling coefficient of coupler 3307 needs to be close to the gain of second power amplifier 3302. Specifically, the absolute value of the coupling coefficient of coupler 3307 should be the difference (in dB) between the gain of second power amplifier 3302 and the power of second power amplifier 3302. Within a preset range, the power added by second power amplifier 3302 should just compensate for the power loss at the coupled end of coupler 3307 relative to the input end, achieving power balance between the carrier branch and the peak branch at the combining point CON. This is when coupler 3307 achieves maximum efficiency. For example, if the gain of second power amplifier 3302 is ≥10 dB, the coupling coefficient of coupler 3307 must also be ≥10 dB. The coupling coefficient of coupler 3307 refers to the difference (in dB) between the power of the output signal at the coupled end of coupler 3307 and the power of the output signal at the through end. Figure 8 The circuit shown is relative to Figure 4The circuit shown in FIG3 removes the power divider 3305, thereby avoiding the 3dB insertion loss caused by the power divider 3305 to the carrier branch and the peak branch, thereby helping to improve the efficiency of the power amplifier circuit 33. However, due to the addition of the coupler 3307, the direct connection from the output end of the first power amplifier 3301 to the coupler 3307 will also introduce insertion loss, reducing the efficiency of the power amplifier circuit 33.

[0101] For this reason, Figure 9 and Figure 10 As shown, an embodiment of the present application provides another power amplifier circuit 33, including: a first power amplifier (also called a carrier power amplifier) ​​3301, a second power amplifier (also called a peak power amplifier) ​​3302, a matching network 3303, a second phase shift network 3306, a coupler 3307, and a matching load R. Figure 9 and Figure 10 The power amplifier circuit 33 shown is removed Figure 8 The first phase shift network 3304 is shown.

[0102] The input terminal of the first power amplifier 3301 is used to Figure 3 The RFIC 32 shown is electrically connected to input an RF signal. The output of the first power amplifier 3301 is electrically connected to the input of the coupler 3307. The through-end of the coupler 3307 and the first end of the matching network 3303 are electrically connected to the junction CON. The second end of the matching network 3303 is used to output the power-amplified RF signal. The isolation end of the coupler 3307 is grounded via the matching load R. The coupled end of the coupler 3307 is electrically connected to the junction CON via the second phase-shifting network 3306 and the second power amplifier 3302. The first power amplifier 3301 is a class AB power amplifier, and the second power amplifier 3302 is a class C power amplifier.

[0103] It should be noted that the embodiment of the present application does not limit the connection relationship between the second phase shift network 3306 and the second power amplifier 3302. Figure 9 As shown, the coupling end of the coupler 3307 is electrically connected to the input end of the second power amplifier 3302 through the second phase shift network 3306, and the output end of the second power amplifier 3302 is electrically connected to the junction point CON. Figure 10 As shown, the coupling end of the coupler 3307 is electrically connected to the input end of the second power amplifier 3302, and the output end of the second power amplifier 3302 is electrically connected to the combining point CON through the second phase shift network 3306.

[0104] Ideally, at the center frequency of the RF signal, the carrier branch and the peak branch are phase-aligned at the junction CON, that is, the two branches have a phase difference of 0 or 360 degrees at the junction CON, so that the two branches are combined in phase at the junction CON to maximize the output power of the power amplifier circuit 33. Taking the RF signal frequency of 3.3 GHz to 4.2 GHz as an example, a constant phase difference of 90 degrees will be generated between the coupling end and the through end of the coupler 3307. Figure 11 As shown, at different input voltages (1.0V-1.3V) and frequencies, second power amplifier 3302 produces a phase shift of approximately 70 degrees. Therefore, the function of second phase shift network 3306 is to adjust the phase shift angle of the peak branch so that the carrier branch and the peak branch are as closely aligned in phase as possible at the combining point CON at the center frequency of the RF signal.

[0105] Second phase-shift network 3306 can be a 180-degree phase-shift network to produce a 180-degree phase shift. The phase difference between the peak branch and the carrier branch is approximately 90+70+180=340 degrees, which is 20 degrees away from 360 degrees. The two branches merge nearly in phase at the combining point CON, nearly maximizing the output power of power amplifier circuit 33.

[0106] Figure 8 or Figure 9 The working principle of the power amplifier circuit 33 to achieve power amplifier power back-off is as follows:

[0107] like Figure 7 As shown in the mid-power back-off point N, because the second power amplifier 3302 is a Class C power amplifier, when the input RF signal is a small signal, the power of the input RF signal is small and insufficient to trigger the second power amplifier 3302 to work. The second power amplifier 3302 is cut off, causing the peak branch to be in an open-circuit state. The second phase-shifting network 3306 produces a 180-degree phase shift, which will not affect the open-circuit characteristics of the second power amplifier 3302, and realizes that the peak branch and the carrier branch are nearly in-phase merged at the junction CON, nearly maximizing the output power of the power amplifier circuit 33. At this time, the straight-through path of the coupler 3307 is equivalent to a quarter-wavelength line, which can achieve a 90-degree phase shift for the RF signal at the center frequency. That is, the straight-through path of the coupler 3307 is equivalent to the first phase-shifting network 3304, but the first phase-shifting network 3304 is reduced in comparison, so the insertion loss of the carrier branch is reduced.

[0108] like Figure 7As shown at the mid-power saturation point P, when the power of the input RF signal continues to increase, causing both the first power amplifier 3301 and the second power amplifier 3302 to reach saturation, the load impedance of the first power amplifier 3301 and the load impedance of the second power amplifier 3302 are both Ropt. Because the carrier branch and the peak branch both have the same phase shift network, the RF signals of the two branches are phase-aligned at the combining point CON. The equivalent impedance of the combining point CON is Ropt / 2, and the overall power amplification efficiency of the power amplifier circuit 33 is maximized.

[0109] For example, Figure 12 As shown, the embodiments of the present application provide several implementations of a 180-degree phase-shift network.

[0110] like Figure 12 As shown in Figure A, the 180-degree phase-shift network can be a microstrip line with an electrical length of 180 degrees. This 180-degree phase-shift network can achieve a 180-degree phase shift on the RF signal at the center frequency, and the phase shift angle changes linearly at other frequencies.

[0111] like Figure 12 As shown in Figure B, the 180-degree phase-shift network can be a series inductor π-type network, including capacitor C1, capacitor C2, capacitor C3, inductor L1, and inductor L2. The first end of inductor L1 is electrically connected to the first end of capacitor C1 and serves as the input end of the series inductor π-type network. The second end of capacitor C1 is grounded, the second end of inductor L1, the first end of inductor L2, and the first end of capacitor C2 are electrically connected, the second end of capacitor C2 is grounded, the second end of inductor L2 is electrically connected to the first end of capacitor C3, and serves as the output end of the series inductor π-type network. The capacitance value of capacitor C2 is 2C, the capacitance values ​​of capacitors C1 and C3 are C, and the inductance of inductor L1 and inductor L2 is L. This 180-degree phase-shift network can achieve a 180-degree phase shift for the RF signal at the center frequency, and the phase shift angle will change linearly at other frequencies.

[0112] like Figure 12As shown in C, the 180-degree phase-shift network can be a series capacitor π-type network, including capacitor C1, capacitor C2, inductor L1, inductor L2, and inductor L3. The first end of capacitor C1 is electrically connected to the first end of inductor L1 and serves as the input end of the series capacitor π-type network. The second end of inductor L1 is grounded. The second end of capacitor C1, the first end of capacitor C2, and the first end of inductor L2 are electrically connected. The second end of inductor L2 is grounded. The second end of capacitor C2 is electrically connected to the first end of inductor L3 and serves as the output end of the series capacitor π-type network. The second end of inductor L3 is grounded. The capacitance value of capacitor C1 and capacitor C2 is C, the inductance of inductor L1 and inductor L3 is L, and the inductance of inductor L2 is L / 2. This 180-degree phase-shift network can achieve a 180-degree phase shift for the RF signal at the center frequency, and the phase shift angle will change linearly at other frequencies.

[0113] like Figure 12 As shown in D, the 180-degree phase-shift network can be a series inductor T-type network, including capacitor C1, capacitor C2, inductor L1, inductor L2, and inductor L3. The first end of inductor L1 serves as the input end of the series inductor T-type network, the second end of inductor L1 and the first end of inductor L2 are electrically connected to the first end of capacitor C1, the second end of capacitor C1 is grounded, the second end of inductor L2 and the first end of inductor L3 are electrically connected to the first end of capacitor C2, the second end of capacitor C2 is grounded, and the second end of inductor L3 serves as the output end of the series inductor T-type network. The capacitance values ​​of capacitors C1 and C2 are C, the inductance values ​​of inductors L1 and L3 are L, and the inductance value of inductor L2 is 2L. This 180-degree phase-shift network can achieve a 180-degree phase shift on the RF signal at the center frequency, and the phase shift angle will change linearly at other frequencies.

[0114] like Figure 12 As shown in Figure E, the 180-degree phase-shift network can be a series capacitor T-type network, including capacitor C1, capacitor C2, capacitor C3, inductor L1, and inductor L2. The first end of capacitor C1 serves as the input end of the series capacitor T-type network, the second end of capacitor C1, the first end of capacitor C2, and the first end of inductor L1 are electrically connected, the second end of inductor L1 is grounded, the second end of capacitor C2, the first end of capacitor C3, and the first end of inductor L2 are electrically connected, the second end of inductor L2 is grounded, and the second end of capacitor C3 serves as the output end of the series capacitor T-type network. The capacitance values ​​of capacitors C1 and C3 are C, the capacitance value of capacitor C2 is C / 2, and the inductance values ​​of inductors L1 and L2 are L. This 180-degree phase-shift network can achieve a 180-degree phase shift for the RF signal at the center frequency, and the phase shift angle will change linearly at other frequencies.

[0115] Figure 12In each network shown, L=Ropt / (2πf), C=1 / (2πfRopt), where f is the center frequency of the RF signal and Ropt is the load impedance of the first power amplifier 3301 when it is saturated.

[0116] Figure 9 or Figure 10 The working principle of the power amplifier circuit 33 to achieve power amplifier power back-off is as follows:

[0117] When the input RF signal is small, the power of the input RF signal is too low to trigger the operation of second power amplifier 3302. Second power amplifier 3302 is cut off and becomes open-circuited, and the load impedance of second power amplifier 3302 is infinite (∞). Therefore, the peak branch is also open-circuited, and the straight-through path of coupler 3307 is equivalent to a quarter-wavelength line, which can achieve a 90-degree phase shift on the RF signal at the center frequency. The straight-through path of coupler 3307 functions similarly to the first phase shift network 3304, but by eliminating the first phase shift network 3304, the insertion loss of the carrier branch is reduced. When the first power amplifier 3301 begins operating, the through path of the coupler 3307 transforms the load impedance of the first power amplifier 3301, increasing the load impedance of the first power amplifier 3301. This increases the output voltage of the first power amplifier 3301, causing the first power amplifier 3301 to enter a saturation state earlier. Specifically, as the input power increases, the first power amplifier 3301 is more likely to enter the saturation region earlier, thereby improving the efficiency of the first power amplifier 3301 in amplifying small signals, and thereby improving the overall power amplification efficiency of the power amplifier circuit 33. The output voltage of the first power amplifier 3301 increases as the power of the input RF signal increases, and the output power of the first power amplifier 3301 also increases as the power of the input RF signal increases.

[0118] When the input RF signal is a large signal, the power of the RF signal is relatively high, triggering the second power amplifier 3302 to start operating. As the power of the input RF signal continues to increase, the load impedance of the first power amplifier 3301 decreases from 2Ropt to Ropt, and the load impedance of the second power amplifier 3302 decreases from infinity (∞) to Ropt. Due to its high output voltage, the first power amplifier 3301 remains in a pre-saturated state, the output voltage remains essentially constant, and the output current increases with the increase in the power of the input RF signal. Therefore, the output power of the first power amplifier 3301 increases with the increase in the power of the input RF signal. The output voltage of the second power amplifier 3302 increases with the increase in the power of the input RF signal. Therefore, the output power of the second power amplifier 3302 increases with the increase in the power of the input RF signal. The overall power amplification efficiency of the power amplifier circuit 33 can still be maintained at a high level.

[0119] Optional, such as Figure 13 or Figure 14 As shown, the power amplifier circuit 33 further includes a third phase shift network 3308, which is connected in series to the branch (peak branch) where the second power amplifier 3302 is located. Figure 13 As shown, the output end of the second power amplifier 3302 is electrically connected to the junction point CON through the third phase shift network 3308, as shown in FIG. Figure 14 As shown, the coupling end of the coupler 3307 is electrically connected to the input end of the second power amplifier 3302 through the third phase shift network 3308.

[0120] The third phase-shift network 3308 can be a 20-degree phase-shift network to produce a 20-degree phase shift. The phase difference between the peak branch and the carrier branch is approximately 90+70+180+20=360 degrees. The two branches merge in phase at the combining point CON, nearly maximizing the output power of the power amplifier circuit 33.

[0121] like Figure 15 As shown, an embodiment of the present application provides another power amplifier circuit 33, including: a first power amplifier (also called a carrier power amplifier) ​​3301, a second power amplifier (also called a peak power amplifier) ​​3302, a matching network 3303, a second phase shift network 3306, a coupler 3307, a third power amplifier (also called a driving power amplifier) ​​3309, and a matching load R.

[0122] The input terminal of the first power amplifier 3301 is used to Figure 3 The RFIC 32 shown is electrically connected to input a radio frequency signal. The output end of the first power amplifier 3301 is electrically connected to the input end of the coupler 3307. The through end of the coupler 3307, the output end of the second power amplifier 3302, and the first end of the matching network 3303 are electrically connected to the junction CON. The coupling end of the coupler 3307 is electrically connected to the input end of the third power amplifier 3309. The isolation end of the coupler 3307 is grounded through the matching load R. The output end of the third power amplifier 3309 is electrically connected to the first end of the second phase shift network 3306. The second end of the second phase shift network 3306 is electrically connected to the input end of the second power amplifier 3302. The second end of the matching network 3303 is used to connect to the Figure 3 The antenna 34 is shown electrically connected, and the second end of the matching network 3303 is used to output the power-amplified radio frequency signal.

[0123] The first power amplifier 3301 is a Class AB power amplifier, the second power amplifier 3302 is a Class B or Class C power amplifier, and the third power amplifier 3309 is a Class C power amplifier. Because the first power amplifier 3301 is a Class AB power amplifier, when the input RF signal is a small signal, the first power amplifier 3301 begins to operate. Because the second power amplifier 3302 is a Class B or Class C power amplifier, when the input RF signal is a large signal, the second power amplifier 3302 begins to operate. Because the third power amplifier 3309 is a Class C power amplifier, when the input RF signal is a small signal, the input RF signal power is too low to trigger the third power amplifier 3309 to operate, causing the third power amplifier 3309 to cut off and enter an open-circuit state. When the power of the input RF signal increases until it reaches the power back-off point, that is, when the RF signal changes from a small signal to a large signal, the second power amplifier 3302 and the third power amplifier 3309 are triggered to start operating. Specifically, first power amplifier 3301 is used to amplify small signals, while second power amplifier 3302 and third power amplifier 3309 are used to amplify large signals. This allows the power amplifier circuit to maintain a high level of overall power amplification efficiency through power back-off. Furthermore, third power amplifier 3309 utilizes a Class C power amplifier, requiring a higher input RF signal power to conduct. The peak branch is opened before the power back-off point is reached, preventing the peak branch from conducting prematurely.

[0124] The absolute value of the coupling coefficient of the coupler 3307 is close to the sum of the gain of the second power amplifier 3302 and the gain of the third power amplifier 3309. That is, the difference between the absolute value of the coupling coefficient of the coupler 3307 and the sum of the gain of the second power amplifier 3302 and the gain of the third power amplifier 3309 is within a preset range. The increased power of the second power amplifier 3302 and the third power amplifier 3309 just compensates for the power reduction of the coupling end of the coupler 3307 relative to the input end. The power of the carrier branch and the peak branch at the junction CON reaches equilibrium. At this time, the efficiency of the coupler 3307 is the highest. For other contents about the first power amplifier 3301, the second power amplifier 3302, the matching network 3303, and the second phase shift network 3306, please refer to Figure 4 The relevant description will not be repeated here.

[0125] The second phase shift network 3306 is located between the second power amplifier 3302 and the third power amplifier 3309, and does not affect the open-circuit characteristics of the third power amplifier 3309 and the second power amplifier 3302, so that the peak branch remains in an open-circuit state.

[0126] Figure 15The working principle of the power amplifier circuit 33 to achieve power amplifier power back-off is as follows:

[0127] like Figure 7 As shown in the mid-power back-off point N, because the third power amplifier 3309 is a Class C power amplifier, when the input RF signal is a small signal, the power of the input RF signal is small and insufficient to trigger the third power amplifier 3309 to operate. The third power amplifier 3309 is cut off, causing the peak branch to be open-circuited. The straight-through path of the coupler 3307 (i.e., from the input end of the coupler 3307 to the straight-through end of the coupler 3307) is equivalent to a quarter-wavelength line, which can achieve a 90-degree phase shift on the RF signal at the center frequency. The straight-through path of the coupler 3307 acts as Figure 8 The first phase-shift network 3304 in the power amplifier circuit shown is omitted, but compared with the first phase-shift network 3304, the insertion loss of the carrier branch is reduced.

[0128] like Figure 7 As shown in the mid-power saturation point P, the carrier branch and the peak branch are power-combined at the junction CON. Since the peak branch is amplified in two stages by the gain of the second power amplifier 3302 and the third power amplifier 3309, Figure 15 The absolute value of the coupling coefficient of the coupler 3307 in Figure 8 The absolute value of the coupling coefficient of the coupler 3307 is larger, which is beneficial to reducing the insertion loss from the output end of the first power amplifier 3301 to the through end of the coupler 3307 and improving the efficiency of the power amplifier circuit 33.

[0129] Figure 15 In the power amplifier circuit 33 shown, although the carrier branch and the peak branch are phase-aligned at the combining point CON at the center frequency of the RF signal, at other frequencies, a certain phase difference will occur between the carrier branch and the peak branch, resulting in a reduction in the output power of the power amplifier circuit 33.

[0130] For this reason, Figure 16 As shown, the embodiment of the present application provides another power amplifier circuit. Figure 15 Based on the power amplifier circuit 33 shown, the power amplifier circuit 33 also includes a fourth phase shift network 3310. The first end of the fourth phase shift network 3310 is electrically connected to the junction point CON, and the second end of the fourth phase shift network 3310 is grounded. That is, the fourth phase shift network 3310 is connected in parallel with the carrier branch and the peak branch. The fourth phase shift network 3310 is a frequency element. For example, the fourth phase shift network 3310 can be the quarter-wavelength line mentioned above. It should be noted that the fourth phase shift network 3310 can also be applied to Figure 8 、 Figure 9 、 Figure 10 、 Figure 13 、 Figure 14In the power amplifier circuit 33.

[0131] At the center frequency, the fourth phase-shifting network 3310 is grounded, which is equivalent to 0 ohms. After the 90-degree phase shift of the fourth phase-shifting network 3310, the equivalent impedance of the fourth phase-shifting network 3310 is infinite, which is equivalent to an open circuit and does not affect the equivalent impedance Ropt / 2 of the junction CON. At frequencies other than the center frequency, the fourth phase-shifting network 3310 can achieve phase shifts at other angles. The equivalent impedance of the fourth phase-shifting network 3310 is not ideally infinite. The farther the frequency of the RF signal is from the center frequency, the lower the equivalent impedance of the fourth phase-shifting network 3310. Since the fourth phase-shifting network 3310 is connected in parallel with the two branches, the equivalent impedance of the junction CON will be reduced during power fallback, thereby increasing the output power of the power amplifier circuit 33, reducing the power loss caused by the phase misalignment of the junction CON, and increasing the efficiency of the power amplifier circuit 33. Therefore, the bandwidth of the RF signal applicable to the power amplifier circuit 33 is larger. Among them, Figure 17 The middle dashed line is the equivalent impedance at each frequency when the characteristic impedance of the fourth phase shift network 3310 is 25 ohms. Figure 17 The solid line in the middle represents the equivalent impedance at each frequency when the characteristic impedance of the fourth phase-shifting network 3310 is 50 ohms. It can be seen that as the characteristic impedance of the fourth phase-shifting network 3310 increases, the equivalent impedance at the center frequency increases, and the closer it approaches infinity, the smaller the impact on the equivalent impedance Ropt / 2 at the combining point CON.

[0132] It should be noted that the fourth phase-shifting network 3310 is not connected in series between the second power amplifier 3302 and the combining point CON. The reason is that at the power back-off point, the second power amplifier 3302 has not yet started working, and the output end of the second power amplifier 3302 is open. Such a series connection will change the output end of the second power amplifier 3302 from an open circuit to a short circuit due to impedance transformation, change the equivalent impedance of the combining point CON, and then change the load impedance of the first power amplifier 3301, thereby reducing the efficiency of power amplification.

[0133] like Figure 7 As shown at the mid-power back-off point N, at the center frequency of the RF signal, the coupler 3307 performs an impedance transformation, transforming the equivalent impedance Ropt / 2 of the junction CON into the load impedance 2Ropt of the first power amplifier 3301. The farther the frequency of the RF signal is from the center frequency, the lower the load impedance of the first power amplifier 3301. At this time, the fourth phase-shifting network 3310 lowers the equivalent impedance of the junction CON, which also slows down the degree of reduction in the load impedance of the first power amplifier 3301 and improves the efficiency of the first power amplifier 3301. For example, Figure 18As shown, it is the ratio of the load impedance of the first power amplifier 3301 to the load impedance Ropt at saturation (i.e., the impedance ratio), wherein the dotted line corresponds to the case where the junction CON is not electrically connected to the fourth phase-shift network 3310, and the solid line corresponds to the case where the junction CON is electrically connected to the fourth phase-shift network 3310.

[0134] It should be noted that if the second phase shift network 336 is optimized so that the peak branch and the carrier branch within the frequency band achieve a constant phase difference of 0 or 360 degrees at the combining point CON, then the combining point CON no longer needs to be electrically connected to the fourth phase shift network 3310.

[0135] The power amplifier circuit and electronic device provided by the embodiment of the present application are as follows: the input end of the first power amplifier is used to input the radio frequency signal, the output end of the first power amplifier is electrically connected to the input end of the coupler, the straight-through end of the coupler and the first end of the matching network are electrically connected to the junction point, the second end of the matching network is used to output the radio frequency signal after power amplification, and the isolation end of the coupler is grounded through the matching load; the coupling end of the coupler is electrically connected to the junction point through the second phase-shifting network and the second power amplifier; when the first power amplifier performs power back-off, the second power amplifier is open-circuited. The branch where the first power amplifier is located is called the carrier branch, and the branch where the second power amplifier is located is called the peak branch. When the power is backed off, the peak branch is open-circuited, and the straight-through path of the coupler is equivalent to a quarter-wavelength line, which can achieve a 90-degree phase shift for the radio frequency signal at the center frequency, which is equivalent to a phase-shifting network, plays the role of impedance transformation, and can change the load impedance of the first power amplifier. Since the phase-shifting network electrically connected to the output end of the first power amplifier is cancelled, the insertion loss of the carrier branch is reduced.

[0136] The above description is merely a specific embodiment of the present application, but the scope of protection of the present application is not limited thereto. Any changes or substitutions that can be easily conceived by a person skilled in the art within the technical scope disclosed in this application should be included in the scope of protection of this application. Therefore, the scope of protection of this application should be based on the scope of protection of the claims.

Claims

1. A power amplifier circuit, characterized in that: include: A first power amplifier, a second power amplifier, a third power amplifier, a matching network, a second phase-shifting network, a coupler, a matching load, a third phase-shifting network, and a fourth phase-shifting network; The input end of the first power amplifier is used to input a radio frequency signal, the output end of the first power amplifier is electrically connected to the input end of the coupler, the through-end of the coupler and the first end of the matching network are directly electrically connected to a junction point, a carrier branch is formed between the through-end of the coupler and the junction point, the second end of the matching network is used to output the power-amplified radio frequency signal, and the isolation end of the coupler is grounded through the matching load; a peak branch is formed between the coupled end of the coupler and the junction point, and the third power amplifier, the second phase-shifting network, the second power amplifier, and the third phase-shifting network are connected in series in the peak branch; a first end of the fourth phase-shifting network is electrically connected to the junction point, and a second end of the fourth phase-shifting network is grounded. At the center frequency of the radio frequency signal, the fourth phase-shifting network is configured to generate a 90-degree phase shift on the signal input to the fourth phase-shifting network; the difference between the absolute value of the coupling coefficient of the coupler and the sum of the gain of the second power amplifier and the gain of the third power amplifier is within a preset range; when the first power amplifier performs power back-off, the second power amplifier is open-circuited; At the center frequency of the radio frequency signal, the second phase shift network is configured to produce a 180-degree phase shift on the signal input to the second phase shift network, the third phase shift network is configured to produce a 20-degree phase shift on the signal input to the third phase shift network, the second power amplifier is configured to amplify the gain of the signal input to the second power amplifier and perform a 70-degree phase shift on the signal input to the second power amplifier, the phase difference of the signal output from the coupling end of the coupler relative to the signal output from the through end is 90 degrees, and the signal output from the carrier branch and the signal output from the peak branch are combined in phase at the combining point.

2. The power amplifier circuit according to claim 1, characterized in that: The first power amplifier is a class AB power amplifier, the second power amplifier is a class B power amplifier or a class C power amplifier, and the third power amplifier is a class C power amplifier.

3. The power amplifier circuit according to claim 1, characterized in that: The first power amplifier is a class AB power amplifier, and the second power amplifier is a class C power amplifier.

4. The power amplifier circuit according to claim 1, wherein: A difference between an absolute value of a coupling coefficient of the coupler and a gain of the second power amplifier is within a preset range.

5. The power amplifier circuit according to claim 1, characterized in that: The fourth phase shift network is a quarter-wavelength line.

6. An electronic device, characterized in that: The invention comprises an antenna and a power amplifier circuit according to any one of claims 1 to 5, wherein the power amplifier circuit is used to output a power-amplified radio frequency signal to the antenna.

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