A two-stage voltage converter with self-balancing flying capacitor voltage

Abstract

The present invention discloses a two-stage voltage converter with self-balancing flying capacitor voltage, which belongs to the field of voltage converters. The two-stage voltage converter includes: a PWM controller, a synchronous square wave generator, a current detection module, a first-stage conversion circuit and a second-stage conversion circuit; the first-stage conversion circuit operates at a fixed resonant frequency and a fixed duty cycle, and the resonant inductor L ZVS The stored energy is only related to the input voltage; the output of the first-stage conversion circuit serves as the input of the second-stage conversion circuit; the output of the second-stage conversion circuit is connected to the load; the second-stage conversion circuit achieves self-balancing of the flying capacitor by shorting the phase node and introducing a balancing inductor. ZVS When stable, the stored energy is independent of the load current. All power devices in the first stage can achieve soft switching within a certain load current range. The self-balancing control part of the flying capacitor voltage in the second-stage conversion circuit only needs to adjust the output voltage. The loop is simple, reducing the system design complexity and total cost.

Description

Technical Field

[0001] The present invention belongs to the field of voltage converters, and more particularly, relates to a two-stage voltage converter with self-balancing flying capacitor voltage. Background Art

[0002] With the development of technologies like big data and cloud computing, data centers are experiencing a dramatic increase in power demand. Traditional 12V power supply systems are struggling to meet these increasing performance requirements, prompting a shift from 12V to 48V. To ensure compatibility between legacy 12V infrastructure and the new 48V power supply system, a 48V-12V DC-DC converter is required.

[0003] For DC-DC converters used in data centers, high power density, high efficiency, and stable output voltage are core requirements. Common 48V-12V DC-DC converters can be configured using either a single-stage or two-stage structure. While a single-stage converter minimizes the number of components and achieves the highest power density, a two-stage converter allows one stage to operate at a significantly higher frequency than the other, thereby increasing the system's control bandwidth and improving transient performance. Furthermore, in a two-stage 4:1 converter, each switch operates under a 50% duty cycle control signal, minimizing inductor ripple voltage and further reducing inductor size compared to a single-stage structure.

[0004] Currently, there are three common topologies for DC-DC conversion: buck-based step-down converters, transformer-based step-down converters, and switched-capacitor-based step-down converters. Common buck-based step-down converters include the traditional buck converter and the flying capacitor converter. Traditional buck converters use inductors for energy transfer and have mature control schemes, but they suffer from large switching stresses on components and large inductor size, limiting power density and efficiency. Flying capacitor converters (FCs) significantly reduce switching stress on each component, increase the equivalent switching frequency at the inductor, and significantly reduce inductor size, resulting in significant improvements in power density and efficiency. However, parasitics, component mismatch, and gate drive delays can lead to voltage imbalances on the flying capacitors, causing the voltage on a particular switch to exceed its rated value, impacting system stability. A common transformer-based step-down converter is the LLC converter. LLC converters achieve regulated output voltage by controlling the switching frequency and offer advantages such as full isolation and high frequency. However, the presence of transformers and large inductors prevents the system from achieving a balance between power density and efficiency. Common switched-capacitor-based step-down converter circuits include switched-capacitor circuits and resonant switched-capacitor circuits. Switched-capacitor circuits use capacitors for energy transfer, significantly improving the system's power density compared to buck circuits, but with charge redistribution losses. Resonant switched-capacitor circuits incorporate a small resonant inductor into the switched-capacitor circuit, avoiding charge redistribution while enabling soft-switching operation of each power device and reducing switching losses. A common two-stage converter uses a resonant switched-capacitor circuit as the high-voltage, low-current first stage and a flying capacitor conversion circuit as the low-voltage, high-current second stage. For such converters, soft switching technology and flying-capacitor voltage balancing are crucial.

[0005] In summary, the energy stored in the resonant inductor in the existing two-stage voltage converter is related to the load current. Changes in the load current will affect the implementation of soft switching, thereby increasing system power consumption. Soft switching may not be possible under low currents. To achieve flying capacitor voltage balance, the voltage / current of each flying capacitor needs to be sampled and detected, which makes the system design complex, the total cost high, the power consumption high, and affects the system stability. Summary of the Invention

[0006] In view of the defects of the related art, the purpose of the present invention is to provide a two-stage voltage converter with self-balancing flying capacitor voltage, aiming to solve the problems of possible failure of resonant operation under small load current and imbalance of flying capacitor voltage.

[0007] To achieve the above object, the present invention provides a two-stage voltage converter with self-balancing flying capacitor voltage, comprising: a PWM controller, a synchronous square wave generator, a current detection module, a first-stage conversion circuit, and a second-stage conversion circuit;

[0008] The first stage conversion circuit is a resonant switched capacitor conversion circuit, including a first three-level flying capacitor conversion circuit, a first stage output capacitor C OUT1 , resonant inductor L ZVS and DC blocking capacitor C DC ; The first terminal input voltage V of the first three-level flying capacitor conversion circuit IN , the second end is grounded; the resonant inductor L ZVS One end of the capacitor C DC connected to the first three-level flying capacitor conversion circuit, and the other end is grounded; the output end of the first three-level flying capacitor conversion circuit is connected to the first-stage output capacitor C OUT1 rear grounding;

[0009] The output of the first stage conversion circuit is used as the input of the second stage conversion circuit, which includes a symmetrically arranged second three-level flying capacitor conversion circuit and a third three-level flying capacitor conversion circuit, and a balancing inductor L bal And the second stage output inductor L out The first end of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit is connected to the output end of the first three-level flying capacitor conversion circuit, and the second end of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit is grounded; the balancing inductor L bal connected between the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit; the output ends of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit are short-circuited and connected to the second-stage output inductor L out The first end of the second stage output inductor L out The second end of is connected to the output end;

[0010] The PWM controller is connected to the synchronous square wave generator and the current detection module, and is connected to the switching device of the second-stage conversion circuit; the synchronous square wave generator is connected to the switching device of the first-stage conversion circuit;

[0011] The PWM controller is used to obtain the second-stage output inductance L according to the current detection module. out The output current and the feedback voltage at the output end of the second-stage conversion circuit generate a synchronization signal sync and a second-stage control signal with adjustable duty cycle; the second-stage control signal is used to control the switching device of the second-stage conversion circuit;

[0012] The synchronous square wave generator is used to generate a first-stage control signal with a fixed duty cycle of 50% using a synchronous signal sync, so as to control the switch device of the first-stage conversion circuit.

[0013] Optionally, the frequency of the switch control signal of the first-stage conversion circuit is twice the frequency of the switch control signal of the second-stage conversion circuit.

[0014] Optionally, the first three-level flying capacitor conversion circuit includes a ninth switch Q 11 , the tenth switch Q 21 、The eleventh switch Q 31 and the twelfth switch Q 41 , and the flying capacitor C res ;

[0015] Flying capacitor C res The two ends of the ninth switch Q are connected 11 The source of the twelfth switch Q 41 The drain, flying capacitor C res and the DC blocking capacitor C DC and resonant inductor L ZVS Connect in series and then ground; the ninth switch Q 11 The drain is connected to the input voltage V IN ; The tenth switch Q 21 The source of each is connected to the first stage output capacitor C OUT1 The first end of the twelfth switch Q 41 The drain of the flying capacitor C res and the DC blocking capacitor C DC Between, the source is grounded; the first stage output capacitor C OUT1 The second end is grounded.

[0016] Optionally, the second three-level flying capacitor conversion circuit includes a first switch Q1, a second switch Q2, a third switch Q3 and a fourth switch Q4 connected in sequence, and a first flying capacitor C fly1 ;

[0017] The drain of the first switch Q1 serves as the first end of the second three-level flying capacitor conversion circuit, and the source of the fourth switch Q4 is grounded;

[0018] The first flying capacitor C fly1 The two ends of the transistor are respectively connected to the source of the first switch Q1 and the drain of the fourth switch Q4.

[0019] Optionally, the third three-level flying capacitor conversion circuit includes a fifth switch Q5, a sixth switch Q6, a seventh switch Q7 and an eighth switch Q8 connected in sequence, and a second flying capacitor C fly2 ;

[0020] The drain of the fifth switch Q5 serves as the first end of the third three-level flying capacitor conversion circuit, and the source of the eighth switch Q8 is grounded;

[0021] The second flying capacitor C fly2The two ends of Q5 are connected to the source of the fifth switch Q5 and the drain of the eighth switch Q8 respectively.

[0022] Optional, second stage output inductor L out The second end is connected to the second stage output capacitor C out And the load resistor R0; the second stage output capacitor C out The second end of the output inductor L out and the first end of the load resistor R0, the output inductor L out And the second end of the load resistor R0 is grounded.

[0023] Optionally, the switching device is a MOS tube, and the PWM controller and the synchronous square wave generator are respectively connected to the gate of the MOS tube.

[0024] Optionally, the PWM controller generates a second-level control signal and in and The rising edge and and The falling edge enables the synchronization signal sync; among which, the rising edge enables the first level control signal when When it rises to high level and the falling edge enables the first level control signal Reduced to low level.

[0025] Compared with the prior art, the above technical solutions conceived by the present invention can achieve the following beneficial effects:

[0026] 1. The present invention provides a two-stage voltage converter with self-balancing flying capacitor voltage, which optimizes the topology structure and adopts a resonant inductor L ZVS When stable, the voltage across the terminals is ±V in / 4, the stored energy is independent of the load current, and all power devices in the first-stage resonant switched capacitor conversion circuit can achieve soft switching within a certain load current range. The topology of the two symmetrical three-level flying capacitor converters in the second-stage conversion circuit is improved, and a flying capacitor voltage self-balancing control technique is proposed. This achieves self-balancing of the flying capacitor voltage by shorting the phase nodes and introducing balancing inductors. Without the addition of additional control circuitry, the voltage across the second-stage flying capacitor is self-balanced, simplifying the control circuit design and improving system stability.

[0027] 2. The present invention provides a two-stage voltage converter with self-balancing flying capacitor voltage. A resonant inductor is connected in parallel across a switching device, and a capacitor is added to isolate the resonant inductor from the DC signal. This ensures that the energy stored in the resonant inductor in the first-stage resonant switched capacitor conversion circuit is only related to the input voltage and has nothing to do with the output current. Even with a small resonant inductor, all power devices can be turned on at zero voltage over a wide load current range. This resolves the contradiction between the resonant inductor size and the load current range in traditional resonant switched capacitor conversion circuits, effectively improving the system's power density. The present invention's self-balancing flying capacitor voltage control loop eliminates the need for sampling and controlling the flying capacitor voltage or phase current, effectively improving the system's power density.

[0028] 3. The present invention provides a two-stage voltage converter with self-balancing flying capacitor voltage. The first stage adopts resonant operation to reduce switching losses. The second stage realizes self-balancing voltage by forcing two flying capacitors to be connected to the load inductor and the ground for two equal periods of time in one switching cycle, thereby improving the inductor current ripple and reducing the conduction loss, while eliminating capacitor redistribution loss.

[0029] 4. The present invention provides a two-stage voltage converter with self-balancing flying capacitor voltage. The energy stored in the first-stage resonant inductor is independent of the load current. The switching tube can also achieve resonant operation under small load current, which widens the lower limit of the load current; the second stage adopts a two-phase structure, which widens the upper limit of the load current. BRIEF DESCRIPTION OF THE DRAWINGS

[0030] Figure 1 It is a topological diagram of an existing two-stage voltage converter;

[0031] Figure 2 This is a timing waveform diagram of control signals of an existing two-stage voltage converter under ideal conditions;

[0032] Figure 3 This is a structural schematic diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention;

[0033] Figure 4 This is a working waveform diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention;

[0034] Figure 5 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 1;

[0035] Figure 6 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 2;

[0036] Figure 71 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 3;

[0037] Figure 8 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 4;

[0038] Figure 9 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 5;

[0039] Figure 10 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 6;

[0040] Figure 11 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 7;

[0041] Figure 12 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 8;

[0042] Figure 13 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 9;

[0043] Figure 14 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 10;

[0044] Figure 15 1 is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 11;

[0045] Figure 16 This is an equivalent circuit diagram of a two-stage voltage converter with self-balancing flying capacitor voltage provided by the present invention in mode 12. DETAILED DESCRIPTION

[0046] In order to make the objectives, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely for the purpose of explaining the present invention and are not intended to limit the present invention. In addition, the technical features involved in the various embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.

[0047] like Figure 1 and Figure 2 As shown in FIG, there is a topological structure diagram of an existing two-stage voltage converter and a timing waveform diagram of a control signal under ideal conditions.

[0048] In order to achieve soft switching technology, the frequency of all control signals is the resonant frequency f of the first-stage flying capacitor and the output inductor. SW , as shown below:

[0049]

[0050] During a switching cycle, the flying capacitor resonantly charges in half a cycle and discharges in half a cycle, eliminating charge redistribution losses. Furthermore, because the switching frequency is equal to the resonant frequency, the inductor current waveform is a rectified sine wave, reaching zero at the phase transition, thus achieving zero-current switching.

[0051] In order to realize the flying capacitor voltage balancing technology, the circuit adopts valley current mode control, which uses the real-time detected output current and feedback voltage to generate the first-level control signal with a duty cycle of 50% in real time. And the second-level control signal with adjustable duty cycle and fixed phase The flying capacitor voltage V Cfly1 V mid / 2 is small, the flying capacitor voltage V Cfly2 V mid / 2 is taken as an example for analysis: When the flying capacitor voltage V Cfly1 V mid / 2 is small, the flying capacitor voltage V Cfly2 V mid / 2 is large, the control signal and The duty cycle increases, and The duty cycle is reduced, and the flying capacitor C fly1 Charge more, the flying capacitor C fly2 Discharge more charge, the flying capacitor voltage is restored. When stable, the voltage of the two flying capacitors is equal to V mid / 2.

[0052] The above-mentioned prior art solutions have the following defects:

[0053] 1. The energy stored in the inductor in the first-stage resonant switched capacitor conversion circuit is related to the load current. Changes in the load current will affect the implementation of soft switching.

[0054] 2. To achieve flying capacitor voltage balancing, existing solutions require detecting the output inductor current of each flying capacitor branch, which greatly increases the total system cost. At the same time, the existence of multiple loops poses a challenge to system stability.

[0055] Based on the above technical deficiencies, an embodiment of the present invention proposes a two-stage voltage converter with self-balancing flying capacitor voltage. The following terms are explained and defined in the two-stage voltage converter with self-balancing flying capacitor voltage provided in this embodiment:

[0056] DC-DC converter: A power conversion circuit that converts direct current (DC) power to a DC power supply of a different voltage.

[0057] Buck circuit: It obtains pulsed energy from the input DC power supply by controlling the on-off state of the switch tube, and uses the LC filter to smooth the pulsed energy, thereby obtaining an output voltage with less ripple at the output end.

[0058] LLC converter: A resonant circuit consisting of two inductors and a capacitor that controls the switching frequency to achieve a constant output voltage.

[0059] Switched capacitor circuit: A fixed current source controlled by a switch charges and discharges the capacitor, changing the voltage level of the capacitor through charging and discharging.

[0060] Soft switching technology: V of MOS tube during normal on / off DS with I DS There will be overlap, and there will be conduction / turn-off losses. By adjusting the V DS / I DS It can reduce the turn-on / off loss. These two strategies correspond to zero voltage switching technology and zero current switching technology respectively.

[0061] PWM: pulse width modulation, a modulation method that keeps the modulation signal frequency constant and changes the modulation signal duty cycle according to the error between the output voltage and the set voltage, so that the output voltage is close to the expected value.

[0062] The following describes the contents involved in the embodiment of the present invention in conjunction with a preferred embodiment.

[0063] The present invention proposes a two-stage voltage converter with self-balancing flying capacitor voltage, with an input voltage range of 48V-60V and a constant output voltage of 12V. The first stage adopts a resonant switched capacitor conversion circuit, and the second stage adopts a two-phase three-level flying capacitor conversion circuit. Figure 3 As shown, a two-stage voltage converter with self-balancing flying capacitor voltage includes: a PWM controller, a synchronous square wave generator, a current detection module, a first-stage conversion circuit and a second-stage conversion circuit;

[0064] The first stage conversion circuit is a resonant switched capacitor conversion circuit, including a first three-level flying capacitor conversion circuit, a first stage output capacitor C OUT1 , resonant inductor LZVS and DC blocking capacitor C DC ; The first terminal input voltage V of the first three-level flying capacitor conversion circuit IN , the second end is grounded; the resonant inductor L ZVS One end of the capacitor C DC connected to the first three-level flying capacitor conversion circuit, and the other end is grounded; the output end of the first three-level flying capacitor conversion circuit is connected to the first-stage output capacitor C OUT1 rear grounding;

[0065] The output of the first stage conversion circuit is used as the input of the second stage conversion circuit, which includes a symmetrically arranged second three-level flying capacitor conversion circuit and a third three-level flying capacitor conversion circuit, and a balancing inductor L bal And the second stage output inductor L out The first end of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit is connected to the output end of the first three-level flying capacitor conversion circuit, and the second end of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit is grounded; the balancing inductor L bal connected between the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit; the output ends of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit are short-circuited and then pass through the second-stage output inductor L out Connect the output end; the output nodes of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit are short-circuited, and the balancing inductor L bal Providing a charging and discharging path for the flying capacitor, a stable voltage of the second-stage conversion circuit, and a periodic inductor current to make the flying capacitor self-balanced;

[0066] The PWM controller is connected to the synchronous square wave generator and the current detection module, and is connected to the switching device of the second-stage conversion circuit; the synchronous square wave generator is connected to the switching device of the first-stage conversion circuit;

[0067] The PWM controller is used to obtain the second-stage output inductance L according to the current detection module. out The output current and the feedback voltage at the output end of the second-stage conversion circuit generate a synchronization signal sync and a second-stage control signal with adjustable duty cycle; the second-stage control signal is used to control the switching device of the second-stage conversion circuit;

[0068] The synchronous square wave generator is used to generate a first-stage control signal with a fixed duty cycle of 50% using a synchronous signal sync, so as to control the switch device of the first-stage conversion circuit.

[0069] The resonant inductor L ZVSIt is used to store energy and control the zero voltage turn-on of the power switch. The stored energy is only related to the input voltage. The first-stage conversion circuit operates at a fixed resonant frequency and a fixed duty cycle. The resonant inductor L ZVS The stored energy is only related to the input voltage. The output nodes of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit are short-circuited, and the balancing inductor L bal Providing a charging and discharging path for the flying capacitor, stabilizing the voltage and periodic inductive current to enable the flying capacitor to achieve self-balancing.

[0070] Continue to refer Figure 3 Optionally, the first three-level flying capacitor conversion circuit includes a ninth switch Q connected in sequence 11 , the tenth switch Q 21 、The eleventh switch Q 31 and the twelfth switch Q 41 , and the flying capacitor C res ;

[0071] Flying capacitor C res The two ends of the ninth switch Q are connected 11 The source of the twelfth switch Q 41 The drain, flying capacitor C res and the DC blocking capacitor C DC and resonant inductor L ZVS Connect in series and then ground; the ninth switch Q 11 The drain is connected to the input voltage V IN ; The tenth switch Q 21 The source of each is connected to the first stage output capacitor C OUT1 The first end of the twelfth switch Q 41 The drain of the flying capacitor C res and the DC blocking capacitor C DC Between, the source is grounded; the first stage output capacitor C OUT1 The second end is grounded.

[0072] Optionally, the second three-level flying capacitor conversion circuit includes a first switch Q1, a second switch Q2, a third switch Q3 and a fourth switch Q4 connected in sequence, and a first flying capacitor C fly1 ;

[0073] The drain of the first switch Q1 serves as the first end of the second three-level flying capacitor conversion circuit, and the source of the fourth switch Q4 is grounded;

[0074] The first flying capacitor C fly1 The two ends of the transistor are respectively connected to the source of the first switch Q1 and the drain of the fourth switch Q4.

[0075] Optionally, the third three-level flying capacitor conversion circuit includes a fifth switch Q5, a sixth switch Q6, a seventh switch Q7 and an eighth switch Q8 connected in sequence, and a second flying capacitor C fly2 ;

[0076] The drain of the fifth switch Q5 serves as the first end of the third three-level flying capacitor conversion circuit, and the source of the eighth switch Q8 is grounded;

[0077] The second flying capacitor C fly2 The two ends of Q5 are connected to the source of the fifth switch Q5 and the drain of the eighth switch Q8 respectively.

[0078] Optional, second stage output inductor L out The second end is connected to the second stage output capacitor C out And the load resistor R0; the second stage output capacitor C out The second end of the output inductor L out and the first end of the load resistor R0, the output inductor L out And the second end of the load resistor R0 is grounded.

[0079] Wherein, each of the above-mentioned switching devices is a MOS tube, and the PWM controller and the synchronous square wave generator are respectively connected to the gate of the MOS tube.

[0080] Optionally, the PWM controller generates a second-level control signal and in and The rising edge and and The falling edge enables the synchronization signal sync; among which, the rising edge enables the first level control signal when When it rises to high level and the falling edge enables the first level control signal Reduced to low level.

[0081] The working principle of the two-stage voltage converter with self-balancing flying capacitor voltage proposed in the embodiment of the present invention is as follows: Figure 4 The working waveform diagram shows that:

[0082] The switching frequency of the second-stage control signal is the rated switching frequency. Since the first-stage switch tube always switches at zero voltage, the switching loss can be ignored. Therefore, the switching frequency of the first-stage control signal can be greater than the rated switching frequency to improve the system bandwidth. Here, the switching frequency of the first-stage control signal is twice the switching frequency of the second-stage control signal. In practical applications, it is difficult for the input voltage to be stabilized at the rated 48V voltage. The following example assumes that the input voltage is higher than 48V. and The circuit working principle is explained by taking the duty cycle less than 50% as an example. The analysis of the circuit working principle when the input voltage is less than or equal to 48V is the same. In a basic cycle, t0-t 12 There are twelve working modes, and the equivalent circuit diagrams of each stage are as follows: Figures 5 to 16 As shown, the MOS tube marked in black indicates that the MOS tube is turned on, and the MOS tube marked in gray indicates that the MOS tube is turned off. Specifically including:

[0083] (1) Mode 1: t0-t1

[0084] The equivalent circuit diagram of this stage is as follows Figure 5 shown.

[0085] For the first stage circuit, Q 11 With Q 31 Zero voltage turn-on, Q 21 With Q 41 Keep the off state. Current i Cres Flow through Q 11 Give the switch capacitor C res Charging, flowing through Q 31 To the output capacitor C OUT1 Since Q 41 Shutdown, Q 41 The drain-source level has a size of V in / 2 voltage difference, the DC blocking capacitor C DC Under the effect of L ZVS There is a V at the upper and lower ends in / 4 positive voltage difference, which has nothing to do with the load current. LZVS Continuously increasing.

[0086] For the second stage circuit, after the dead time back In this stage, the flying capacitor C fly2 Charged from the input, connected to V through Q7 ph2 is the output inductor L out Power supply, C fly1 Connect to V through Q2 ph1 Towards the output inductor L out Discharge; inductor current i Lout Increase; balance inductance I bal The current on I Lbal,pk Start to lower.

[0087] (2) Mode 2: t1-t2

[0088] The equivalent circuit diagram of this stage is as follows Figure 6 shown.

[0089] For the first stage circuit, Q 11 With Q31 Shutdown, Q 21 With Q 41 Keep it in the off state. LZVS Q 11 Junction capacitance C OSS,11 With Q 31 Junction capacitance C OSS,31 Charge to C OSS *V in / 2, C OSS,21 and C OSS,41 Empty. Because at this time LZVS Greater than 0, Q 21 With Q 41 The body diode is turned on, which is the Q 21 With Q 41 Prepare for zero voltage turn-on.

[0090] For the second-stage circuit, the control signal at this stage does not change, and the circuit continues to operate as described in Mode 1.

[0091] (3) Mode 3: t2-t3

[0092] The equivalent circuit diagram of this stage is as follows Figure 7 shown.

[0093] For the first stage circuit, Q 21 With Q 41 Zero voltage turn-on, Q 11 With Q 31 Keep the switch in the off state. res By Q 21 Give the output capacitor C OUT1 Discharge. Since Q 41 conduction, Q 41 The drain-source voltage difference is 0, and the DC blocking capacitor C DC Under the effect of L ZVS There is a V at the upper and lower ends in / 4 negative voltage difference, which has nothing to do with the load current. LZVS Continuously decreasing.

[0094] For the second-stage circuit, the control signal at this stage does not change, and the circuit continues to operate as described in Mode 2.

[0095] (4) Mode 4: t3-t4

[0096] The equivalent circuit diagram of this stage is as follows Figure 8 shown.

[0097] For the first-stage circuit, the control signal at this stage does not change, and the circuit continues to operate as described in Mode 3.

[0098] For the second stage circuit, Q2 and Q5 are turned off at t=t3. The relevant driver stage enters the dead zone, the body diodes of Q3 and Q8 are turned on, and the inductor current i Lout Decrease.

[0099] (5) Mode 5: t4-t5

[0100] The equivalent circuit diagram of this stage is as follows Figure 9 shown.

[0101] For the first-stage circuit, the control signal at this stage does not change, and the circuit continues to operate as described in Mode 4.

[0102] For the second stage circuit, when t=t4, the dead time has passed. back Rising to a high level, Q3 and Q8 are turned on, and the inductor current is -V out / L out The slope decreases.

[0103] (6) Mode 6: t5-t6

[0104] The equivalent circuit diagram of this stage is as follows Figure 10 shown.

[0105] For the first stage circuit, Q 21 With Q 41 Shutdown, Q 11 With Q 31 Keep it in the off state. LZVS Q 21 Junction capacitance C OSS,21 With Q 41 Junction capacitance C OSS,41 Charge to C OSS *V in / 2, C OSS,11 and C OSS,31 Empty. Because at this time LZVS Less than 0, Q 11 With Q 31 The body diode is turned on, which is the Q 11 With Q 31 Prepare for zero voltage turn-on.

[0106] For the second stage circuit, Q4 and Q7 are turned off at t=t5. The relevant driver stage enters the dead zone, the body diodes of Q4 and Q7 are turned on, and the inductor current i Lout It continues to decrease and reaches the minimum value at t=t6.

[0107] (7) Mode 7: t6-t7

[0108] The equivalent circuit diagram of this stage is as follows Figure 11 shown.

[0109] For the first stage circuit, Q 11 With Q 31 Zero voltage turn-on, Q 21 With Q 41 Keep the off state. Current i Cres Flow through Q 11 Give the switch capacitor C res Charging, flowing through Q 31 To the output capacitor C OUT1 Since Q 41 Shutdown, Q 41 The drain-source level has a size of V in / 2 voltage difference, the DC blocking capacitor C DC Under the effect of L ZVS There is a V at the upper and lower ends in / 4 positive voltage difference, which has nothing to do with the load current. LZVS Continuously increasing.

[0110] For the second stage circuit, when t=t6, the dead time has passed. back Rising to a high level, Q1 and Q6 are turned on, and the flying capacitor C fly1 Charged from the input, connected to V through Q3 ph1 is the output inductor L out Power supply, C fly2 Connect to V through Q6 ph2 Towards the output inductor L out Discharge; Balanced inductance I bal The current on -I Lbal,pk Starts to increase; the inductor current i Lout Increase.

[0111] (8) Mode 8: t7-t8

[0112] The equivalent circuit diagram of this stage is as follows Figure 12 shown.

[0113] For the first stage circuit, Q 11 With Q 31 Shutdown, Q 21 With Q 41 Keep it in the off state. LZVS Q 11 Junction capacitance C OSS,11 With Q 31 Junction capacitance C OSS,31 Charge to C OSS *V in / 2, C OSS,21 and C OSS,41 Empty. Because at this time LZVSGreater than 0, Q 21 With Q 41 The body diode is turned on, which is the Q 21 With Q 41 Prepare for zero voltage turn-on.

[0114] For the second-stage circuit, the control signal at this stage does not change, and the circuit continues to operate according to the conditions described in Mode 7.

[0115] (9) Mode 9: t8-t9

[0116] The equivalent circuit diagram of this stage is as follows Figure 13 shown.

[0117] For the first stage circuit, Q 21 With Q 41 Zero voltage turn-on, Q 11 With Q 31 Keep the switch in the off state. res By Q 21 Give the output capacitor C OUT1 Discharge. Since Q 41 conduction, Q 41 The drain-source voltage difference is 0, and the DC blocking capacitor C DC Under the effect of L ZVS There is a V at the upper and lower ends in / 4 negative voltage difference, which has nothing to do with the load current. LZVS Continuously decreasing.

[0118] For the second-stage circuit, the control signal at this stage does not change, and the circuit continues to operate according to the conditions described in Mode 8.

[0119] (10) Mode 10: t9-t 10

[0120] The equivalent circuit diagram of this stage is as follows Figure 14 shown.

[0121] For the first-stage circuit, the control signal at this stage does not change, and the circuit continues to operate according to the conditions described in Mode 9.

[0122] For the second stage circuit, Q1 and Q6 are turned off at t=t9. The relevant driver stage enters the dead zone, the body diodes of Q4 and Q7 are turned on, and the inductor current i Lout Decrease.

[0123] (11) Mode 11: t 10 -t 11

[0124] The equivalent circuit diagram of this stage is as follows Figure 15 shown.

[0125] For the first-stage circuit, the control signal at this stage does not change, and the circuit continues to operate according to the situation described in Mode 10.

[0126] For the second stage circuit, t = t 10 Dead time back Rising to a high level, Q1 and Q6 are turned on, and the inductor current is -V out / L out The slope decreases.

[0127] (12) Mode 12: t 11 -t 12

[0128] The equivalent circuit diagram of this stage is as follows Figure 16 shown.

[0129] For the first stage circuit, Q 21 With Q 41 Shutdown, Q 11 With Q 31 Keep it in the off state. LZVS Q 21 Junction capacitance C OSS,21 With Q 41 Junction capacitance C OSS,41 Charge to C OSS *V in / 2, C OSS,11 and C OSS,31 Empty. Because at this time LZVS Less than 0, Q 11 With Q 31 The body diode is turned on, which is the Q 11 With Q 31 Prepare for zero voltage turn-on.

[0130] For the second stage circuit, t = t 11 hour drops to a low level, Q3 and Q8 are turned off, and the inductor current i Lout Continue to decrease, at t=t 12 Reduce to a minimum value.

[0131] In the above-mentioned two-stage voltage converter with self-balancing flying capacitor voltage, the topology is optimized and the resonant inductor L ZVS When stable, the voltage across the terminals is ±V in / 4, the stored energy has nothing to do with the load current. In the present invention, all power devices in the first-stage resonant switched capacitor conversion circuit can achieve soft switching within a certain load current range.

[0132] Furthermore, the topology of the two symmetrical three-level flying capacitor converters in the second-stage conversion circuit is improved, and a flying capacitor voltage self-balancing control technology is proposed. By short-circuiting the phase nodes and referencing balancing inductors, the flying capacitor voltage is self-balanced.

[0133] The short circuit of the phase node mainly affects the periods t0-t3 and t6-t9. fly1 The lower plate of the grounded, C fly2 The upper plate of the first stage is connected to the output, and during the period t6-t9, C fly1 The upper plate is connected to the first stage output, C fly2 The lower plate of the inductor is grounded. According to the topology, the inductor voltage is equal to the difference between the voltage of the flying capacitor connected to ground and the output voltage. Since the inductor current changes periodically during stability, the two flying capacitors will self-equalize the voltage during the switching process. V Cfly1 With V Cfly2 The voltage deviation is defined as follows:

[0134]

[0135] When the phase node is short-circuited, the relationship between the flying capacitor voltage deviation and time is shown in the following formula:

[0136] ΔV(t)=ΔV init ·e -t / τ (3)

[0137] It can be seen that the voltage deviation decreases exponentially with the increase of time.

[0138] Where △V init is the initial flying capacitor mismatch voltage, ζ is the equilibrium time constant, and the expression is shown in formula (4):

[0139]

[0140] Among them, R cond represents the parasitic resistance from the bottom capacitor to the output terminal, f res , D adj The expressions are shown in formula (5) and (6) respectively:

[0141]

[0142]

[0143] Furthermore, a balancing inductor is introduced into the second-stage conversion circuit. The short-circuited phase node at a given timing can make the flying capacitor mismatch voltage exponentially decay. However, under extreme non-ideal conditions such as strong drive delay, the stability of the flying capacitor voltage cannot be guaranteed. Therefore, the present invention introduces a balancing inductor L bal .

[0144] For the balancing inductor L bal , its volt-second balance expression is shown in formula (7):

[0145]

[0146] Among them, DCR Lbal Indicates L bal The DC resistance, I bal,dc Indicates flow through L bal of DC current.

[0147] Under normal circumstances, T onφa ≈T onφb , V Cfly1 ≈V Cfly2 ≈V out1 / 2. If a strong drive delay occurs (T onφa <T onφb For example), V cfly1 >V out1 / 2,V cfly2 <V out1 / 2, at this time there is a positive current flowing through L bal Make C fly1 The upper part of the charge is transferred to C fly2 , causing the voltages of the two flying capacitors to change toward the equilibrium voltage.

[0148] For the flying capacitor C fly1 , the charging charge and discharging charge expressions in a basic switching cycle are shown in equations (8) and (9):

[0149]

[0150] According to the law of charge conservation, the balanced inductor DC current I can be obtained as shown in formula (10): bal,dc :

[0151]

[0152] From Equation (10), we can see that even with strong driving delay, I bal,dc < out , you can choose a small size balanced inductor L bal , which will not have much impact on the system power density.

[0153] ​The topology of the first-stage switching converter in the existing scheme is improved (a resonant inductor is connected in parallel across the switching device, and a capacitor is added to isolate the resonant inductor from the DC signal). This makes the energy stored in the resonant inductor dependent on the input voltage and independent of the output current. Within a given load current range, all first-stage power devices can achieve soft switching. For the second-stage two-phase three-level flying capacitor converter in the existing scheme, a flying capacitor voltage self-balancing control technology is proposed. Under this topology and control scheme, each flying capacitor can achieve voltage self-balancing, ensuring system stability.

[0154] The PWM controller discretizes the sampled feedback voltage and output current, compares the discretized feedback voltage with the reference voltage to generate an error signal, and then generates a digital compensation reference through the compensation network. Finally, the digital pulse width modulator and the driver stage compare it with the discretized output current and slope compensation to obtain the second-level control signal. and in and Rising edge and and The synchronous signal sync is enabled by the falling edge, and the first-level control signal is enabled when the rising edge is valid. When it rises to high level and the falling edge enables the first level control signal Reduced to low level.

[0155] The embodiment of the present invention optimizes the topology structure and adopts the resonant inductor L ZVS When stable, the voltage across the terminals is ±V in / 4, the stored energy is independent of the load current. All power devices in the first-stage resonant switched capacitor conversion circuit can achieve soft switching within a certain load current range, reducing system power consumption while widening the lower limit of the load current range. The topology of the two symmetrical three-level flying capacitor converters in the second-stage conversion circuit is improved, and a flying capacitor voltage self-balancing control technology is proposed. Under the control timing proposed in this patent, the flying capacitor voltage is self-balanced by shorting the output phase node and referencing a balancing inductor. The control loop for the self-balancing flying capacitor voltage is simple, reducing system design complexity and overall cost.

[0156] It will be easily understood by those skilled in the art that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A two-stage voltage converter with self-balancing flying capacitor voltage, characterized in that: include: PWM controller, synchronous square wave generator, current detection module, first stage conversion circuit and second stage conversion circuit; The first stage conversion circuit is a resonant switched capacitor conversion circuit, including a first three-level flying capacitor conversion circuit, a first stage output capacitor C OUT1 , resonant inductor L ZVS and DC blocking capacitor C DC ; The first terminal input voltage V of the first three-level flying capacitor conversion circuit IN , the second end is grounded; the resonant inductor L ZVS One end of the capacitor C DC connected to the first three-level flying capacitor conversion circuit, and the other end is grounded; the output end of the first three-level flying capacitor conversion circuit is connected to the first-stage output capacitor C OUT1 rear grounding; The output of the first stage conversion circuit is used as the input of the second stage conversion circuit, which includes a symmetrically arranged second three-level flying capacitor conversion circuit and a third three-level flying capacitor conversion circuit, and a balancing inductor L bal And the second stage output inductor L out The first end of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit is connected to the output end of the first three-level flying capacitor conversion circuit, and the second end of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit is grounded; the balancing inductor L bal connected between the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit; the output ends of the second three-level flying capacitor conversion circuit and the third three-level flying capacitor conversion circuit are short-circuited and then pass through the second-stage output inductor L out Connect the output terminal; The PWM controller is connected to the synchronous square wave generator and the current detection module, and is connected to the switching device of the second-stage conversion circuit; the synchronous square wave generator is connected to the switching device of the first-stage conversion circuit; The PWM controller is used to obtain the second-stage output inductance L according to the current detection module. out The output current and the feedback voltage at the output end of the second-stage conversion circuit generate a synchronization signal sync and a second-stage control signal with adjustable duty cycle; the second-stage control signal is used to control the switching device of the second-stage conversion circuit; The synchronous square wave generator is used to generate a first-stage control signal with a fixed duty cycle of 50% using a synchronous signal sync, so as to control the switching device of the first-stage conversion circuit.

2. The voltage converter according to claim 1, wherein: The frequency of the switch control signal of the first-stage conversion circuit is twice the frequency of the switch control signal of the second-stage conversion circuit.

3. The voltage converter according to claim 1, wherein: The first three-level flying capacitor conversion circuit includes a ninth switch Q connected in sequence 11 , the tenth switch Q 21 、The eleventh switch Q 31 and the twelfth switch Q 41 , and the flying capacitor C res ; Flying capacitor C res The two ends of the ninth switch Q are connected 11 The source of the twelfth switch Q 41 The drain, flying capacitor C res and the DC blocking capacitor C DC and resonant inductor L ZVS Connect in series and then ground; the ninth switch Q 11 The drain is connected to the input voltage V IN ; The tenth switch Q 21 The source of the first stage output capacitor C OUT1 The first end of the twelfth switch Q 41 The drain of the flying capacitor C res and the DC blocking capacitor C DC Between, the source is grounded; the first stage output capacitor C OUT1 The second end is grounded.

4. The voltage converter according to claim 1, wherein: The second three-level flying capacitor conversion circuit includes a first switch Q1, a second switch Q2, a third switch Q3 and a fourth switch Q4 connected in sequence, and a first flying capacitor C fly1 ; The drain of the first switch Q1 serves as the first end of the second three-level flying capacitor conversion circuit, and the source of the fourth switch Q4 is grounded; The first flying capacitor C fly1 The two ends of the transistor are respectively connected to the source of the first switch Q1 and the drain of the fourth switch Q4.

5. The voltage converter according to claim 4, wherein: The third three-level flying capacitor conversion circuit includes a fifth switch Q5, a sixth switch Q6, a seventh switch Q7 and an eighth switch Q8 connected in sequence, and a second flying capacitor C fly2 ; The drain of the fifth switch Q5 serves as the first end of the third three-level flying capacitor conversion circuit, and the source of the eighth switch Q8 is grounded; The second flying capacitor C fly2 The two ends of Q5 are connected to the source of the fifth switch Q5 and the drain of the eighth switch Q8 respectively.

6. The voltage converter according to claim 5, wherein: The second stage output inductor L out The second end is connected to the second stage output capacitor C out and the first end of the load resistor R0; the output capacitor C out And the second end of the load resistor R0 is grounded.

7. The voltage converter according to claim 1, wherein: The switching device is a MOS tube, and the PWM controller and the synchronous square wave generator are respectively connected to the gate of the MOS tube.

8. The voltage converter according to claim 1, wherein: The PWM controller generates a second-level control signal , and in and The rising edge of and The falling edge enables the synchronization signal sync; among which, the rising edge enables the first level control signal when When it rises to high level and the falling edge enables the first level control signal Reduced to low level.

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