A zero input current ripple bipolar self-balanced DC converter and control method

Abstract

A zero-input current ripple bipolar self-equalizing DC converter and control method. The primary side of the converter consists of a two-level interleaved full-bridge circuit and a clamping capacitor, and the secondary side consists of a full-bridge module and a half-bridge module, which share the same arm switching transistors. S 3 and S 4. The primary and secondary sides are connected by two high-frequency transformers. The full bridges on the primary and secondary sides are both driven by fixed 0.5 duty cycle pulses. The two-phase interleaved structure on the primary side achieves zero-input current ripple, and the secondary side achieves self-equalizing output. The switching transistor S 1 lags behind the switching transistor S 3 by an angle D . By controlling D , high voltage boost ratio can realized along with zero voltage switching for all switching transistors on the secondary side.. The invention effectively solves the problems such as zero current ripple, high voltage boost ratio required in photovoltaic power generation systems and voltage imbalance in bipolar DC distribution networks improves the power quality, simplifies system structure, increases power density efficiency simultaneously.

Description

Technical Field

[0001] The present invention belongs to the technical field of power electronics, and in particular relates to a zero-input current ripple bipolar self-balanced DC converter and a control method. Background Art

[0002] The application of photovoltaic power generation system in China is growing rapidly. Photovoltaic power generation system adopts parallel configuration to make full use of electric energy. However, due to the parallel configuration, the output voltage of photovoltaic array is low. In addition, the output current ripple of photovoltaic array is too large, which greatly reduces the life of photovoltaic array. Therefore, DC converter with high step-up ratio and low input current ripple is indispensable in photovoltaic power generation system. In addition, bipolar DC distribution network is superior to traditional unipolar DC distribution network in terms of efficiency, reliability, transmission capacity and safety, and photovoltaic power generation system can be used in bipolar DC distribution network. However, when the load access is unbalanced, it will cause voltage imbalance between the two poles. The literature (W. Wang, Y. Wang, Y. Guan, T. Yao, Y. Wang and D. Xu, "A Family of Impedance Source DC-DC Converters With Zero Input Current Ripple," in IEEE Transactions on Industrial Electronics, vol. 70, no. 9, pp. 8883-8894) proposed a series of impedance source DC converters with zero input current ripple, which achieve ripple-free control by designing the number of coupled inductor turns and input inductance. However, it has unipolar output and low reliability. At the same time, the switch is hard-off, and the switching loss is high. The literature (Z. Yan et al., "Rip ple-Free Bidirectional DC–DC Converter With Wide ZVS Range for Battery Charging / Discharging System," in IEEE Transactions on Industrial Electronics, vol. 70, no. 10, pp. 9992-10002) proposes a DC converter with zero input current ripple and wide ZVS range, but it is a non-isolated structure and the output voltage should not be too high, which limits the voltage gain of the converter.

[0003] The technical differences between this application and the prior art are as follows:

[0004] Patent CN202111119433.5 "A zero current switching full-bridge boost DC converter and its control method"

[0005] Technology Comparison

[0006] Patent CN202111119433.5 adjusts the phase shift time of the leading bridge arm and the lagging bridge arm of the primary full bridge to adjust the voltage gain, and cannot achieve balanced control of the output voltage and zero ripple processing of the input current. The present invention can adjust the voltage gain under different transformer turns ratios by adjusting the phase shift time between the secondary bridge arms of the transformer. At the same time, both the primary full bridge and the secondary full bridge adopt a fixed 0.5 duty cycle control. The primary side achieves zero ripple processing of the input current through parallel staggered inductance; the secondary side can achieve balanced control of the bipolar output voltage.

[0007] Patent CN202111119433.5 achieves zero current switching in a wide load range for all switching devices through frequency modulation control. In the present invention, the volt-second balance of the power transfer inductor within a half-switching cycle is achieved to achieve zero voltage switching in a wide load range for the secondary switch tube; and the zero voltage switching in a wide load range for the primary switch tube is achieved by utilizing the primary excitation inductor current under no-load conditions.

[0008] Technical comparison with patent CN202111119433.5 "A modular DC transformer topology and its control method"

[0009] Patent CN202111119433.5 integrates staggered parallel buck / boost converters to form an MMC structure DAB converter, thereby solving the problem of small input voltage range of MMC structure DAB. However, the present invention is not a modular circuit, and its purpose is to achieve zero ripple processing of converter input current and balanced control of bipolar output voltage.

[0010] Patent CN202111119433.5 achieves voltage matching of bridge arm inductance through decoupling control, and designs corresponding parameters according to the control method to achieve zero current switching of all switches. The present invention adjusts the phase difference between the secondary bridge arms to ensure volt-second balance within half a cycle of the power transmission inductance and, in the no-load state, uses the current ripple flowing through the primary excitation inductance of the transformer to achieve zero voltage switching of all switches within a wide load range. Summary of the invention

[0011] Aiming at the application of bipolar DC distribution network in photovoltaic power generation system, and taking full consideration of factors such as cost, system reliability and loss, the present invention proposes a zero-input current ripple bipolar self-balanced DC converter and control method. The converter realizes zero input current ripple and bipolar output voltage self-balancing, optimizes the control mode, and realizes zero-voltage turn-on of all semiconductor devices in a wide load range. It has the characteristics of low cost, high transmission efficiency and high reliability.

[0012] To achieve the above object, the technical solution adopted by the present invention is:

[0013] A zero input current ripple bipolar self-balanced DC converter, the zero input current ripple bipolar self-balanced DC converter comprising a two-level interleaved full-bridge circuit, a clamping capacitor C c , high frequency transformer T1, T2, high frequency transformer equivalent leakage inductance L r , full-bridge module and half-bridge module; the two-level staggered full-bridge circuit is composed of a full-bridge circuit and inductors L1 and L2, the full-bridge circuit is composed of switch tubes Q1~Q4, one end of the inductors L1 and L2 is connected to the power supply side, and the other end is respectively connected to the series connection point a of the primary first switch tube Q1 and the primary second switch tube Q2 and the series connection point b of the primary third switch tube Q3 and the primary fourth switch tube Q4; the clamping capacitor C c Connected to the third switch tube Q3 and the fourth switch tube Q4 of the full-bridge circuit, the voltage is constant at V c The full-bridge module is composed of an advanced bridge arm and a lagging bridge arm. The advanced bridge arm is composed of the third switch tube S3 of the secondary side and the fourth switch tube S4 of the secondary side connected in forward series, and the lagging bridge arm is composed of the first switch tube S1 and the second switch tube S2 of the secondary side connected in forward series. The two bridge arms are connected in forward parallel. The half-bridge module is composed of the advanced bridge arm of the full-bridge module and a bipolar output capacitor. The bipolar output capacitor is composed of an output capacitor C o1 With the output two-terminal capacitor C o2 The high-frequency transformer T1 and T2 are connected in series in a forward direction and are connected in parallel with the lagging bridge arm of the full-bridge module; the primary windings of the high-frequency transformers T1 and T2 are connected in a forward direction in series between the series point a of the primary first switch tube Q1 and the primary second switch tube Q2 and the series point b of the primary third switch tube Q3 and the primary fourth switch tube Q4; the secondary winding of the high-frequency transformer T1 is connected in a forward direction in series between the series point e of the secondary third switch tube S3 and the secondary fourth switch tube S2 and the series point g of the secondary first switch tube S1 and the secondary second switch tube S2; the secondary winding of the high-frequency transformer T2 is connected in a forward direction in series between the series point e of the secondary third switch tube S3 and the secondary fourth switch tube S4 and the output capacitor C o1 With the output two-terminal capacitor C o2 Between the series points f.

[0014] The DC converter of the present invention is further improved, the converter adopts phase shift control, and the switching period is T s .

[0015] The present invention provides a control method for a zero input current ripple bipolar self-balanced DC converter, wherein the driving signals of the primary switch tubes Q1-Q4 have the same fixed duty cycle of 0.5, so that the current ripples of the inductors L1 and L2 cancel each other out, and realize zero input current ripple, and the driving signals of the secondary switch tubes S1-S4 have the same fixed duty cycle of 0.5, so as to realize balanced control of the bipolar output voltage, and the driving signal of the first primary switch tube Q1 also leads the driving signal of the third secondary switch tube S3 by a time of Through closed loop control Output power regulation can be achieved;

[0016] To ensure that the secondary switch tubes S1~S4 achieve zero voltage switching, the clamping capacitor voltage is stabilized at 2V in The time lag of the first switch tube S1 on the secondary side and the third switch tube on the secondary side is DT s By adjusting D, zero voltage switching in the full load range of the third switch tube S3 and the fourth switch tube S4 on the secondary side and zero voltage switching in the wide load range of the first switch tube S1 and the second switch tube S2 on the secondary side can be achieved;

[0017] Since in the no-load condition, that is, the bipolar output resistance R o1 and R o2 When both are 0, no load current flows through the primary switch tubes Q1~Q4 of the transformer, and their switching action will generate ripple current in the inductors L1 and L2. The ripple current can d The charging and discharging process of the junction capacitance of the primary switch tubes Q1 to Q4 is completed, providing the necessary conditions for zero voltage switching, thereby realizing zero voltage switching of the primary semiconductor devices within a wide load range.

[0018] The method of the present invention is further improved, wherein the converter has a switching cycle T s It is divided into 6 switching modes. The 6 switching modes within the switching period [t0~t6] are as follows;

[0019] Mode 1 t0~t1: Before t0, switch tubes S2 and S4 are turned on. At t0, switch tubes Q1 and Q4 are turned on with zero voltage, and the voltage between points ab is v ab =V c ; the voltage between points eg is 0; the voltage between points ef is v ef -V o / 2, since the turns ratio of high-frequency transformers T1 and T2 is N, the voltage between points cd is v cd V o / 2N, inductor L1 current i L1 The inductor L2 current i L2 The inductor current iLr The expression is:

[0020]

[0021] Mode 2 t1~t2: At t1, the switch tube S4 is turned off, and the secondary current i of high-frequency transformer T1 and T2 is g with i f The sum of the junction capacitance of the switch tube S3 is discharged, and the junction capacitance of the switch tube S4 is charged until the drain-source voltage of the switch tube S3 drops to zero. Next, the body diode of the switch tube S3 is turned on to ensure that the switch tube S3 is turned on at zero voltage in the next stage. Since the junction capacitance value is small, its charging and discharging process is ignored. Therefore, the ZVS condition of the switch tube S3 is:

[0022] i g (t1)+i f (t1)<0

[0023] Mode 3 t2~t3: At t2, the switch tube S3 is turned on with zero voltage, and the voltage between points ab is v ab =V c ;eg the voltage between two points is V o ; ef two-point voltage v ef V o / 2, since the turns ratio of high-frequency transformers T1 and T2 is N, the voltage between points cd is v cd 3V o / 2N, at this time the inductor current i Lr The expression is:

[0024]

[0025] Mode 4 t3~t4: At t3, the switch tube S2 is turned off, and the secondary current i of the high-frequency transformer T1 is g The junction capacitance of the switch tube S2 is charged, and the junction capacitance of the switch tube S1 is discharged until the drain-source voltage of the switch tube S1 drops to zero. Next, the body diode of the switch tube S1 is turned on to ensure that the switch tube S1 is turned on at zero voltage in the next stage. Since the junction capacitance value is small, its charging and discharging process is ignored. Therefore, the ZVS condition of the switch tube S1 is:

[0026] i g (t3)>0

[0027] Mode 5 t4~t5: At t4, the switch tube S1 is turned on with zero voltage, and the voltage between points ab is v ab =V c ; the voltage between points eg is 0; the voltage between points ef is v ef V o / 2, since the turns ratio of high-frequency transformers T1 and T2 is N, the voltage between points cd is v cd V o / 2N, at this time the inductor current i Lr The expression is:

[0028]

[0029] Mode 6 t5~t6: At t5, switch tubes Q1 and Q4 are turned off, and the inductor current i L1 and leakage inductance current i Lr The sum of the two charges the junction capacitance of the switch tube Q1 and discharges the junction capacitance of the switch tube Q2 until the drain-source voltage of the switch tube Q2 drops to zero. In addition, the inductor current i L2 and leakage inductance current i Lr The difference between the two charges the junction capacitance of switch tube Q4 and discharges the junction capacitance of switch tube Q3 until the drain-source voltage of switch tube Q3 drops to zero. Next, the body diodes of switch tubes Q2 and Q3 are turned on to ensure that the next stage of switch tubes Q2 and Q3 are turned on at zero voltage. Since the junction capacitance is small, its charging and discharging process is ignored. Therefore, the ZVS conditions of switch tubes Q2 and Q3 are respectively

[0030] i L1 (t5)+i Lr (t5)<0

[0031] i L2 (t5)-i Lr (t5)>0.

[0032] Compared with the prior art, the present invention has the following advantages:

[0033] 1) The converter achieves zero input current ripple and self-balancing control of output voltage by adopting a pulse drive with a fixed duty cycle of 0.5;

[0034] 2) The converter can achieve zero voltage switching of all semiconductor devices within a wide load range by adopting a phase-shift control method and utilizing the current ripple of the input inductor. BRIEF DESCRIPTION OF THE DRAWINGS

[0035] Figure 1 is the main circuit of the present invention;

[0036] Figure 2 is the current flow diagram of the converter in mode 1;

[0037] Figure 3 is the current flow diagram of the converter in mode 2;

[0038] Figure 4 is the current flow diagram of the converter in mode three;

[0039] Figure 5 is the current flow diagram of the converter in mode four;

[0040] Figure 6 is the current flow diagram of the converter in mode five;

[0041] Figure 7 is the current flow diagram of the converter in mode six;

[0042] Figure 8 It is the main waveform diagram of the converter;

[0043] Fig. 9 is the leakage inductance L of the converter r Voltage waveform at both ends;

[0044] Fig.10 This is the relationship between the voltage gain M and the duty cycle D of the converter under different turns ratios N;

[0045] Fig.11 This is the volt-second balance principle diagram of the secondary winding N2 of the high-frequency transformer of the converter;

[0046] Fig.12 The leakage inductance L of the converter is balanced when the bipolar load is r Voltage and current waveforms;

[0047] Fig.13 This is the zero voltage switching range diagram of the secondary side switches S1 and S2 of the converter;

[0048] Fig.14 This is the waveform of the primary switch tube current when the converter is unloaded;

[0049] Fig.15 The leakage inductance L of the converter is balanced when the bipolar load is r Voltage and current waveforms. DETAILED DESCRIPTION

[0050] The present invention is further described in detail below in conjunction with the accompanying drawings and specific embodiments:

[0051] Implementation plan: See Figure 1 , is the main circuit of the zero input current ripple bipolar self-balanced voltage w DC converter, including a two-level interleaved full-bridge circuit, a clamping capacitor C c , high frequency transformer T1, T2, high frequency transformer equivalent leakage inductance L r, full-bridge module, half-bridge module; the two-level staggered full-bridge circuit is composed of a full-bridge circuit and inductors L1 and L2, the full-bridge circuit is composed of switch tubes Q1~Q4, one end of the inductors L1 and L2 is connected to the power supply side, and the other end is respectively connected to the series connection point a of the primary first switch tube Q1 and the primary second switch tube Q2 and the series connection point b of the primary third switch tube Q3 and the primary fourth switch tube Q4; the clamping capacitor C c Connected to the third switch tube Q3 and the fourth switch tube Q4 of the full-bridge circuit, the voltage is constant at V c The full-bridge module is composed of an advanced bridge arm and a lagging bridge arm. The advanced bridge arm is composed of the third switch tube S3 of the secondary side and the fourth switch tube S4 of the secondary side connected in forward series, and the lagging bridge arm is composed of the first switch tube S1 and the second switch tube S2 of the secondary side connected in forward series. The two bridge arms are connected in forward parallel. The half-bridge module is composed of the advanced bridge arm of the full-bridge module and a bipolar output capacitor. The bipolar output capacitor is composed of an output capacitor C o1 With the output two-terminal capacitor C o2 The high-frequency transformer T1 and T2 are connected in series in a forward direction and are connected in parallel with the lagging bridge arm of the full-bridge module; the primary windings of the high-frequency transformers T1 and T2 are connected in a forward direction in series between the series point a of the primary first switch tube Q1 and the primary second switch tube Q2 and the series point b of the primary third switch tube Q3 and the primary fourth switch tube Q4; the secondary winding of the high-frequency transformer T1 is connected in a forward direction in series between the series point e of the secondary third switch tube S3 and the secondary fourth switch tube S2 and the series point g of the secondary first switch tube S1 and the secondary second switch tube S2; the secondary winding of the high-frequency transformer T2 is connected in a forward direction in series between the series point e of the secondary third switch tube S3 and the secondary fourth switch tube S4 and the output capacitor C o1 With the output two-terminal capacitor C o2 The converter adopts phase shift control, and the switching period is T s , where the phase shift control mode is that the driving signals of the primary switch tubes Q1~Q4 have the same fixed duty cycle of 0.5, so that the current ripples of the inductors L1 and L2 cancel each other out and achieve zero input current ripple. The driving signals of the secondary switch tubes S1~S4 have the same fixed duty cycle of 0.5, so as to achieve balanced control of the bipolar output voltage. The driving signal of the first primary switch tube Q1 leads the driving signal of the third secondary switch tube S3 by a time of Through closed loop control Output power regulation can be achieved. At the same time, the driving signal of the third switch tube S3 on the secondary side also leads the time of the first switch tube S1 on the secondary side by DT s , clamping capacitor C c Voltage is 2V in .

[0052] The proposed converter has a switching cycle T s It can be divided into 6 switching modes, and its main waveforms are as follows Figure 8 As shown. The following is a detailed description of the six switching modes within the switching cycle [t0~t6].

[0053] Mode 1 (t0~t1): The current path of this mode is as follows Figure 2 As shown. Before t0, switch tubes S2 and S4 are turned on. At t0, switch tubes Q1 and Q4 are turned on with zero voltage. The voltage between points ab is v ab =V c ; the voltage between points eg is 0; the voltage between points ef is v ef -V o / 2. Since the turns ratio of high-frequency transformers T1 and T2 is N, the voltage between points cd is v cd V o / 2N. Inductor L1 current i L1 The inductor L2 current i L2 Linear rise, due to the staggered structure, the input current can be ripple-free.

[0054] At this time, the inductor current i Lr The expression is

[0055]

[0056] Mode 2 (t1~t2): The current path of this mode is as follows Figure 3 At time t1, the switch tube S4 is turned off, and the secondary current i of high-frequency transformers T1 and T2 is g with i f The sum of the junction capacitance of switch tube S3 is discharged, and the junction capacitance of switch tube S4 is charged, until the drain-source voltage of switch tube S3 drops to zero. Next, the body diode of switch tube S3 is turned on to ensure that the switch tube S3 is turned on at zero voltage in the next stage. Since the junction capacitance is small, its charging and discharging process can be ignored. Therefore, the ZVS condition of switch tube S3 is

[0057] i g (t1)+i f (t1)<0 (2)

[0058] Mode 3 (t2~t3): The current path of this mode is as follows Figure 4 As shown. At t2, the switch tube S3 is turned on with zero voltage. The voltage between points ab is v ab =V c ;eg the voltage between two points is V o ; ef two-point voltage v ef V o / 2. Since the turns ratio of high-frequency transformers T1 and T2 is N, the voltage between points cd is v cd 3V o / 2N. At this time, the inductor current i Lr The expression is

[0059]

[0060] Mode 4 (t3~t4): The current path of this mode is as follows Figure 5 As shown. At t3, the switch tube S2 is turned off. The secondary current i g The junction capacitance of switch S2 is charged and the junction capacitance of switch S1 is discharged until the drain-source voltage of switch S1 drops to zero. Next, the body diode of switch S1 is turned on to ensure that the switch S1 is turned on at zero voltage in the next stage. Since the junction capacitance is small, its charging and discharging process can be ignored. Therefore, the ZVS condition of switch S1 is

[0061] i g (t3)>0 (4)

[0062] Mode 5 (t4-t5): The current path of this mode is as follows Figure 6 As shown. At t4, the switch tube S1 is turned on with zero voltage. The voltage between points ab is v ab =V c ; the voltage between points eg is 0; the voltage between points ef is v ef V o / 2. Since the turns ratio of high-frequency transformers T1 and T2 is N, the voltage between points cd is v cd V o / 2N. At this time, the inductor current i Lr The expression is

[0063]

[0064] Mode 6 (t5-t6): The current path of this mode is as follows Figure 7 As shown. At t5, the switch tubes Q1 and Q4 are turned off. The inductor current i L1 and leakage inductance current i Lr The sum of the two charges the junction capacitance of the switch tube Q1 and discharges the junction capacitance of the switch tube Q2 until the drain-source voltage of the switch tube Q2 drops to zero. In addition, the inductor current i L2 and leakage inductance current i Lr The difference between the two charges the junction capacitance of switch tube Q4 and discharges the junction capacitance of switch tube Q3 until the drain-source voltage of switch tube Q3 drops to zero. Next, the body diodes of switch tubes Q2 and Q3 are turned on to ensure that the next stage of switch tubes Q2 and Q3 are turned on at zero voltage. Since the junction capacitance is small, its charging and discharging process can be ignored. Therefore, the ZVS conditions of switch tubes Q2 and Q3 are respectively

[0065] i L1 (t5)+iLr (t5)<0 (6)

[0066] i L2 (t5)-i Lr (t5)>0 (7)

[0067] Depend on Figure 2 It can be seen that the input side switches Q1, Q2, Q3, Q4 are driven by fixed 50% duty cycle pulses, and the drive signals are staggered 180° with each other. L1 and i L2 The current ripples can cancel each other out, resulting in a zero ripple input current. r The voltage is v ab With v cd The difference, where v ab is a two-level waveform, v cd It is a four-level waveform to meet the larger input voltage variation range. Therefore, the corresponding volt-second balance schematic diagram is as follows Fig. 9 As shown, the inductor L can be realized r The volt-second balance. Where S1 and S2 are v ab With v cd The positive volt-second area.

[0068]

[0069] To realize the leakage inductance L r The volt-second balance is improved, and the zero voltage switching range of the secondary switch tube is increased, then S1 = S2, and the expression of the voltage gain M is derived as follows:

[0070]

[0071] The relationship between the converter voltage gain and duty cycle D under different transformer turns ratios N is as follows: Fig.10 As shown in the figure, it can be seen that the voltage gain range of the designed converter increases with the increase of the turns ratio N, and the voltage gain becomes smaller with the increase of the duty cycle D at the same turns ratio N. To simplify the design, the transformer turns ratio can be designed as a fixed value, and the voltage gain can be adjusted by adjusting the duty cycle D.

[0072] From formula (10), it can be concluded that when S1=S2, the expression of duty cycle D is:

[0073]

[0074] Transformer T2 secondary winding N s2 The relationship between the voltage and the secondary switch tube drive signal waveform is as follows: Fig.11Assume that the duty cycle of the driving pulses of the secondary switch tubes S1 and S4 is G, and the duty cycle of the switch tubes S2 and S3 is 1-G, then according to the volt-second balance principle, we can get:

[0075] V o1 (1-G)T s -V o2 GT s =0 (12)

[0076] From equation (12), the bipolar output voltage V o1 With V o2 The relationship is:

[0077]

[0078] From equation (12), it can be seen that as long as the secondary switch of the converter is controlled by 50% duty cycle, its bipolar voltage level can be balanced. In addition, since equation (13) is valid under all load conditions, the bipolar voltage level can be balanced over the entire range of unbalanced loads, even when one side of the output end is unloaded.

[0079] Based on the analysis of the working mode, the zero voltage switching conditions of the primary and secondary side switches under bipolar load balance are shown in Table 1.

[0080] Table 1 Zero voltage turn-on conditions of the primary and secondary switch tubes

[0081]

[0082] When the bipolar load is balanced, the leakage inductance L r Voltage and current waveforms are as follows Fig.12 Ignore the dead zone of the switch tube and assume that the excitation inductance is large enough, because the bipolar load is balanced at this time, the current i Lr There is no DC component passing through, so we can get

[0083]

[0084] Since the inductor current i Lr Symmetry, according to equations (1), (3), and (5), we can obtain the following relationship:

[0085]

[0086] Through equations (14) and (15), we can obtain:

[0087]

[0088] Substituting equation (16) into Table 1, the zero voltage turn-on condition of the secondary side switch tube is shown in Table 2.

[0089] Table 2 Zero voltage turn-on conditions for the secondary switch

[0090]

[0091] If the L deduced above is satisfied r The volt-second balance principle within half a switching cycle can be simplified according to the table in formula (11) as follows:

[0092] Table 3: Zero voltage turn-on conditions of the secondary switch when equation (11) is satisfied

[0093]

[0094] As shown in Table 3, the transformer secondary leading bridge arm switch tubes S3 and S4 can achieve zero voltage switching within the full load range. The ZVS range of the lagging bridge arm switch tubes S1 and S2 is as follows: Fig.13 As shown in Table 2, according to the converter operating conditions, its ZVS range is wider than that in Table 2.

[0095] The current expression of the primary DC inductor L1, L2 is:

[0096]

[0097] Substituting equation (17) into Table 1, the zero voltage turn-on condition of the primary switch can be expressed as

[0098] Table 4 Zero voltage turn-on conditions for the primary switch

[0099]

[0100] For the primary switch tube of the transformer, the current ripple of inductors L1 and L2 can be used to charge and discharge the junction capacitor under light load or even no-load conditions, and the ZVS range is expanded. Under no-load conditions, that is, At this time, the zero voltage turn-on conditions of the primary switch tube are shown in Table 5.

[0101] Table 5 Zero voltage turn-on conditions of the primary switch tube under no-load conditions

[0102]

[0103]

[0104] It can be seen that the primary switch tube can achieve zero voltage switching in the full load range including no-load, and the ZVS range is expanded. Fig.14 As shown, during the dead time t d There is still enough negative current flowing through the primary switch tube to complete the charging and discharging process of the primary switch tube junction capacitance. Its peak value i bias(peak) for

[0105]

[0106] When the bipolar load is unbalanced, assuming that the bipolar output resistance R o1 >R o2 , there will be a DC bias current I at the midpoint of the bipolar capacitor dc flows through, its value is V o2 / R o2 -V o1 / R o1 , that is, the secondary current i f Compared to the bipolar equilibrium state, the magnitude is I dc The DC bias component of the transformer. According to the transformer turns ratio conversion relationship, the primary leakage inductance current i Lr and the secondary current i g The load balancing state has a size of NI dc and I dc The DC bias component. At this time, L r The voltage and current waveforms are as follows Fig.15 shown.

[0107] According to formula (15), the inductor current value at each moment is

[0108]

[0109] Under the condition of unbalanced bipolar load, the zero voltage turn-on range of the compound switch is shown in Table 6.

[0110] Table 6: Zero voltage turn-on conditions for the secondary switch when the bipolar load is unbalanced

[0111]

[0112] Compared with Table 3, when the bipolar output resistor R o1 >R o2 , that is I dc >0, the ZVS range of switches S1 and S2 becomes larger, and the greater the unbalanced degree of bipolar load, the wider the ZVS range. The ZVS range of switches S3 and S4 becomes smaller, and the greater the unbalanced degree of bipolar load, the narrower the ZVS range. When the bipolar output resistor R o1 <R o2 , that is I dc <0, the ZVS range of switch tubes S1 and S2 becomes smaller, and the greater the unbalanced degree of bipolar load, the narrower the ZVS range. While the ZVS range of switch tubes S3 and S4 becomes larger, and the greater the unbalanced degree of bipolar load, the wider the ZVS range.

[0113] For the primary switch tube, according to Table 4, when the bipolar output resistor Ro1 >R o2 , that is I dc >0, the ZVS range of the secondary switch becomes larger, and the greater the unbalanced degree of the bipolar load, the wider the ZVS range; when the bipolar output resistance R o1 <R o2 , that is I dc <0, the ZVS range of the secondary side switch tube becomes smaller, and the greater the imbalance of the bipolar load, the narrower the ZVS range.

[0114] The above description is only a preferred embodiment of the present invention and does not constitute any other form of limitation to the present invention. Any modification or equivalent change made based on the technical essence of the present invention still falls within the scope of protection required by the present invention.

Claims

1. A zero input current ripple bipolar self-balanced DC converter, characterized in that: The zero input current ripple bipolar self-balanced DC converter comprises a two-level interleaved full-bridge circuit, a clamping capacitor C c , high frequency transformer T1, high frequency transformer T2, high frequency transformer equivalent leakage inductance L r , full-bridge module and half-bridge module; the two-level staggered full-bridge circuit is composed of a full-bridge circuit and inductors L1 and L2, the full-bridge circuit is composed of switch tubes Q1~Q4, one end of the inductors L1 and L2 is connected to the power supply side, and the other end is respectively connected to the series connection point a of the primary first switch tube Q1 and the primary second switch tube Q2 and the series connection point b of the primary third switch tube Q3 and the primary fourth switch tube Q4; the clamping capacitor C c Connected to the third switch tube Q3 and the fourth switch tube Q4 of the full-bridge circuit, the voltage is constant at V c The full-bridge module is composed of an advanced bridge arm and a lagging bridge arm. The advanced bridge arm is composed of the third switch tube S3 of the secondary side and the fourth switch tube S4 of the secondary side connected in forward series, and the lagging bridge arm is composed of the first switch tube S1 and the second switch tube S2 of the secondary side connected in forward series. The two bridge arms are connected in forward parallel. The half-bridge module is composed of the advanced bridge arm of the full-bridge module and a bipolar output capacitor. The bipolar output capacitor is composed of an output capacitor C o1 With the output two-terminal capacitor C o2 The high-frequency transformer T1 and the high-frequency transformer T2 are connected in series in a forward direction and are connected in parallel with the lagging bridge arm of the full-bridge module; the primary windings of the high-frequency transformer T1 and the high-frequency transformer T2 are connected in a forward direction in series between the series point a of the primary first switch tube Q1 and the primary second switch tube Q2 and the series point b of the primary third switch tube Q3 and the primary fourth switch tube Q4; the secondary winding of the high-frequency transformer T1 is connected in a forward direction in series between the series point e of the secondary third switch tube S3 and the secondary fourth switch tube S2 and the series point g of the secondary first switch tube S1 and the secondary second switch tube S2; the secondary winding of the high-frequency transformer T2 is connected in a forward direction in series between the series point e of the secondary third switch tube S3 and the secondary fourth switch tube S4 and the output capacitor C o1 With the output two-terminal capacitor C o2 Between the series points f; The converter adopts phase shift control with a switching period of T s , wherein the phase shift control mode is that the driving signals of the primary switch tubes Q1 to Q4 have the same fixed duty cycle of 0.5, the driving signals of the secondary switch tubes S1 to S4 have the same fixed duty cycle of 0.5, and the driving signal of the first primary switch tube Q1 leads the driving signal of the third secondary switch tube S3 by a time of At the same time, the driving signal of the third switch tube S3 on the secondary side also leads the driving signal of the first switch tube S1 on the secondary side by a time DT s .

2. The control method of the zero input current ripple bipolar self-balanced DC converter according to claim 1, characterized in that: The converter has a switching cycle T s It is divided into 6 switching modes. The 6 switching modes in the switching cycle [t0~t6] are as follows; Mode 1 t0~t1: Before t0, switch tubes S2 and S4 are turned on. At t0, switch tubes Q1 and Q4 are turned on with zero voltage. The voltage v between points ab is ab =V c ; the voltage between points eg is 0; the voltage between points ef is v ef -V o / 2, since the turns ratio of high-frequency transformers T1 and T2 is N, the voltage between points cd is v cd V o / 2N, inductor L1 current i L1 The inductor L2 current i L2 Linear rise, due to the staggered structure, the input current can be ripple-free. At this time, the inductor current i Lr The expression is: Mode 2 t1~t2: At t1, the switch tube S4 is turned off, and the secondary current i of high-frequency transformer T1 and T2 is g with i f The sum of the junction capacitance of the switch tube S3 is discharged, and the junction capacitance of the switch tube S4 is charged until the drain-source voltage of the switch tube S3 drops to zero. Next, the body diode of the switch tube S3 is turned on to ensure that the switch tube S3 is turned on at zero voltage in the next stage. Since the junction capacitance value is small, its charging and discharging process is ignored. Therefore, the ZVS condition of the switch tube S3 is: in g (t1)+i f (t1)<0 Mode 3 t2~t3: At t2, the switch tube S3 is turned on with zero voltage, and the voltage between points ab is v ab =V c ;eg the voltage between two points is V o ; ef two-point voltage v ef V o / 2, since the turns ratio of high-frequency transformers T1 and T2 is N, the voltage between points cd is v cd 3V o / 2N, at this time the inductor current i Lr The expression is: Mode 4 t3~t4: At t3, the switch tube S2 is turned off, and the secondary current i of the high-frequency transformer T1 is g The junction capacitance of the switch tube S2 is charged, and the junction capacitance of the switch tube S1 is discharged until the drain-source voltage of the switch tube S1 drops to zero. Next, the body diode of the switch tube S1 is turned on to ensure that the switch tube S1 is turned on at zero voltage in the next stage. Since the junction capacitance value is small, its charging and discharging process is ignored. Therefore, the ZVS condition of the switch tube S1 is: i g (t3)>0 Mode 5 t4~t5: At t4, the switch tube S1 is turned on with zero voltage, and the voltage between points ab is v ab =V c ; the voltage between points eg is 0; the voltage between points ef is v ef V o / 2, since the turns ratio of high-frequency transformers T1 and T2 is N, the voltage between points cd is v cd V o / 2N, at this time the inductor current i Lr The expression is: Mode 6 t5~t6: At t5, the switch tubes Q1 and Q4 are turned off, and the inductor current i L1 and leakage inductance current i Lr The sum of the two charges the junction capacitance of the switch tube Q1 and discharges the junction capacitance of the switch tube Q2 until the drain-source voltage of the switch tube Q2 drops to zero. In addition, the inductor current i L2 and leakage inductance current i Lr The difference between the two charges the junction capacitance of switch tube Q4 and discharges the junction capacitance of switch tube Q3 until the drain-source voltage of switch tube Q3 drops to zero. Next, the body diodes of switch tubes Q2 and Q3 are turned on to ensure that the next stage of switch tubes Q2 and Q3 are turned on at zero voltage. Since the junction capacitance is small, its charging and discharging process is ignored. Therefore, the ZVS conditions of switch tubes Q2 and Q3 are respectively i L1 (t5)+i Lr (t5)<0 i L2 (t5)-i Lr (t5)>0。

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