Transient process control method of dual active bridge series resonant converter in discontinuous conduction mode

Abstract

The present invention discloses a transient process control method for a dual active bridge series resonant converter in discontinuous conduction mode, which belongs to the field of DC converter control. When the converter transmits a power instruction that suddenly changes in the kth switching cycle, the duty cycle of the first half of the switching cycle is calculated based on the charge amount change value before and after the transient state, and the steady-state duty cycle D is restored from the second half of the switching cycle. k , thus ending the transient process within half a switching cycle. In order to achieve fast control within a larger transmission power mutation range, the time of the first half cycle in the kth switching cycle is set to half a resonant cycle, and the duty cycle of the second half switching cycle is used for fast transient control, and the steady-state duty cycle D is restored from the k+1th switching cycle. k This method can eliminate the overshoot and bias of the state variables during the sudden change of the transmission power command, and achieve a smooth and fast transition between two power levels within one to two switching cycles, thereby improving the dynamic performance of the converter.

Description

Technical Field

[0001] The present invention belongs to the field of direct current converter control, and more specifically, relates to a transient process control method of a dual active bridge series resonant converter in a discontinuous conduction mode. Background Art

[0002] With the rapid development of renewable energy generation and the increasing use of DC loads, DC transformers have become a research hotspot in the power electronics field in recent years. The dual-bridge series resonant converter (DBSRC) connects two full-bridge modules via a series resonant circuit and a high-frequency transformer, resulting in a near-sinusoidal current waveform and minimal current stress on the switch tubes. This structure offers the DBSRC advantages of low switching stress, a wide soft switching range, and high power density. Consequently, DBSRC has found widespread application in various fields, including renewable energy generation, electric vehicles, and energy storage systems.

[0003] However, under light load conditions, DBSRCs tend to lose their soft-switching characteristics, reducing converter efficiency. Existing literature typically utilizes discontinuous conduction mode (DCM) operation of the inductor current to achieve zero-current switching, thereby reducing switching losses. This approach has recently become a hot topic for improving DBSRC light-load efficiency. In actual operation, DC transformers often need to switch between different power levels, and the dynamic performance of this transient process can affect the transformer's reliability and stability. In continuous conduction mode (CCM), it has been clearly pointed out that the coupling between the resonant inductor and resonant capacitor can cause severe oscillations in the DBSRC transient process, and this has been analyzed in detail. The transient process in DCM differs slightly from that in CCM in the following ways. First, the inductor current is discontinuous during each switching cycle, and its initial value is always zero, which does not affect the transient process. Second, when the controlled variable undergoes a step change, the initial and final steady-state values ​​of the capacitor voltage are not equal. Since the capacitor voltage cannot change suddenly immediately, a transient process is required for the capacitor voltage to transition to the new steady-state. Furthermore, due to the coupling between the inductor current and capacitor voltage in the resonant cavity, even though the current is intermittent within each switching cycle, the transient state of the capacitor voltage also causes the inductor current to experience a transient state. Finally, simulations revealed that the transition time of the DCM transient process is still much longer than the switching period, hindering the improvement of closed-loop control performance.

[0004] Existing methods improve closed-loop control performance by optimizing controllers, such as proportional-integral controllers and model predictive controllers. However, these methods design controller parameters based on steady-state models, without considering transient processes. Furthermore, these methods place high demands on parameters for optimal control performance. Therefore, analyzing the transient processes of DCM modulation and developing a fast transient control method are crucial for improving both the open-loop and closed-loop dynamic performance of converters. Summary of the Invention

[0005] In response to the defects of the existing technology and the need for improvement, the present invention provides a transient process control method for a dual active bridge series resonant converter in discontinuous conduction mode. The purpose is to ensure that when the transmission power instruction changes suddenly, that is, when the control duty cycle changes in a step, the DBSRC can end the transient process within one to two switching cycles and smoothly transition to a new stable operating state, thereby improving the dynamic adjustment speed of the converter.

[0006] To achieve the above object, according to one aspect of the present invention, a transient process control method in a discontinuous conduction mode of a dual active bridge series resonant converter is provided. The dual active bridge series resonant converter includes a primary full bridge H1 and a secondary full bridge H2. H1 is connected to the primary side of the transformer through an LC resonant cavity, and H2 is connected to the secondary side of the transformer. When f s <f r , Q k >Q k-1 And ΔQ k -Q k-1 <4C(V1-NV2), or when f s <f r , Q k k-1 And ΔQ k-1 -Q k <4CNV2, or when f s >f r and|Q k -Q k-1 When |>ΔQ, the method includes: in the current switching cycle, the duty cycle of the AC side voltage of H1 is controlled to be equal to D in the first and second half switching cycles respectively. c 、D k ; From the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k , D c for:

[0007]

[0008] Among them, D c is the first intermediate duty cycle, f s is the switching frequency, f r is the resonant frequency, Q k D k The charge of the down converter, Q k-1 D k-1 The charge of the down converter, D k is the duty cycle instruction of the current switching cycle, D k-1 ​​​is the duty cycle instruction of the previous switching cycle, ΔQ is the minimum charge difference of the converter during transient process, C is the resonant capacitance in the LC resonant cavity, N represents the turns ratio of the transformer, V1 is the input DC voltage, V2 is the load voltage, and F n is the frequency ratio, F n =f s / f r , M is the voltage conversion ratio.

[0009] Furthermore, when f s <f r , Q k >Q k-1 And 4C (V1-NV2) k -Q k-1 <8C(V1-NV2), the method further includes: in the current switching cycle, the duty cycle of the AC side voltage of H1 is controlled to be equal to δ·(F n / 2), the second intermediate duty cycle D c1 , δ is the correction coefficient; from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

[0010] Furthermore, when f s <f r , Q k >Q k-1 And 8C (V1-NV2) k -Q k-1 When the method further includes: S11, in the current switching cycle, controlling the duty cycle of the AC side voltage of H1 to be equal to δ·(F n / 2), and Q k-1 Updated to Q k-1 +8C(V1-NV2), δ is the correction coefficient; S12, if the updated Q k -Q k-1 <4C(V1-NV2), in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1 to be equal to D c 、D k , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k ; If updated 4C (V1-NV2) k -Q k-1 <8C(V1-NV2), in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1, which is equal to δ·(F n 2) Second intermediate duty cycle D c1 , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k ​​​; If updated to 8C (V1-NV2) k -Q k-1 , the next switching cycle is taken as the new current switching cycle, and the process returns to S11 again until Q k -Q k-1 <8C(V1-NV2) no longer returns to execute the above S11.

[0011] Furthermore, D c1 for:

[0012]

[0013] Furthermore, when f s <f r , Q k k-1 And 4CNV2 k-1 -Q k <8CNV2, the method further includes: in the current switching cycle, the first and second half switching cycles respectively control the duty cycle of the H1 AC side voltage to be equal to 0, the third intermediate duty cycle D c2 ; From the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

[0014] Furthermore, when f s <f r , Q k k-1 And 8CNV2 k-1 -Q k The method further includes: S21, in the current switching cycle, controlling the duty cycle of the AC side voltage of H1 to be equal to 0, and setting Q k-1 Updated to Q k-1 -8CNV2; S22, if Q is updated k-1 -Q k <4CNV2, in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1 to be equal to D c 、D k , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k ; If 4CNV2 is updated k-1 -Q k <8CNV2, in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1 to be equal to 0, the third intermediate duty cycle D c2 , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k ; If updated 8CNV2 k-1 -Q k ​​​​​​​, the next switching cycle is taken as the new current switching cycle, and the process returns to S21 again until Q k-1 -Q k <8CNV2 no longer returns to execute S21.

[0015] Furthermore, D c2 for:

[0016]

[0017] Furthermore, when |Q k -Q k-1 |≤Δq, the method further includes: directly updating the duty cycle of the AC side voltage of H1 to D k .

[0018] According to another aspect of the present invention, a transient process controller for a dual active bridge series resonant converter in discontinuous conduction mode is provided, wherein the controller is used to execute the transient process control method for the dual active bridge series resonant converter in discontinuous conduction mode as described above.

[0019] In general, the above technical solutions conceived by the present invention can achieve the following beneficial effects:

[0020] (1) A transient process control method for a dual active bridge series resonant converter in discontinuous conduction mode is provided. When a transient event (i.e. |Q k -Q k-1 |>ΔQ), by modifying the duty cycle of the primary full-bridge AC side voltage in the first half switching cycle after the transient occurs to D c , the transient transition time can be controlled within half a switching cycle, greatly shortening the transient transition time, achieving a smooth transition of the transient process, protecting the safe operation of the converter, and having good dynamic adjustment capabilities;

[0021] (2) Further, considering that only D c Acting within half a switching cycle after the transient occurs, it is possible that D c When the limit value is reached, the fast transient control method (FTCM) reaches the maximum adjustment limit. On this basis, an improved fast transient control method (IFTCM) is proposed for the cases of sudden increase and sudden decrease of duty cycle respectively.

[0022] Specifically, when the duty cycle suddenly increases, according to Q k -Q k-1 The relationship between 8C (V1-NV2) and 4C (V1-NV2) is as follows: k -Q k-1The relationship between 8CNV2 and 4CNV2 controls the duty cycle of the primary full-bridge AC side voltage after a transient occurs, avoiding reaching the maximum regulation limit. This keeps the transient transition time within 1-2 switching cycles, ensuring dynamic regulation capability.

[0023] (3) The simulation results show that this method improves the dynamic performance of DBSRC in open-loop and closed-loop conditions, laying the foundation for the reliable operation and wide application of the converter. On this basis, this method is combined with direct charge closed-loop control to improve the effect of closed-loop control. BRIEF DESCRIPTION OF THE DRAWINGS

[0024] Figure 1 An overall block diagram of the DBSRC transient process control provided by an embodiment of the present invention;

[0025] Figure 2 Waveform diagram of the DCM modulation method of DBSRC provided in an embodiment of the present invention;

[0026] Figure 3 A schematic diagram of an FTCM provided in an embodiment of the present invention;

[0027] Figure 4 A schematic diagram of an IFTCM when the duty cycle suddenly increases according to an embodiment of the present invention;

[0028] Figure 5 A control flow chart of IFTCM when the duty cycle suddenly increases according to an embodiment of the present invention;

[0029] Figure 6 A schematic diagram of an IFTCM when the duty cycle suddenly decreases according to an embodiment of the present invention;

[0030] Figure 7 A control flow chart of IFTCM when the duty cycle suddenly decreases provided by an embodiment of the present invention;

[0031] Figure 8 An open-loop transient control simulation diagram provided by an embodiment of the present invention;

[0032] Figure 9 A closed-loop simulation diagram without transient control provided by an embodiment of the present invention;

[0033] Figure 10 A closed-loop simulation diagram using transient control is provided in an embodiment of the present invention. DETAILED DESCRIPTION

[0034] In order to make the objectives, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely for the purpose of explaining the present invention and are not intended to limit the present invention. In addition, the technical features involved in the various embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.

[0035] In the present invention, the terms "first", "second", etc. (if any) in the present invention and the drawings are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence.

[0036] Example 1

[0037] A transient process control method for a dual active bridge series resonant converter in discontinuous conduction mode, see Figure 1 , combined with Figure 2-Figure 10 , the transient process control method of the dual active bridge series resonant converter in the discontinuous conduction mode in this embodiment is described in detail.

[0038] The dual-active-bridge series resonant converter (DBSRC) includes a primary full-bridge H1 and a secondary full-bridge H2. H1 is connected to the primary side of the transformer via an LC resonant cavity, while H2 is connected to the secondary side of the transformer. In this embodiment, the DBSRC uses discontinuous conduction mode modulation for steady-state operation. The transient control method is as follows.

[0039] When f s <f r , Q k >Q k-1 And ΔQ k -Q k-1 <4C(V1-NV2), or when f s <f r , Q k k-1 And ΔQ k-1 -Q k <4CNV2, or when f s >f r and|Q k -Q k-1 When |>ΔQ, the method includes: in the current switching cycle, the duty cycle of the AC side voltage of H1 is controlled to be equal to D in the first and second half switching cycles respectively. c 、D k ; From the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k , this method is called FTCM. c for:

[0040] ​​​

[0041] Among them, D c is the first intermediate duty cycle, f s is the switching frequency, f r is the resonant frequency, Q k D k The charge of the down converter, Q k-1 D k-1 The charge of the down converter, D k is the duty cycle instruction of the current switching cycle, D k-1 is the duty cycle instruction of the previous switching cycle, ΔQ is the minimum charge difference of the converter during transient process, C is the resonant capacitance in the LC resonant cavity, N represents the turns ratio of the transformer, V1 is the input DC voltage, V2 is the load voltage, and F n is the frequency ratio, F n =f s / f r , M is the voltage conversion ratio.

[0042] When|Q k -Q k-1 When |≤ΔQ, it is considered that the converter has not undergone a sudden change in command and is in a stable operating state. The method also includes: directly updating the duty cycle of the AC side voltage of H1 to D k , this method is called DCM.

[0043] When|Q k -Q k-1 When |>ΔQ, it is considered that the duty cycle of the converter has a step change, that is, a transient event occurs, and the corresponding FTCM or IFTCM is executed, such as Figure 5 and Figure 7 As shown, the details are as follows.

[0044] (1) When f s <f r , Q k >Q k-1 And ΔQ k -Q k-1 <4C(V1-NV2), or when f s <f r , Q k k-1 And ΔQ k-1 -Q k <4CNV2, or when f s >f r and|Q k -Q k-1 When |>ΔQ, only half of the switching cycle is used for FTCM. The method includes: in the current switching cycle, the duty cycle of the AC side voltage of H1 is controlled to be equal to D in the first and second half switching cycles respectively. c 、D​​​k ; From the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

[0045] (2) When f s <f r , Q k >Q k-1 And 4C (V1-NV2) k -Q k-1 <8C(V1-NV2), using one switching cycle to perform IFTCM when the duty cycle suddenly increases, the method further includes: in the current switching cycle, the duty cycle of the AC side voltage of H1 is controlled to be equal to δ·(F n / 2), the second intermediate duty cycle D c1 , δ is the correction coefficient; from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k Preferably, 0.5<δ<1.

[0046] (3) When f s <f r , Q k >Q k-1 And 8C (V1-NV2) k -Q k-1 , perform the following sub-operation S11-sub-operation S12.

[0047] In sub-operation S11, in the current switching cycle, the duty cycle of the AC side voltage of H1 is controlled to be equal to δ·(F n / 2), and Q k-1 Updated to Q k-1 +8C(V1-NV2), δ is the correction coefficient.

[0048] In sub-operation S12, there are three cases:

[0049] ① If the Q k -Q k-1 <4C(V1-NV2), execute FTCM, in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1 to be equal to D c 、D k , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

[0050] ② If 4C (V1-NV2) after update k -Q k-1 ​​​<8C(V1-NV2), IFTCM is executed when the duty cycle suddenly increases. In the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled in the first and second half switching cycles respectively, which is equal to δ·(F n / 2), the second intermediate duty cycle D c1 , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

[0051] ③ If updated to 8C (V1-NV2) k -Q k-1 , the next switching cycle is the new current switching cycle, and the sub-operation S11 is executed again until Q k -Q k-1 <8C(V1-NV2) no longer returns to execute sub-operation S11. It should be noted that when returning to execute sub-operation S11 again, Q k-1 The latest Q k-1 , Q k Remain unchanged.

[0052] The second intermediate duty cycle D C1 for:

[0053]

[0054] (4) When f s <f r , Q k k-1 And 4CNV2 k-1 -Q k <8CNV2, a switching cycle is used to execute IFTCM when the duty cycle suddenly decreases. Specifically, in the current switching cycle, the duty cycle of the first and second half switching cycles respectively controls the H1 AC side voltage to be equal to 0, the third intermediate duty cycle D c2 ; From the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

[0055] (5) When f s <f r , Q k k-1 And 8CNV2 k-1 -Q k , perform the following sub-operation S21-sub-operation S22.

[0056] In sub-operation S21, in the current switching cycle, the duty cycle of the AC side voltage of H1 is controlled to be equal to 0, and Q k-1 Updated to Q k-1 -8CNV2.

[0057] ​​​​​In sub-operation S22, there are three cases:

[0058] ① If the Q k-1 -Q k <4CNV2, execute FTCM, in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1 to be equal to D c 、D k , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

[0059] ② If 4CNV2 after update k-1 -Q k <8CNV2, when the IFTCM is executed when the duty cycle is suddenly reduced, in the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to be equal to 0 in the first and second half switching cycles, and the third intermediate duty cycle D c2 , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

[0060] ③ If 8CNV2 after update k-1 -Q k , the next switching cycle is taken as the new current switching cycle, and the sub-operation S21 is returned to be executed again until Q k-1 -Q k <8CNV2 no longer returns to execute sub-operation S21. It should be noted that when returning to execute sub-operation S21 again, Q k-1 The latest Q k-1 , Q k Remain unchanged.

[0061] The third intermediate duty cycle D c2 for:

[0062]

[0063] In this embodiment, the following problems are considered in the transmission power adjustment process: the duty cycle of DBSRC changes from the current cycle D k-1 The step changes to the next switching cycle D k When the duty cycle changes, the transmission power cannot change suddenly. Instead, it requires a transient process to transition. The larger the duty cycle mutation range, the longer the transition time. On the other hand, in closed-loop control, although charge is used as a control variable to improve dynamic response speed, the trade-off between overshoot and adjustment time still needs to be considered during the transient process. It is precisely based on these issues that this method is proposed. By changing the duty cycle in the first half of the switching cycle after the transient occurs, the transient transition time is controlled within half a switching cycle.

[0064] ​​The following describes the design principle of the transient process control method of the dual active bridge series resonant converter in the discontinuous conduction mode according to an embodiment of the present invention.

[0065] See Figure 1 The dual active bridge series resonant converter includes: input DC power supply V1, primary full bridge H1, resonant inductor L, resonant capacitor C, high frequency isolation transformer T with a turns ratio of N:1, secondary full bridge H2, output capacitor C o 、Load R o , the voltage on the load is expressed as V2. The resonant inductor L and the resonant capacitor C form an LC resonant cavity; H1 includes switching tubes Q1-Q4; H2 includes switching tubes Q5-Q8. The AC side voltage of H1 is v p ; The AC side voltage of H2 is v s ;i L is the current flowing through the resonant inductor; u c is the voltage on the resonant capacitor; half the switching cycle of the converter is T hs , the switching period is T s ;ω s =2πf s is the switching angular frequency; ω r =2πf r is the resonant angular frequency; M = NV2 / V1 is the voltage conversion ratio; is the characteristic impedance of the resonant cavity.

[0066] The converter discontinuous conduction mode targeted by this method has a resonant tank waveform such as Figure 2 See Figure 2 , in the positive half cycle, the inductor current can be divided into three stages: the first stage, v p =V1, v s =NV2,i L Rising; second stage, v p =0, v s =NV2,i L Decline; third stage, i L After dropping to zero, intermittent occurs. Lx and U cx Respectively expressed in t x The inductor current value and capacitor voltage value at the moment are expressed as follows:

[0067]

[0068] See Figure 2 In steady state, during half a switching cycle, the total charge flowing through the inductor causes the capacitor voltage to change from U c0 Rise to U c2 .but:

[0069]

[0070] exist Figure 2 In the discontinuous conduction mode shown, the transmission power or charge of the DBSRC has a one-to-one correspondence with the duty cycle, so a sudden change in the transmission power instruction is equivalent to a step change in the charge.

[0071] See Figure 3 , shows the schematic diagram of FTCM, the duty cycle from D k-1 The step changes to D k , define t k is the starting time of the transient process, FTCM uses an intermediate duty cycle D c The introduction of i L and u c transient process.

[0072] See Figure 3 In this embodiment, the duty cycle is changed before the sudden change (the duty cycle is D k-1 ) is called the initial steady state, and the steady state reached after the mutation is called the final steady state (the duty cycle is D k ). The FTCM method takes the first positive half cycle v after the transient occurs p The duty cycle is used as the control duty cycle D c , let it act for half a switching cycle, and then directly transition to the duty cycle D in the final steady state k This method only introduces one control parameter d c .

[0073] Because the current is discontinuous in DCM, in each half switching cycle, i L will return to zero and will not affect the next switching cycle. Therefore, the initial and final values ​​of the current and current in each switching cycle are equal. k+0.5 Time u c When the voltage is equal to the voltage value at the corresponding moment in the final steady state, the rapid switching between the two steady-state working conditions can be completed.

[0074] Although the capacitor voltage cannot jump at the moment of duty cycle mutation, it can be adjusted by adjusting D c Change the transient trajectory of the capacitor voltage so that it is k -t k+0.5 ) transitions to the target value, as shown in formula (2), and a fast transition between two DCM operating conditions can be achieved within half a switching cycle. k+0.5 , the duty cycle has been updated to the duty cycle D at the final steady state k , the converter continues to operate stably under the new working conditions.

[0075] u C (θ k2 )=UC2 (D k ) (2)

[0076] Although transient processes can cause DC voltage fluctuations, the control time of FTCM and IFTCM is only one to two switching cycles, so parasitic resistance and DC voltage fluctuations within the switching cycle can be ignored in the derivation and analysis. k -t k+0.5 The control period of i can be divided into three intervals. L and u c Iterative calculation can get u c (θ k2 ), as shown in formula (3), where θ k1 =2D c π.

[0077]

[0078] Combining equations (1) and (3), we can obtain the first intermediate duty cycle D based on charge control: c :

[0079]

[0080] Combined with D c From the expression, we can know that we only need to know the charge Q in one switching cycle before and after the transient process. k and Q k-1 , you can use D c By implementing FTCM, the transient process can be ended after half a switching cycle, which greatly improves the dynamic performance of the converter in DCM.

[0081] Since the control range of FTCM is limited, the following are the f s <f r When the duty cycle increases suddenly (Q k >Q k-1 ) and duty cycle reduction (Q k k-1 ) situation, FTCM is improved to form IFTCM.

[0082] Figure 4 This is the control method of IFTCM when the duty cycle suddenly increases. s <f r Time T s >T r , at t k -t k+0.5 Inside, D c The limit value may be reached. To solve this problem, IFTCM requires that in the first control interval t0-t k1 Inside, the voltage on the resonant cavity is v​LC =V1-NV2, when its action is 0.5T r (ie t k1 -t0=0.5T r ), that is, D c =F n / 2, can i L (θ k1 ) resonates to zero, at this time θ k1 =θ k2 ,like Figure 4 The shaded area shows that during this half resonant cycle, the total charge Q transferred from the input is c That is the area of ​​half a sine wave, as shown in formula (4). c The capacitor voltage u c (θ k2 ) reaches U c2 (D k ), that is, Equation (5), which indicates that the transient process cannot end in half a switching cycle. Substituting Equation (1) into Equation (5), we can obtain the limiting conditions for FTCM to achieve the expected goal, as shown in Equation (6).

[0083]

[0084] Q c <C[U C2 (D k )-U C0 (D k-1 )] (5)

[0085] Q′-Q<4C[V1-NV2] (6)

[0086] When the difference in charge exceeds the limit of formula (6), IFTCM should be used. First, control the duty cycle D c =F n / 2, ensuring the maximum duty cycle in the first half of the switching cycle, reducing the difference in charge. Then, in the second half of the switching cycle (θ k3 -θ k1 ), and then introduce a new control duty cycle D c1 Due to D c =F n / 2, so IFTCM also has only one control variable. c1 To adjust the transient trajectory of the capacitor voltage so that at the end of a switching cycle (t k -t k+1 ) transitions to the target value, as shown in formula (7), and a fast transition between two DCM operating conditions can be achieved within one switching cycle. k+1 , the duty cycle has been updated to the duty cycle D at the final steady state k, the converter continues to operate stably under the new working conditions.

[0087] u C (θ k5 )=U C0 (D k ) (7)

[0088] The time domain method can be used to derive the control quantity D required by IFTCM when the duty cycle suddenly increases. c1 as follows:

[0089]

[0090] When D1=F n / 2, indicating that IFTCM has also reached its maximum adjustment limit. c1 is the area of ​​the sine wave in the negative half switching cycle, as follows:

[0091] Q c1 =-2C[-V1+NV2-u C (θ k3 )]=C[u C (θ k5 )-u C (θ k3 )] (8)

[0092] Combining equations (4) and (8), we can see that after IFTCM is adjusted to the maximum extent, k5 The charge Q(θ k5 ) and the initial charge Q k-1 Relationship:

[0093] Q(θ k5 )=-2Cu C (θ k5 )=Q k-1 +8C(V1-NV2)

[0094] If Q(θ k5 ) k , then IFTCM can realize the transient control of DBSRC under DCM modulation within one switching cycle. c =D c1 =F n / 2, that is, IFTCM is modulated with the maximum charge, and then Q(θ k5 ) replaces the initial Q k-1 , and then perform transient control again in the manner of FTCM or IFTCM. The above description of transient control when the duty cycle suddenly increases under DCM modulation can be fully summarized as follows: Figure 5 The iterative process shown.

[0095] ​It should be noted that in the DBSRC converter, although L m >>L, but in reality L m is not infinite, where L m is the excitation inductance. k -t k1 (t k2 ) period, due to v p =V1, v s =NV2, then the excitation inductance current i m At positive voltage v s According to KCL, as i m Gradually increases, when i m >i L When L2 <0. At this time, since Q5 is turned on, the reverse current i L2 After Q5 and D7, v s = 0, the excitation inductance is short-circuited. p Still maintain the positive voltage V1, the voltage V LC =v p -v s =V1>0. Therefore, at positive voltage V LC Under the effect of L It will increase again, causing the resonant tank LC and the excitation inductance to resonate together. L =i m , so the resonant inductor current cannot be completely discontinuous.

[0096] In order to solve the k -t k1 (t k2 ) may occur in LC and L m The problem of common resonance, achieving i L Completely intermittent, need to ensure i L2 Reverse and L m After being short-circuited, the voltage V LC ≤0. Because i L =i m Usually occurs in L Falling stage, if V LC is always less than or equal to 0, which can keep the downward trend until it decreases to zero, thus achieving discontinuity. L =i m Front V LC ≤0, the switch tube Q4 should be turned off in advance to ensure v p in i L =i m Return to zero level.

[0097] It is beneficial to the time domain method, combined with formula (0), i L and i m In t k -t k1 The expression of the stage is:

[0098]

[0099] Let i L (t) = i m (t) is used to solve the moment when Q4 is turned off in advance during the transient process. It can be seen that this equation is a transcendental equation. Therefore, in order to simplify the calculation, the sine function is expanded by the third order Taylor, and t2 = 0 is set. Finally, the solution is i L (t) = i m The approximate time t of (t) eq for:

[0100]

[0101] In fact, the time when the current is equal is slightly greater than t eq , so the t calculated by formula (10) eq This is the conservative time to turn off Q4 in advance. Turning off Q4 in advance according to this conservative time can also reduce the parasitic resistance and L in actual situations. m The impact of measurement error.

[0102] Definition eq The corresponding duty cycle is t eq / T s =D eq . Define t eq The ratio of half the resonant period is the coefficient δ. Then D eq =δ·(F n / 2). Due to i m The rising speed is less than i L , so they must intersect at i L Therefore, 0.5<δ<1. After the correction of IFTCM, although the charge of the positive half cycle in the transient process is the same as that before the correction, c There is an error, but in the DBSRC converter, L m Usually larger, the excitation current rises more slowly, so t eq The time from t2 is short, and i L will eventually decrease to zero. Therefore, the corrected Q c The error can be ignored. In the negative half cycle, the duty cycle Q c1 for transient control.

[0103] Figure 6This is the control method of IFTCM when the duty cycle suddenly decreases. First, the duty cycle in the first half of the switching cycle is minimized to reduce the difference with the target charge Q. k This requires D c =0, at this time Q c =2C[-NV2-U C0 (D k-1 )]. When Q c The capacitor voltage u c (θ k2 ) reaches U C2 (D k ), that is, |Q k >Q k-1 |>4CNV2, then introduce a new control duty cycle D in the second half of the switching cycle c2 , to achieve the purpose of ending the transient process within one switching cycle. The control quantity D during sudden reduction can be derived by using the time domain method c2 for:

[0104]

[0105] If D c2 = 0, then IFTCM also reaches the maximum adjustment limit, at this time the charge Q (θ k5 ) and the initial charge Q k-1 The relationship is: Q(θ k5 )=Q k-1 -8CNV2. Similar to the case of a sudden increase in duty cycle, if Q(θ k5 )<θ k , then IFTCM can be used to achieve transient control. Otherwise, it is only necessary to control the duty cycle D c =D c2 =0, then let Q(θ k5 ) replaces the initial D k-1 , and then perform transient control according to the FTCM or IFTCM method. In summary, the idea of ​​transient control when the duty cycle suddenly decreases can be described as Figure 7 The flowchart shown.

[0106] In order to verify the effectiveness of the proposed transient process control method in the discontinuous conduction mode of the dual active bridge series resonant converter in closed-loop control, the embodiment of the present invention is based on direct charge control, and the transfer charge of DBSRC is controlled by a proportional integral controller, thereby adjusting the duty cycle, which not only achieves the output voltage V o The overall block diagram of this principle is as follows: Figure 1 The DBSRC involved includes a direct charge quantity control unit, a transient process control unit, a pulse width generation unit and a main circuit.

[0107] When the direct charge quantity control unit controls the control quantity Q of the current cycle k The control amount Q of the previous beat k-1 If the error exceeds a given value, a transient event is considered to have occurred in the DBSRC, and transient control is implemented using the FTCM provided by the present invention. Otherwise, the converter is considered to be in steady-state operation, and DCM modulation is maintained. After DCM or transient control, the control duty cycle is determined and sent to the pulse width generation unit. This pulse width generation unit controls the on / off switching of the switches in the H1 and H2 bridge circuits, achieving rapid and stable control of the output voltage.

[0108] In summary, the design principle of transient control method is to add intermediate control duty cycle D c 、D c1 and D c2 , when the transmission power instruction changes suddenly, it is only necessary to c 、D c1 and D c2 By using the expression of FTCM or IFTCM modulation principle, the transient process can be controlled within 1-2 switching cycles, and the transient process of the inductor current and capacitor voltage can be smoother, which greatly improves the dynamic performance of the converter.

[0109] Set the circuit parameters of DBSRC as follows: V1 = 160V, V2 = 120V, f s =10kHz, N=1, L=214μH, C=0.613μF. Simulation was performed in MATLAB / Simulink according to the above parameters.

[0110] First, an uncontrolled open-loop simulation was performed, i.e., both sides of the converter were connected to a DC voltage source. The simulation results are shown in Figure 2. Figure 8 shown. Figure 8 The left picture in the middle is the effect picture under DCM, and the right picture is the control effect picture after using this method. k-1 =0.05, D k =0.1, after calculation, we know that the control quantity D required by FTCM c is 0.0867; in the middle two rows, D k-1 =0.2, D k =0.05, the control amount D required by FTCM c is 0.096; in the last two rows, D k-1 =0.05, D k =0.2, using IFTCM, let D c =F n / 2*0.8,D c1= 0.2297. It can be seen that under DCM, when the duty cycle suddenly changes, the DBSRC resonant cavity undergoes a relatively long transient process, and the peak value of the state variable during this transient process will experience shocks and offsets. After optimization using this method, the transient process can smoothly transition to a new steady state within basically one switching cycle.

[0111] Furthermore, a closed-loop simulation based on charge control was performed, where the converter output was connected to the load. o Switching between 80Ω and 40Ω, the results are as follows Figure 9 As shown, Figure 9 The left figure in the middle shows the simulation when switching from 80Ω to 40Ω, and the right figure shows the simulation when switching from 40Ω to 80Ω. It can be seen that due to the regulation of the PI controller, a certain overshoot will occur in the transient process during load switching, which is determined by the controller parameters.

[0112] Applying FTCM and IFTCM to closed-loop control, the control results are as follows Figure 10 As shown, Figure 10 The left-hand figure shows the control results when switching from 80Ω to 40Ω, while the right-hand figure shows the control results when switching from 40Ω to 80Ω. It can be seen that the implementation of transient control improves the response speed of closed-loop control and also smooths the transient processes of the output voltage, inductor current, and capacitor voltage, significantly improving the dynamic performance of closed-loop control.

[0113] Example 2

[0114] A transient process controller for a dual active bridge series resonant converter in discontinuous conduction mode is provided. The controller is configured to execute the transient process control method for the dual active bridge series resonant converter in discontinuous conduction mode. The related technical solutions are the same as those in the first embodiment and will not be described in detail here.

[0115] It will be easily understood by those skilled in the art that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A transient process control method for a dual-active-bridge series resonant converter in discontinuous conduction mode, wherein the dual-active-bridge series resonant converter comprises a primary full-bridge H1 and a secondary full-bridge H2, wherein H1 is connected to the primary side of a transformer via an LC resonant cavity, and H2 is connected to the secondary side of the transformer, wherein: When f s <f r , Q k >Q k-1 And ΔQ k -Q k-1 <4C(V1-NV2), or when f s <f r , Q k k-1 And ΔQ k-1 -Q k <4CNV2, or when f s >f r and|Q k -Q k-1 When |>ΔQ, the methods include:​​​ In the current switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1, which is equal to D c 、D k ; From the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k , D c for: Among them, D c is the first intermediate duty cycle, f s is the switching frequency, f r is the resonant frequency, Q k D k The charge of the down converter, Q k-1 D k-1 The charge of the down converter, D k is the duty cycle instruction of the current switching cycle, D k-1 is the duty cycle instruction of the previous switching cycle, ΔQ is the minimum charge difference of the converter during transient process, C is the resonant capacitance in the LC resonant cavity, N represents the turns ratio of the transformer, V1 is the input DC voltage, V2 is the load voltage, and F n is the frequency ratio, F n =f s / f r , M is the voltage conversion ratio.

2. The transient process control method in the discontinuous conduction mode of the dual active bridge series resonant converter according to claim 1, characterized in that: When f s <f r , Q k >Q k-1 And 4C (V1-NV2) k -Q k-1 When <8C (V1-NV2), the method also includes:​ In the current switching cycle, the duty ratio of the first and second half switching cycles respectively controlling the voltage on the AC side of H1 is equal to δ·(F n / 2), the second intermediate duty cycle D c1 , δ is the correction coefficient; From the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

3. The transient process control method in the discontinuous conduction mode of the dual active bridge series resonant converter according to claim 1, characterized in that: When f s <f r , Q k >Q k-1 And 8C (V1-NV2) k -Q k-1 When, the method further includes:​ S11, in the current switching cycle, the duty cycle of the AC side voltage of H1 is controlled to be equal to δ·(F n / 2), and Q k-1 Updated to Q k-1 +8C(V1-NV2), δ is the correction factor; S12, if Q is updated k -Q k-1 <4C(V1-NV2), in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1 to be equal to D c 、D k , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k ; If updated 4C (V1-NV2) k -Q k-1 <8C(V1-NV2), in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1, which is equal to δ·(F n / 2), the second intermediate duty cycle D c1 , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k ;​ If updated to 8C (V1-NV2) k -Q k-1 , the next switching cycle is taken as the new current switching cycle, and the process returns to S11 again until Q k -Q k-1 <8C(V1-NV2) no longer returns to execute the above S11.​ 4. The transient process control method in the discontinuous conduction mode of the dual active bridge series resonant converter according to claim 2 or 3, characterized in that: D c1 for:

5. The transient process control method in the discontinuous conduction mode of the dual active bridge series resonant converter according to claim 1, characterized in that: When f s <f r , Q k k-1 And 4CNV2 k-1 -Q k When <8CNV2, the method also includes:​​ In the current switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1 to be equal to 0, the third intermediate duty cycle D c2 ; From the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k .

6. The transient process control method in the discontinuous conduction mode of the dual active bridge series resonant converter according to claim 1, characterized in that: When f s <f r , Q k k-1 And 8CNV2 k-1 -Q k When, the method further includes:​​ S21, in the current switching cycle, controls the duty cycle of the AC side voltage of H1 to be equal to 0, and sets Q k-1 Updated to Q k-1 -8CNV2; S22, if Q is updated k-1 -Q k <4CNV2, in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1 to be equal to D c 、D k , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k ; If 4CNV2 is updated k-1 -Q k <8CNV2, in the next switching cycle, the duty cycle of the first and second half switching cycles respectively controls the voltage on the AC side of H1 to be equal to 0, the third intermediate duty cycle D c2 , from the next switching cycle, the duty cycle of the AC side voltage of H1 is controlled to return to D k ;​ If 8CNV2 is updated k-1 -Q k , the next switching cycle is taken as the new current switching cycle, and the process returns to S21 again until Q k-1 -Q k <8CNV2 no longer returns to execute S21.​ 7. The transient process control method in the discontinuous conduction mode of the dual active bridge series resonant converter according to claim 5 or 6, characterized in that: D c2 for:

8. The transient process control method in the discontinuous conduction mode of the dual active bridge series resonant converter according to claim 1, characterized in that: When|Q k -Q k-1 When |≤ΔQ, the method further includes: directly updating the duty cycle of the AC side voltage of H1 to D k .

9. A transient process controller for a dual active bridge series resonant converter in discontinuous conduction mode, characterized in that: The controller is used to execute the transient process control method in the discontinuous conduction mode of the dual active bridge series resonant converter according to any one of claims 1 to 8.

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