Hybrid rectifier bridge llc resonant converter with wide output voltage range and control method thereof

Abstract

The application discloses a hybrid rectifier bridge LLC resonant converter with wide output voltage range and a control method thereof, and belongs to the technical field of power generation, power transformation or power distribution. The converter comprises a primary side inverter network, a resonant network, a high-frequency transformer group and a secondary side hybrid rectifier network with bidirectional switches, and realizes super wide output voltage gain. On the control method, the primary side adopts phase shift control, and the secondary side adaptively switches the working modes of the full-bridge rectifier network and the voltage doubling rectifier network through a superimposed asymmetric PWM addition and phase shift control mode, realizes wide regulation voltage range, provides half cycle soft switching for the bidirectional switches, greatly reduces diode loss, realizes the optimization between low switching loss and low conduction loss, and realizes the free switching of high voltage and low voltage modes while retaining the original advantages of the LLC. The control method is simple and effective, can realize super wide output voltage range, and can also ensure the improvement of conversion efficiency.

Description

TECHNICAL FIELD

[0001] The application discloses a hybrid rectifier bridge LLC resonant converter with a wide output voltage range and a control method thereof, relates to power electronic technology, in particular to DC-DC power conversion and converter control technology, and belongs to the technical field of power generation, power transformation or power distribution. BACKGROUND

[0002] In the fields of new energy vehicles, spacecrafts, new energy power generation and data centers, the end voltage of a load has a wide range characteristic, and thus the power supply system needs to realize wide voltage range power conversion while ensuring high conversion efficiency in the whole range.

[0003] A power converter is a core device of a power supply system. The LLC resonant converter is widely welcomed due to small size, high efficiency and large power density. In order to widen the output voltage gain range, phase-shift control is widely applied. However, when the phase-shift angle is too large, one of the switch tubes of the primary side circuit of the LLC converter loses the soft switching characteristic, resulting in loss of efficiency. On this basis, a pair of bidirectional switches is added to change the secondary side structure into a good solution, and the secondary side variable structure rectifier bridge with the added bidirectional switches can be configured into a bridge rectifier and a voltage doubler rectifier in multiple modes, so as to cover a wider gain range in multiple modes with different gains. However, the added bidirectional switches increase the switching loss and the conduction loss, greatly affecting the efficiency of the converter. In order to reduce the loss, the existing research proposes to add a single switch tube to reduce the loss. However, in the high voltage mode, the single switch tube can only be in the voltage doubler rectifier mode, losing the advantage of flexible switching of the hybrid rectifier bridge, and can improve the efficiency of the converter, but cannot meet the requirement of widening the output voltage range. In order to reduce the conduction loss, the existing research proposes a PWM control method to reduce the conduction loss. However, the symmetrical PWM control needs to give priority to the output voltage control, so that the diode conduction loss can only be appropriately reduced, and the hard switching still brings about the switching loss, that is, the output voltage range is widened while the efficiency of the converter is reduced. Therefore, in the application of the secondary side variable structure rectifier bridge, it is a challenging problem to propose an adaptive, efficient, reliable and simple control method. SUMMARY

[0004] The application aims at solving the technical problem of the contradiction between the wide gain and high efficiency of the existing LLC converter, and achieving the application purpose of simplifying the control and improving the efficiency while widening the output voltage range.

[0005] The application adopts the following technical scheme to achieve the above application purpose:

[0006] The wide output voltage range hybrid rectifier bridge LLC resonant converter comprises a primary side half-bridge inverter network, a resonant network, a high-frequency transformer set and a secondary side hybrid rectifier network.

[0007] The primary side half-bridge inverter network comprises a first half-bridge inverter unit and a second half-bridge inverter unit, and the first half-bridge inverter unit and the second half-bridge inverter unit are connected in parallel with an input DC power supply; the first half-bridge inverter unit is composed of a first switch tube and a second switch tube connected in series; and the second half-bridge inverter unit is composed of a third switch tube and a fourth switch tube connected in series.

[0008] The resonant network comprises a first resonant tank and a second resonant tank, and the first resonant tank is connected to an output end of the first half-bridge inverter unit, and the second resonant tank is connected to an output end of the second half-bridge inverter unit.

[0009] The high-frequency transformer set comprises an anti-phase high-frequency transformer and a positive-phase high-frequency transformer, and the primary winding of the anti-phase high-frequency transformer is connected in parallel with a magnetizing inductance in the first resonant tank, and the primary winding of the positive-phase high-frequency transformer is connected in parallel with a magnetizing inductance in the second resonant tank; and the secondary winding of the anti-phase high-frequency transformer and the secondary winding of the positive-phase high-frequency transformer are connected in series in positive polarity.

[0010] The secondary side hybrid rectifier network comprises a full-bridge rectifier network and a pair of bidirectional switches, and the pair of bidirectional switches comprises a fifth switch tube and a sixth switch tube connected in common source; the positive polarity end of the anti-phase high-frequency transformer and the positive polarity end of the positive-phase high-frequency transformer connected in series in positive polarity are connected to a first bridge arm midpoint of the full-bridge rectifier network; the negative polarity end of the anti-phase high-frequency transformer and the positive polarity end of the positive-phase high-frequency transformer connected in series in positive polarity and the drain of the fifth switch tube are connected to a second bridge arm midpoint of the full-bridge rectifier network; and the drain of the sixth switch tube is connected to a filter output unit.

[0011] As a further optimization scheme of the wide output voltage range hybrid rectifier bridge LLC resonant converter, in the first half-bridge inverter unit, the drain of the first switch tube is connected to the positive pole of the input DC power supply, the drain of the second switch tube is connected to the source of the first switch tube, the source of the second switch tube is connected to the negative pole of the input DC power supply, and the drain and the source of the second switch tube constitute an output end of the first half-bridge inverter unit; and in the second half-bridge inverter unit, the drain of the third switch tube is connected to the positive pole of the input DC power supply, the drain of the fourth switch tube is connected to the source of the third switch tube, the source of the fourth switch tube is connected to the negative pole of the input DC power supply, and the drain and the source of the fourth switch tube constitute an output end of the second half-bridge inverter unit.

[0012] As a further optimization scheme of the hybrid rectifier bridge LLC resonant converter with wide output voltage range, the first resonant tank comprises: a first resonant inductor, a first excitation inductor and a first resonant capacitor, one end of the first resonant inductor is connected with the source electrode of the first switch tube and the drain electrode of the second switch tube, the other end of the first resonant inductor is connected with one end of the first excitation inductor, the other end of the first excitation inductor is connected with the positive electrode of the first resonant capacitor, the negative electrode of the first resonant capacitor is connected with the source electrode of the second switch tube, and the original excitation inductor of the anti-phase high-frequency transformer serves as the first excitation inductor; the second resonant tank comprises: a second resonant inductor, a second excitation inductor and a second resonant capacitor, one end of the second resonant inductor is connected with the source electrode of the third switch tube and the drain electrode of the fourth switch tube, the other end of the second resonant inductor is connected with one end of the second excitation inductor, the other end of the second excitation inductor is connected with the positive electrode of the second resonant capacitor, and the negative electrode of the second resonant capacitor is connected with the source electrode of the fourth switch tube, and the original excitation inductor of the positive-phase high-frequency transformer serves as the second excitation inductor.

[0013] As a further optimization scheme of the hybrid rectifier bridge LLC resonant converter with wide output voltage range, the first end of the original winding of the anti-phase high-frequency transformer is connected with the other end of the first resonant inductor and one end of the first excitation inductor, and the second end of the original winding of the anti-phase high-frequency transformer is connected with the positive electrode of the first resonant capacitor; the first end of the original winding of the positive-phase high-frequency transformer is connected with the other end of the second resonant inductor and one end of the second excitation inductor, and the second end of the original winding of the positive-phase high-frequency transformer is connected with the positive electrode of the second resonant capacitor.

[0014] As a further optimization scheme of the hybrid rectifier bridge LLC resonant converter with wide output voltage range, the full-bridge rectifier network comprises: first to fourth diodes, the cathode of the first diode is connected with the cathode of the third diode, the cathode of the second diode is connected with the anode of the first diode to form a first bridge arm midpoint of the full-bridge rectifier network, the cathode of the fourth diode is connected with the anode of the third diode to form a second bridge arm midpoint of the full-bridge rectifier network, and the anode of the second diode is connected with the anode of the fourth diode; the first end of the secondary winding of the anti-phase high-frequency transformer is connected with the first bridge arm midpoint of the full-bridge rectifier network as a positive polarity end, the first end of the secondary winding of the positive-phase high-frequency transformer is connected with the second end of the secondary winding of the anti-phase high-frequency transformer, the second end of the secondary winding of the positive-phase high-frequency transformer is connected with the second bridge arm midpoint of the full-bridge rectifier network as a negative polarity end, the first end of the secondary winding of the anti-phase high-frequency transformer and the second end of the original winding of the anti-phase high-frequency transformer are homonymous ends, and the first end of the secondary winding of the positive-phase high-frequency transformer and the first end of the original winding of the positive-phase high-frequency transformer are homonymous ends.

[0015] A control method of a hybrid rectifier bridge LLC resonant converter with wide output voltage range,

[0016] The wide output voltage range hybrid rectifier bridge LLC resonant converter and its control method include two operating modes, each employing a different control approach. In low-voltage mode, primary-side phase-shift control is used; in high-voltage mode, primary-side phase-shift control is used, while the secondary side employs a superimposed asymmetric PWM and phase-shift control strategy. This achieves ultra-wide output voltage regulation while improving conversion efficiency. The specific steps are as follows:

[0017] Step 1: The control system controls the sampling chip to collect the output voltage and secondary current of the converter, and determines whether the desired voltage belongs to high voltage mode or low voltage mode. If the desired voltage is within the output voltage adjustment range of high voltage mode, proceed to step 3; if the desired voltage is within the output voltage adjustment range of low voltage mode, proceed to step 2.

[0018] Step 2: The difference between the converter output voltage acquired in Step 1 and the desired voltage is adjusted by the voltage regulator to obtain the required phase shift angle α. The control signals of the first and second switches are phase-shifted to obtain the control signals of the third and fourth switches. The required control signals of the first, second, third, and fourth switches are output to adjust the converter output voltage. The fifth and sixth switches are not controlled. The converter operation mode is similar to that of the phase-shifted full bridge, and the normalized voltage gain is 0 to 1.

[0019] Step 3: Enter high voltage mode, perform 180° phase shift adjustment on the control signals of the first and second switching transistors to obtain the control signals of the third and fourth switching transistors, and output the required control signals of the first, second, third, and fourth switching transistors.

[0020] Step 4: The difference between the converter output voltage acquired in Step 1 and the desired voltage is sent to the voltage regulator for adjustment, and the first duty cycle output is D. y1 The first PWM wave control waveform of the fifth switching transistor;

[0021] Step 5, similar to step 4, yields the first duty cycle as D. y1 The PWM wave is then adjusted by 180° phase shift to obtain the first PWM wave control waveform of the sixth switch. The fifth and sixth switches driven by the first PWM wave of the fifth switch and the first PWM wave of the sixth switch are turned on, and the body diodes are naturally turned on to achieve wide output voltage regulation.

[0022] Step 6, according to the formula Δt1 is obtained through D y1 D y2 The second duty cycle, calculated from the relationship with Δt1, is D. y2The PWM wave of the fifth switch tube is obtained by phase-shifting 180 degrees, and the current originally flowing through the body diode of the fifth switch tube is changed to flow through the channel of the switch tube, thereby reducing the conduction loss of the diode, and the time difference of Δt1 provides conditions for realizing zero-voltage turn-on of the fifth switch tube;

[0023] In step 7, similarly, a second duty ratio D y2 The PWM wave of the sixth switch tube is obtained, so as to reduce the conduction loss of the diode of the sixth switch tube and realize zero-voltage turn-on of the sixth switch tube, thereby reducing the conduction loss and switching loss.

[0024] In step 8, the secondary side current i sec is sent to the current zero-crossing detection circuit to obtain the zero-crossing time t zero , so as to obtain the angle θ of the current zero-crossing point lagging behind the primary side voltage, for phase-shift control correction.

[0025] In step 9, the first PWM wave and the second PWM wave control signals of the fifth switch tube are superimposed, and the obtained signal is phase-shifted by an angle θ to obtain the control signal of the fifth switch tube in the final high-voltage mode.

[0026] In step 10, similarly, the first PWM wave and the second PWM wave control signals of the sixth switch tube are superimposed, and the obtained signal is phase-shifted by an angle θ to obtain the final control signal of the sixth switch tube.

[0027] The application adopts the above technical scheme to solve the technical problem of the contradiction between wide gain and high efficiency of the existing LLC converter, and has the following beneficial effects:

[0028] (1) The wide output voltage range hybrid rectifier bridge LLC resonant converter proposed in the application realizes super-wide output voltage gain on the basis of the converter topology architecture of parallel connection of the primary side and series connection of the secondary side, and transmits the energy generated by the parallel connection of the primary side to the series connection of the secondary side through a positive phase high-frequency transformer and a negative phase high-frequency transformer.

[0029] (2) Compared with the existing high-efficiency LLC resonant converter technology, the control method of the wide output voltage range hybrid rectifier bridge LLC resonant converter proposed in the application combines the primary side phase-shift control strategy and the secondary side superimposed asymmetric PWM and phase-shift control strategy, can reduce the conduction time of the diode and realize soft switching of half a period of the bidirectional switch under the premise of super-wide output voltage gain, realizes the optimization balance between reducing switching loss and conduction loss, suppresses voltage spikes, and is simple and convenient to control.

[0030] (3) Compared with the existing wide gain LLC resonant converter technology, the wide output voltage range hybrid rectifier bridge LLC resonant converter control method provided by the application does not need frequency modulation and additional excessive magnetic elements, has less topological devices and simple structure; meanwhile, the converter always works at a resonant point, the magnetic element is easy to design; only the software is used to realize the working mode switching of two working modes, so that the voltage gain can be greatly widened, and the realization method is simple and reliable. BRIEF DESCRIPTION OF DRAWINGS

[0031] Figure 1 is the topological structure diagram of the wide output voltage range hybrid rectifier bridge LLC resonant converter provided by the application.

[0032] Fig. 2(a) and Fig. 2(b) are the switching timing waveform diagrams of the converter provided by the application in a low-voltage mode and a high-voltage mode.

[0033] Figure 3 is the key waveform diagram under the superimposed asymmetric PWM and phase shift control provided by the application.

[0034] Figure 4 is the equivalent circuit diagram under the superimposed asymmetric PWM and phase shift control provided by the application.

[0035] Fig. 5(a) is a relationship curve diagram of the voltage gain and the phase shift angle a of the converter provided by the application in the low-voltage mode, and Fig. 5(b) is a relationship curve diagram of the voltage gain and the duty cycle D of the converter provided by the application in the high-voltage mode. y1

[0036] Fig. 6(a) is the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube, the control signal S6 of the first switch tube and the voltage waveform v ds6 of an embodiment of the application under the working condition of the output voltage of 300V and the output power of 3000W, and Fig. 6(b) is the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube, the secondary side voltage V sec and the resonant current i Lr1 , the excitation current waveform i Lm1 of an embodiment of the application under the working condition of the output voltage of 300V and the output power of 3000W.

[0037] Fig. 7(a) is the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube, the control signal S6 of the first switch tube and the voltage waveform v ds6 ​Fig. 7(b) is the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube and the secondary side voltage V sec and the resonant current i Lr1 , the excitation current waveform i Lm1 .

[0038] Fig. 7(b) is the control signal S1 of the first switch tube, the control signal S5 of the fifth switch tube and the secondary side voltage V in , input DC power supply, Q1, first switch tube, Q2, second switch tube, L r1 , first resonant inductor, C r1 , first resonant capacitor, L m1 , first excitation inductor, Q3, third switch tube, Q4, fourth switch tube, L r2 , second resonant inductor, C r2 , second resonant capacitor, L m2 , second excitation inductor, T1, anti-phase high-frequency transformer, T2, positive-phase high-frequency transformer, D1-D4, first to fourth diodes, Q5, fifth switch tube, Q6, sixth switch tube, C o1 , first filter capacitor, C o2 , second filter capacitor, R o , load resistor, V o , output DC voltage. DETAILED DESCRIPTION

[0039] The technical solutions of the present application will be described in detail below with reference to the drawings.

[0040] As Figure 1 shown is the topology structure of the mixed rectifier bridge LLC resonant converter with wide output voltage range, including primary side half-bridge inverter network, resonant network, high-frequency transformer group and secondary side mixed rectifier network. The primary side half-bridge inverter network is composed of two half-bridge inverter units sharing the secondary side mixed rectifier network, each half-bridge inverter unit is connected in parallel with the input DC power supply V in , each output end of the half-bridge inverter unit is connected with a resonant tank, each resonant tank is composed of a resonant inductor, an excitation inductor and a resonant capacitor in series, the anti-phase high-frequency transformer primary excitation inductor and the positive-phase high-frequency transformer primary excitation inductor are used as the excitation inductor to participate in the resonant tank operation, due to the different winding methods of the two transformers, the energy polarities transmitted to the secondary side through the transformers are different; the secondary side mixed rectifier network is composed of a full-bridge rectifier network and a pair of common-source back-to-back bidirectional switches in series, the anti-phase high-frequency transformer secondary winding and the positive-phase high-frequency transformer secondary winding are connected in series and then connected between the middle of the bridge arms of the full-bridge rectifier network; the special structure of the original parallel anti-series can realize ultra-wide range voltage output in low voltage mode.

[0041] Figure 1 The specific connection relationship of the shown conversion topology is as follows: the first switch tube Q1 and the second switch tube Q2 connected in series to form a first half-bridge inverter network, the drain electrode of the first switch tube Q1 is connected with the positive pole of the input DC power supply V in , the source electrode of the second switch tube Q2 is connected with the negative pole of the input DC power supply V in , the first resonant inductor L r1 has one end connected with the source electrode of the first switch tube Q1 and the drain electrode of the second switch tube Q2, the first resonant inductor L r1 has the other end connected with the first excitation inductor L m1 , the first excitation inductor L m1 has one end connected with the first resonant capacitor C r1 , the first resonant capacitor C r1 has the other end connected with the positive pole of the input DC power supply V r1 , the source electrode of the second switch tube Q2 is connected with the first resonant capacitor C r1 , and the connection point of the negative pole is b1; the third switch tube Q3 and the fourth switch tube Q4 connected in series to form a second half-bridge inverter network, the third switch tube Q3 is connected with the positive pole of the input DC power supply V in , the source electrode of the fourth switch tube Q4 is connected with the negative pole of the input DC power supply V in , the second resonant inductor L r2 has one end connected with the source electrode of the third switch tube Q3 and the drain electrode of the fourth switch tube Q4, the second resonant inductor L r2 has the other end connected with the second excitation inductor L m2 , the second excitation inductor L m2 has one end connected with the second resonant capacitor C r2 , the second resonant capacitor C r2 has the other end connected with the positive pole of the input DC power supply V r2 , the source electrode of the fourth switch tube Q4 is connected with the second resonant capacitor C r2 , and the connection point of the negative pole is b2; the primary winding of the anti-phase high-frequency transformer T1 is connected in parallel with the first excitation inductor L m1 , the primary winding of the positive-phase high-frequency transformer T2 is connected in parallel with the second excitation inductor L m2 , the first excitation inductor L m1 is realized by the primary excitation inductor of the anti-phase high-frequency transformer, and the second excitation inductor L m2The rectifier is achieved through the primary excitation inductance of the positive-phase high-frequency transformer. The full-bridge rectifier network consists of diodes D1 to D4. Diodes D1 and D2 are connected in series to form one rectifier bridge arm, and diodes D3 and D4 are connected in series to form another rectifier bridge arm. The first end of the secondary winding of the inverting high-frequency transformer T1 is connected to one end of the anode of diode D1 and the cathode of diode D2. The first end of the secondary winding of the inverting high-frequency transformer T1 is connected to the first resonant capacitor C. r1 The second terminals of the primary windings of the positive-phase anti-phase high-frequency transformer T1 are identical terminals. The first terminal of the secondary winding of the positive-phase high-frequency transformer T2 is connected to the second terminal of the secondary winding of the anti-phase high-frequency transformer T1. The first terminal of the secondary winding of the positive-phase high-frequency transformer T2 is connected to the second resonant inductor L. r2 The first terminals of the primary winding of the positive-phase high-frequency transformer T2 are identical terminals. The first terminal of the secondary winding of the positive-phase high-frequency transformer T2 is connected to the anode of the third diode D3 and the cathode of the fourth diode D4. A pair of common-source back-to-back bidirectional switches are composed of the fifth switch Q5 and the sixth switch Q6 connected in series. The drain of the fifth switch Q5 is connected to the anode of the third diode D3 and the cathode of the fourth diode D4. The source of the fifth switch Q5 is connected to the source of the sixth switch Q6. The drain of the sixth switch Q6 is connected to the first filter capacitor C. o1 First and second filter capacitors C o2 One terminal is connected, the first filter capacitor C o1 The other pole, the second filter capacitor C o2 A load resistor R is connected in parallel between the other two poles. o The two resonant slots have the same parameters, namely L. r1 =L r2 C r1 =C r2 ,L m1 =L m2 .

[0042] against Figure 1 The converter shown can reduce conduction losses while achieving ultra-wide range voltage output in high-voltage mode by adjusting the duty cycle and phase of a pair of variable structure rectifier bidirectional switches on the secondary side.

[0043] Figures 2(a) and 2(b) show the switching timing waveforms of the converter proposed in this invention in low-voltage and high-voltage operating modes. The low-voltage mode includes one degree of freedom: the phase difference between the turn-on times of switching transistors Q1 and Q3, which is defined here as the phase shift angle α, used to achieve smooth control of the voltage range from 0 to 1. The high-voltage mode includes three degrees of freedom: the duty cycle D of control signals S5' and S6'. y1 The duty cycle D of control signals S5” and S6” y2 And the phase shift angle θ between the turn-off times of switches Q5 and Q6 and the turn-off time of switch Q1, i.e., the secondary current isec The angle of the primary side voltage after the zero crossing point, the voltage between a1 and b1, is θ = ω0·Δt2. Δt1 must satisfy the soft switching critical condition of the switch tubes Q5 and Q6, that is, Δt1 must satisfy Δt2 is equal to the secondary side current i sec The natural zero time length. By adjusting D y1 to control the output voltage, control D y2 to realize the optimization of conduction loss and switching loss, and the phase shift angle θ is used for phase correction, so as to realize the peak suppression of the secondary side voltage. Since the control signal of the switch tube Q5 is formed by superposition of S5' and S5'', and the control signal of the switch tube Q6 is formed by superposition of S6' and S6'', and the conduction duty ratios of S5' and S5'' are different, and the conduction duty ratios of S6' and S6'' are different, and the phase shift control correction is used, the control mode in the high voltage mode is called superimposed asymmetric PWM and phase shift control method.

[0044] Figure 3 The key waveforms under the superimposed asymmetric PWM and phase shift control of the application are shown, the control waveforms of each switch tube and important voltage and current waveforms are given, the switch tubes Q1 and Q2 are complementarily turned on with a dead zone, the control signals of Q1 and Q2 are phase shifted by 180° to obtain the control signals of Q3 and Q4, and the two half-bridge networks transfer energy to the secondary side through the resonant network. The control signals of the switch tubes Q5 and Q6 are consistent with those in Fig. 2(b), for the convenience of discussion, the variables are unified as angles for explanation. In the π(1-D y1 ) angle, the resonant current is equal to the excitation current, and then the two-element resonance continues, the current transferred to the secondary side naturally commutates, the switch tubes Q5 and Q6 are not turned on, the bidirectional switch is turned off, and the working mode of the hybrid rectifier bridge is self-adapted to full-bridge rectification, V sec =-V o In the ω0·Δt1 angle, the switch tube Q5 is turned on, and the switch tube Q6 is still in the off state. Since the current i sec transferred from the primary side to the secondary side is negative, this negative current discharges the junction capacitor of the switch tube Q6 and then flows through the body diode of the switch tube Q6, forming a voltage doubling network, and the working mode of the hybrid rectifier bridge is changed to voltage doubling rectification, V sec =-1 / 2V o In the πD y2 angle, the switch tubes Q5 and Q6 are both in the on state, and the hybrid rectifier bridge still works in the voltage doubling rectification mode, V sec =-1 / 2V o . By using phase shift control correction, the turn-off time of the switch tubes Q5 and Q6 is adjusted to be consistent with the secondary side current i secThe commutation moment is same, and the secondary voltage peak is effectively inhibited. According to the waveform of V sec It can be seen from the waveform that the output voltage can be adjusted by changing the duty ratio and phase shift angle of the secondary superimposed asymmetric PWM wave, and then changing the time of the hybrid rectifier network mode change.

[0045] Figure 4 The control circuit of the phase shift control and superimposed asymmetric PWM and phase shift control strategy is shown, and the control mode is divided into two modes: high voltage mode and low voltage mode.

[0046] When the given voltage V o_ref is less than 300V, the converter will be in low voltage mode, and the phase shift control strategy is used for the primary side, which is simple to control. At this time, the output voltage V o is compared with the given voltage V o_ref , and the difference is sent to the voltage regulator. The output value of the voltage regulator is obtained by the carrier phase shift controller to obtain the phase shift angle α, that is, the obtained carrier V saw2 phase lags behind the basic digital carrier V saw by an angle α. The switching tubes Q3 and Q4 lag behind the switching tubes Q1 and Q2, so that a part of the power is not transmitted to the secondary side, thereby realizing super wide output voltage regulation.

[0047] When the given voltage V o_ref is not less than 300V, the converter will be in high voltage mode, and the phase shift control strategy is used for the primary side, and the superimposed asymmetric PWM and phase shift control strategy is used for the secondary side. The phase shift control strategy used for the primary side in high voltage mode is that the switching tubes Q3 and Q4 lag behind the upper switching tubes Q1 and Q2 by a fixed angle of 180°, and the two switching tubes provide energy to the secondary side in staggered manner. In combination with FIG. 2(a) and FIG. 2(b), under the superimposed asymmetric PWM and phase shift control strategy, the moment when the secondary side current is zero is the turn-off moment of Q5 and Q6, so a zero-crossing detection circuit can be added to control the turn-off of Q5 and Q6, and the voltage peak of the secondary side voltage can be inhibited. The turn-on moment of Q5 and Q6 is adjusted according to the output voltage V o , and the difference between the output voltage V o and the given voltage V o_ref is sent to the voltage regulator. After comparison with the carrier V saw , the duty ratio D y1control signals S5' and S6', the control signal S6' lags behind the control signal S5' by 180°, Q5 and Q6 under the control of the control signals S5' and S6' can realize adaptive working mode switching of full-bridge rectification and voltage doubling rectification at the secondary side, and super wide range output voltage regulation can be realized by controlling the conduction of Q5 and Q6 and the natural freewheeling conduction of the body diode of Q5 and Q6 through S5' and S6'. Since the positive current flows through the channel of Q6 and the body diode of Q5, and the negative current flows through the channel of Q5 and the body diode of Q6, the conduction of the diode brings great conduction loss; the junction capacitance discharge time Δt1 of Q5 and Q6 is sent into a multiplier to obtain the duty cycle D y1 of the asymmetric PWM wave S5' and S6' after subtraction y2 , and the phase-shifted 180° PWM wave S6" is obtained to replace the natural freewheeling conduction function of the body diode, thereby reducing the conduction loss, and the time difference Δt1 reduces the switching loss; the secondary side current i sec collected in step 1 is sent into a current zero-crossing detection circuit to obtain the zero-crossing time t zero , so as to obtain the angle θ at which the current zero-crossing point lags behind the primary side voltage, which is used for phase-shifted control correction; the asymmetric PWM waves S5' and S5" are superimposed to obtain the final control signal of the switch Q5 after phase-shifted control correction at an angle θ, and the asymmetric PWM waves S6' and S6" are superimposed to obtain the final control signal of the switch Q6 after phase-shifted control correction at an angle θ, so as to realize wide range change of the output voltage, reduce the conduction loss and switching loss, and suppress voltage spikes.

[0048] When the converter is in the low voltage mode, the relationship between the phase shift angle α and the voltage gain G satisfies the following formula:

[0049]

[0050] wherein Q is the quality factor;

[0051] When the converter is in the high voltage mode, the superimposed asymmetric PWM plus phase-shifted control mode is adopted, the duty cycle D y1 , D y2 and Δt1 satisfy the following relationship:

[0052]

[0053] wherein C eq is the equivalent junction capacitance of the switch, and ω0 is the resonant angular frequency.

[0054] The turn-off time of S5' and S6" is the same as the turn-off time of S1, and the turn-off time of S6' and S5" is the same as the turn-on time of S1. The zero-crossing time t zero of the secondary side current is obtained by using the zero-crossing detection circuit, and t zeroThe difference Δt2 between the off time or the on time of S1 is obtained, and the phase shift angle is θ=ω0·Δt2.

[0055] The control method combines phase shift control and superimposed asymmetric PWM and phase shift control mode, controls the phase shift angle on the primary side, controls the duty cycle and phase of the fifth switch Q5 and the sixth switch Q6 on the secondary side, and realizes high-efficiency voltage regulation. When the converter works in the low-voltage regulation mode, the control signals of Q1 and Q2 are phase-shifted to obtain the control signals of Q3 and Q4, and the fifth switch Q5 and the sixth switch Q6 are not controlled, and the working mode is similar to the phase-shift full-bridge, and the normalized voltage gain is 0-1. When the converter works in the high-voltage regulation mode, the control signals of Q1 and Q2 are phase-shifted by 180° to obtain the control signals of Q3 and Q4, and then the duty cycle of the fifth switch Q5 and the sixth switch Q6 is adjusted and the phase is corrected, when the duty cycle is 0, the normalized gain is 1, and when the duty cycle is 1, the normalized gain is 2. The control method can suppress voltage spikes and realize continuous regulation of the normalized gain in the range of 1-2 without adjusting the frequency by adjusting the duty cycle and phase of the superimposed asymmetric PWM.

[0056] Fig. 5(a) is a relationship curve of the voltage gain and the phase shift angle α of the converter in the low-voltage mode, and Fig. 5(b) is a relationship curve of the voltage gain and the duty cycle D of the converter in the high-voltage mode. y1 In the low-voltage mode, it is found that the time-domain equation is consistent with that of the phase-shift control full-bridge LLC, and the drawing analysis can be performed by using it to obtain Fig. 5(a); in the high-voltage mode, the time-domain equation is listed according to the key waveform and the equivalent circuit, and the drawing is obtained by solving it by using MATLAB software to obtain Fig. 5(b), and it is found that the obtained curve has weak load dependence, and is more conducive to stable voltage range regulation. The switching control between the low-voltage mode and the high-voltage mode greatly widens the output voltage range. Figure 3

[0057] The simulation example of the application is given below, and the main parameters of the converter are as follows: the input voltage is 380V, the output voltage is 5-500V, and the transformer ratio is 38:30.

[0058] Fig. 6(a) shows the control signals S1, S5 and S6 of the first switch, the fifth switch and the sixth switch and the voltage waveforms v ds6 of the first switch, the fifth switch and the sixth switch under the working conditions of the output voltage of 300V and the output power of 3000W in one embodiment of the application, and the duty cycle D y1 ​Small, the sixth switch tube Q6 to achieve half cycle soft switching; Figure 6 (b) shows an embodiment of the present application in the output voltage is 300V, the output power is 3000W under the working conditions of the first switch tube control signal S1, the fifth switch tube control signal S5 and the secondary side voltage V sec And the resonant current i Lr1 , excitation current waveform i Lm1 , the phase-shift control correction Δt2 to achieve secondary side voltage peak suppression, waveform and theoretical analysis is completely consistent.

[0059] Figure 7 (a) shows an embodiment of the present application in the output voltage is 500V, the output power is 3000W under the working conditions of the first switch tube control signal S1, the fifth switch tube control signal S5 and the first switch tube control signal S6 and its voltage waveform v ds6 , the duty cycle D y1 Comparatively large, consistent with the voltage gain curve analysis, the sixth switch tube Q6 to achieve half cycle soft switching, while optimizing the on loss; Figure 7 (b) shows an embodiment of the present application in the output voltage is 500V, the output power is 3000W under the working conditions of the first switch tube control signal S1, the fifth switch tube control signal S5 and the secondary side voltage V sec And the resonant current i Lr1 , excitation current waveform i Lm1 , the phase-shift control correction Δt2 to achieve secondary side voltage peak suppression, waveform and theoretical analysis is completely consistent.

[0060] The above examples are only to illustrate the technical idea of the present application, can not be limited by the protection scope of the present application, according to the technical idea of the present application, on the basis of technical solutions made any changes, all fall within the scope of the present application.

Claims

1. Control method for a hybrid rectifier bridge LLC resonant converter with a wide output voltage range. The wide output voltage range hybrid rectifier bridge LLC resonant converter includes: The primary half-bridge inverter network includes a first half-bridge inverter unit and a second half-bridge inverter unit. The first and second half-bridge inverter units are connected in parallel with the input DC power supply. The first half-bridge inverter unit consists of a first switch and a second switch connected in series, and the second half-bridge inverter unit consists of a third switch and a fourth switch connected in series. The resonant network includes a first resonant slot and a second resonant slot. The first resonant slot is connected to the output terminal of the first half-bridge inverter unit, and the second resonant slot is connected to the output terminal of the second half-bridge inverter unit. A high-frequency transformer bank includes a reverse-phase high-frequency transformer and a forward-phase high-frequency transformer. The primary winding of the reverse-phase high-frequency transformer is connected in parallel with the magnetizing inductor in the first resonant slot. The primary winding of the forward-phase high-frequency transformer is connected in parallel with the magnetizing inductor in the second resonant slot. The secondary winding of the reverse-phase high-frequency transformer is connected in forward series with the secondary winding of the forward-phase high-frequency transformer. The secondary-side hybrid rectifier network includes a full-bridge rectifier network and a pair of bidirectional switches. The pair of bidirectional switches includes a fifth switch and a sixth switch connected by a common source. The positive terminal of the secondary winding of the inverting high-frequency transformer connected in series with the secondary winding of the positive high-frequency transformer is connected to the midpoint of the first bridge arm of the full-bridge rectifier network. The negative terminal of the secondary winding of the inverting high-frequency transformer connected in series with the secondary winding of the positive high-frequency transformer and the drain of the fifth switch are connected to the midpoint of the second bridge arm of the full-bridge rectifier network. The drain of the sixth switch is connected to the filter output unit. The control method is characterized in that: When the converter operates in low-voltage mode, a phase-shift control strategy is adopted on the primary side. When the converter operates in high-voltage mode, a phase-shift control strategy is adopted on the primary side, and a superimposed asymmetric PWM and phase-shift control strategy is adopted on the secondary side. Specifically, the superimposed asymmetric PWM and phase-shift control strategy on the secondary side is as follows: After adjusting the voltage difference between the sampled output voltage value and the desired voltage, the first duty cycle is obtained as follows: D y1 The PWM wave, denoted as the first duty cycle as D y1 The PWM wave is the first PWM wave of the fifth switching transistor. By performing a 180° phase shift adjustment on the first PWM wave of the fifth switching transistor, the first PWM wave of the sixth switching transistor is obtained. Based on the critical conditions for soft switching of the fifth and sixth switching transistors and the first duty cycle D y1 Obtain the second duty cycle as D y2 The PWM wave, denoted as the second duty cycle as D y2 The PWM wave is the second PWM wave of the fifth switching transistor. By performing a 180° phase shift adjustment on the second PWM wave of the fifth switching transistor, the second PWM wave of the sixth switching transistor is obtained. Based on the zero-crossing time of the secondary current of the converter, calculate the angle by which the zero-crossing point of the secondary current lags behind the primary voltage. θ , Based on the angle by which the secondary current lags behind the primary voltage at its zero-crossing point. θ Phase-shift control is applied to the waveform of the superimposed first and second PWM waves of the fifth switch to obtain the control signal for the fifth switch. Based on the angle by which the secondary current lags behind the primary voltage at its zero-crossing point. θ The waveform of the superimposed first and second PWM waves of the sixth switch is corrected by phase shift control to obtain the control signal of the sixth switch.

2. The control method according to claim 1, characterized in that, When the converter operates in low-voltage mode, a phase-shift control strategy is adopted on the primary side. Specifically, the voltage difference between the sampled output voltage value and the desired voltage is used to adjust the voltage and obtain the phase shift angle. α The control signals for the first and second switching transistors are phase-shifted by an angle of . α Phase shift adjustment is used to obtain control signals for the third and fourth switching transistors.

3. The control method according to claim 2, characterized in that, When the converter operates in high-voltage mode, a phase-shift control strategy is adopted on the primary side. Specifically, the control signals of the first and second switching transistors are adjusted by 180° phase shift to obtain the control signals of the third and fourth switching transistors.

4. The control method according to claim 2, characterized in that, The phase shift angle α With voltage gain G The relationship between them is satisfied The critical conditions for soft switching of the fifth and sixth switching transistors are as follows: Second duty cycle D y2 for The angle at which the secondary current crosses zero lags behind the primary voltage. θ for ,in, Q For quality factor, i sec For the secondary side current, Δ t 1 represents the discharge time of the junction capacitance of the fifth and sixth switching transistors. C eq The equivalent junction capacitance of the switching transistor. V sec This is the secondary side voltage. V O The converter output voltage is ω0, the resonant angular frequency is Δ t 2 represents the time it takes for the secondary current to naturally return to zero after the primary-side switch is switched.

5. The control method according to claim 1, characterized in that, In the first half-bridge inverter unit, the drain of the first switch is connected to the positive terminal of the input DC power supply, the drain of the second switch is connected to the source of the first switch, the source of the second switch is connected to the negative terminal of the input DC power supply, and the drain and source of the second switch constitute the output terminal of the first half-bridge inverter unit. In the second half-bridge inverter unit, the drain of the third switch is connected to the positive terminal of the input DC power supply, the drain of the fourth switch is connected to the source of the third switch, and the source of the fourth switch is connected to the negative terminal of the input DC power supply. The drain and source of the fourth switch constitute the output terminal of the second half-bridge inverter unit.

6. The control method according to claim 5, characterized in that, The first resonant slot includes: a first resonant inductor, a first magnetizing inductor, and a first resonant capacitor. One end of the first resonant inductor is connected to the source of the first switching transistor and the drain of the second switching transistor. The other end of the first resonant inductor is connected to one end of the first magnetizing inductor. The other end of the first magnetizing inductor is connected to the positive terminal of the first resonant capacitor. The negative terminal of the first resonant capacitor is connected to the source of the second switching transistor. The magnetizing inductor of the primary side of the inverting high-frequency transformer serves as the first magnetizing inductor. The second resonant slot includes: a second resonant inductor, a second magnetizing inductor, and a second resonant capacitor. One end of the second resonant inductor is connected to the source of the third switch and the drain of the fourth switch. The other end of the second resonant inductor is connected to one end of the second magnetizing inductor. The other end of the second magnetizing inductor is connected to the positive terminal of the second resonant capacitor. The negative terminal of the second resonant capacitor is connected to the source of the fourth switch. The magnetizing inductor of the primary side of the positive phase high-frequency transformer serves as the second magnetizing inductor.

7. The control method according to claim 6, characterized in that, The first end of the primary winding of the reverse high-frequency transformer is connected to the other end of the first resonant inductor and one end of the first magnetizing inductor, and the second end of the primary winding of the reverse high-frequency transformer is connected to the positive terminal of the first resonant capacitor. The first end of the primary winding of the positive phase high-frequency transformer is connected to the other end of the second resonant inductor and one end of the second magnetizing inductor, and the second end of the primary winding of the positive phase high-frequency transformer is connected to the positive terminal of the second resonant capacitor.

8. The control method according to claim 7, characterized in that, The full-bridge rectifier network includes: a first to a fourth diode, wherein the cathode of the first diode is connected to the cathode of the third diode, the cathode of the second diode is connected to the anode of the first diode to form the midpoint of the first bridge arm of the full-bridge rectifier network, the cathode of the fourth diode is connected to the anode of the third diode to form the midpoint of the second bridge arm of the full-bridge rectifier network, and the anode of the second diode is connected to the anode of the fourth diode; The first end of the secondary winding of the reverse high-frequency transformer is connected as the positive polarity terminal to the midpoint of the first bridge arm of the full-bridge rectifier network. The first end of the secondary winding of the positive high-frequency transformer is connected to the second end of the secondary winding of the reverse high-frequency transformer. The second end of the secondary winding of the positive high-frequency transformer is connected as the negative polarity terminal to the midpoint of the second bridge arm of the full-bridge rectifier network. The first end of the secondary winding of the reverse high-frequency transformer and the second end of the primary winding of the reverse high-frequency transformer are the same-name terminals. The first end of the secondary winding of the positive high-frequency transformer and the first end of the primary winding of the positive high-frequency transformer are the same-name terminals.

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