An integral sliding mode current control method for AC-DC converter for electric vehicles

Abstract

The present invention discloses an integral sliding mode current control method and system for an AC-DC converter for electric vehicles, and relates to the technical field of integral sliding mode current control for AC-DC converters for electric vehicles, including using an improved integral sliding mode control strategy with a nonlinear reaching law to reduce jitter, to track and control the input current. The method of the present invention ensures the accuracy of current control and the robustness of the system by using an integral sliding mode control strategy and nonlinear reaching law regulation, while optimizing the dynamic response of the system and reducing the jitter phenomenon.

Description

Technical Field

[0001] The present invention relates to the technical field of electric vehicles, and in particular to an integral sliding mode current control method and system of an AC-DC converter for an electric vehicle. Background Art

[0002] With the booming development of the electric vehicle industry, on-board chargers, as core components of electric vehicles, are mainly focused on high-frequency, high-efficiency, high-power density and high-performance control technologies. Gallium nitride high electron mobility transistors (GaN High Electron Mobility Transistor, GaN HEMT) have the characteristics of fast switching speed and low loss, which is conducive to improving the switching frequency and overall efficiency of on-board chargers, but it also brings challenges in circuit topology and control technology. Therefore, this paper conducts an in-depth study on the control method of a two-stage bidirectional on-board charger based on GaN HEMT.

[0003] First, the on-board charger must meet the requirements of power quality, that is, it must have functions such as improving power factor and reducing current THD to reduce harmonic pollution to the power grid; secondly, it needs to have a large output voltage range to meet the charging needs of the battery. Since GaN HEMT can effectively suppress reverse recovery, the totem pole bridgeless PFC has been widely used and has become a hot topic in high-frequency PFC topology research. Reference [1] mainly studies the three-way staggered parallel totem pole bridgeless PFC. Since PI control is difficult to track sinusoidal signals, quasi-PR control is used to study G2V, V2G and V2L. The existing technology uses critical conduction mode and adds auxiliary inductance to achieve topological soft switching and reduce conduction loss. The back stage of the on-board charger mainly provides safe and reliable wide-range voltage output. At present, CLLLC resonant converter is the most common. The existing technology studies the synchronous rectification of CLLLC resonant converter to further reduce the conduction loss caused by the body diode.

[0004] In summary, the current research on on-board chargers is mainly focused on improving performance and efficiency, which is reflected in the realization of power factor correction, ripple reduction and harmonic suppression in the front stage, and the realization of efficient isolation and wide range output of DC voltage in the back stage. However, the dynamic performance of PI control is limited. In order to further improve the performance of the whole machine, it is necessary to conduct in-depth research on excellent control strategies. In response to these shortcomings, we have invented an integral sliding mode current control method for AC-DC converters for electric vehicles. Summary of the invention

[0005] The purpose of this section is to summarize some aspects of embodiments of the present invention and briefly introduce some preferred embodiments. Some simplifications or omissions may be made in this section and the specification abstract and the invention title of this application to avoid blurring the purpose of this section, the specification abstract and the invention title, and such simplifications or omissions cannot be used to limit the scope of the present invention.

[0006] In view of the above problems or problems existing in the prior art, the present invention is proposed.

[0007] Therefore, the technical problem solved by the present invention is that the dynamic performance of PI control is limited.

[0008] To solve the above technical problems, the present invention provides the following technical solutions: an integral sliding mode current control method for an AC-DC converter for an electric vehicle, comprising receiving an AC input through a totem pole bridgeless PFC circuit and reducing current ripple through staggered parallel connection;

[0009] Use a software phase-locked loop to obtain the phase information of the grid-side voltage to control the phase of the input current and achieve power factor correction;

[0010] An improved chattering-weakened nonlinear reaching law integral sliding mode control strategy is adopted to track the input current.

[0011] As a preferred solution of the integral sliding mode current control method of the AC-DC converter for electric vehicles of the present invention, the totem pole bridgeless PFC includes improving the totem pole bridgeless PFC.

[0012] The improvements include:

[0013] On the basis of the original single-channel totem pole bridgeless PFC, three parallel high-frequency half-bridges are added. Through phase shifting, the adjacent PWMs are shifted by 90°, and the rise and fall of the inductor current are also shifted by 90°. The four-channel inductor currents are superimposed to reduce the current ripple.

[0014] Through the modal equivalent of the single-channel totem pole bridgeless PFC, the topological mode after interleaving parallel connection is obtained, with the input inductor current i L And the output voltage u o is the state variable. Taking the positive half cycle as an example, the state equation of the inductor charging is obtained as follows:

[0015]

[0016] Among them, the AC input U ac Charge the inductor L to store energy; output filter capacitor C on the output side o For load R L Power supply; set input voltage u in =U ac ;

[0017] The state equation of the inductor current during freewheeling is:

[0018]

[0019] Among them, the AC input U ac The output capacitor and the load are powered together with the inductor L, and the output capacitor is charged; i L is the input inductor current; u o is the output capacitor voltage.

[0020] As a preferred solution of the integral sliding mode current control method of the AC-DC converter for electric vehicles of the present invention, wherein: the reduction of current ripple by staggered parallel connection includes:

[0021] Assume the total current ripple Δi s is the sum of the ripples of the four inductor currents, expressed as:

[0022] Δi s =k(D)Δi L

[0023] Assume that all switches can be switched instantaneously without dead time. Assume that when the carrier is less than the duty cycle, the main switch is turned on to charge the boost inductor. According to the volt-second balance principle, k(D) is expressed as:

[0024]

[0025] Where D is the duty cycle.

[0026] As a preferred solution of the integral sliding mode current control method of the AC-DC converter for electric vehicles of the present invention, wherein: the use of a software phase-locked loop to obtain the phase information of the grid-side voltage to control the phase of the input current and realize power factor correction includes establishing a current loop mathematical model based on a small signal analysis method, and the current loop mathematical model includes:

[0027] The state equation of the totem pole bridgeless PFC in CCM mode is obtained by using the inductor current iL and the output capacitor uc as state variables, which is expressed as:

[0028]

[0029] Among them, u s is the input voltage, d is the main switch duty cycle, R is the output resistance, C is

[0030] Output capacitor value, L is the boost inductor value;

[0031] The interleaved parallel totem pole bridgeless PFC in the positive half cycle of AC is simplified to a Boost converter, where the input current is the inductor current and the output voltage is the capacitor voltage, as shown in the following formula:

[0032]

[0033] Assume the drain-source voltage of the main switch is u T (t), the current on the freewheeling switch is i D (t), the large signal model of the converter within one switching cycle is obtained as follows:

[0034]

[0035] Assume the drain-source voltage u of the main switch tube T (t), the freewheeling secondary switch tube flows through the current i D (t), input current i s (t), output voltage u o The steady-state components of the off duty cycle d′(t) of the main switch are U T ,I D ,I s , U o , D′, and introduce the corresponding small signal perturbations as Adding a small signal perturbation to the above equation yields the following equation:

[0036]

[0037] Since the second-order term of the disturbance can be ignored, it is simplified to:

[0038]

[0039] Get the average switch model in the totem pole bridgeless PFC, and connect the average switch model to the topology to get the required current loop small signal model;

[0040] The Laplace transform of the small signal model is expressed as:

[0041]

[0042] Eliminate the small disturbance of output voltage in the above formula The following formula is obtained:

[0043]

[0044] The transfer function of input current with respect to duty cycle is Is the following formula:

[0045]

[0046] Based on the above formula, obtain the frequency response Bode diagram of the current loop under different duty cycles;

[0047] In practical applications, the output capacitor has a large capacitance, the disturbance can be ignored and is considered to be zero. Duty cycle disturbance Equal, further simplify the above formula and express it as:

[0048]

[0049] Through the small signal analysis method, the mathematical model of the voltage loop is obtained:

[0050]

[0051] The entire system is controlled by a negative feedback double closed-loop control structure. The outer loop of the double closed-loop is a voltage loop, which is used to control the output voltage of the PFC to achieve output voltage regulation; the inner loop of the double closed-loop is a current loop, which is used to control the input current and control the input current to track the phase of the input voltage to achieve power factor correction.

[0052] As a preferred solution of the integral sliding mode current control method of the AC-DC converter for electric vehicles of the present invention, wherein: the improved vibration weakening nonlinear reaching law integral sliding mode control strategy is adopted to track and control the input current, including:

[0053] Proportional terms and integral terms of output voltage and input current are introduced into the control system to form an integral sliding surface;

[0054] According to the sliding surface, the corresponding reaching law is designed to make the system state move toward the sliding surface and finally converge.

[0055] As a preferred solution of the integral sliding mode current control method of the AC-DC converter for electric vehicles of the present invention, wherein: the proportional term and the integral term of the output voltage and the input current are introduced into the control system to form an integral sliding mode surface, including:

[0056] The main object of the integral sliding mode control is the input current is. In order to improve the anti-interference performance of the output voltage, the output voltage U is introduced into the design of the sliding mode surface. o , the sliding surface S is composed of the proportional term and integral term of the current and voltage errors, as shown in the following formula:

[0057]

[0058] Among them, α1, α2, α3, and α4 are the parameters of the four components of the sliding surface. and are the given values ​​of output voltage and input current respectively.

[0059] As a preferred solution of the integral sliding mode current control method of the AC-DC converter for electric vehicles of the present invention, wherein: the design of the corresponding convergence law according to the sliding mode surface so that the system state moves toward the sliding mode surface and finally converges includes:

[0060] The improved reaching law is shown as follows:

[0061]

[0062] The stability is proved by Lyapunov function, which is expressed as:

[0063]

[0064] After derivation, we have:

[0065]

[0066] Since a>0, the stability condition is satisfied:

[0067]

[0068] According to the sliding surface formula, the derivative can be obtained:

[0069]

[0070] Among them, the given current of the current tracking controller is given by the voltage loop. Since the voltage loop adopts PI control, it can be obtained:

[0071]

[0072] Among them, K i is the voltage loop integral parameter;

[0073] Based on the topological mathematical model, the simultaneous equation can be obtained:

[0074]

[0075] Combining the reaching law and the above equation, we can get the duty cycle as:

[0076]

[0077] Beneficial effects of the present invention: The present invention significantly improves the efficiency of the AC-DC converter, reduces the loss of the system, and enhances the power factor correction capability by adopting GaN HEMT devices and four-way staggered parallel totem pole bridgeless PFC circuits. By improving the input current waveform, the harmonic pollution to the power grid is reduced and the power quality is improved. In addition, the present invention uses an integral sliding mode control strategy and nonlinear convergence law regulation to ensure the accuracy of current control and the robustness of the system, while optimizing the dynamic response of the system and reducing the jitter phenomenon. The PI control strategy based on duty cycle compensation further suppresses the zero-crossing current spike to ensure the smooth operation of the system. Through reasonable parameter design and inductor selection, the present invention has high design flexibility and adaptability, and can meet diverse application requirements. BRIEF DESCRIPTION OF THE DRAWINGS

[0078] In order to more clearly illustrate the technical solutions of the embodiments of the present invention, the following briefly introduces the drawings required for describing the embodiments. Obviously, the drawings described below are only some embodiments of the present invention. For ordinary technicians in this field, other drawings can be obtained based on these drawings without creative labor. Among them:

[0079] Figure 1 Schematic diagram of single-channel totem pole bridgeless PFC.

[0080] Figure 2 Schematic diagram of four-way staggered parallel totem pole bridgeless PFC.

[0081] Figure 3 Schematic diagram of the working mode of a single-channel totem pole bridgeless PFC.

[0082] Figure 4 The carrier diagram of the four-way staggered parallel totem pole bridgeless PFC.

[0083] Figure 5 This is a schematic diagram of the current loop small signal model.

[0084] Figure 6 The Bode diagram is the frequency response of input current with respect to duty cycle under different duty cycles.

[0085] Figure 7 To simplify the Bode plot of the frequency response of the input current with respect to the duty cycle.

[0086] Figure 8 The figure is the control block diagram of four-way staggered parallel totem pole bridgeless PFC.

[0087] Fig. 9 It is the zero-crossing current spike of the totem pole bridgeless PFC.

[0088] Fig.10This is the ideal high-frequency lower tube duty cycle waveform of the totem pole bridgeless PFC.

[0089] Fig.11 To improve the reaching law function image.

[0090] Fig.12 This is the structural block diagram of the current tracking controller based on sliding mode control.

[0091] Fig.13 This is a comparison chart of power factor correction results.

[0092] Fig.14 Comparison of input current FFT of integral sliding mode control with different reaching laws.

[0093] Fig.15 The figure is a comparison chart of the output voltage anti-disturbance performance of integral sliding mode control with different reaching laws.

[0094] Fig.16 This is a comparison chart of the four-way single inductor current and the total input current.

[0095] Fig.17 Comparison chart of experimental waveforms of different current control strategies.

[0096] Fig.18 This is the key waveform diagram of half-load state.

[0097] Fig.19 This is the waveform diagram of the four-way staggered parallel experiment.

[0098] Fig. 20 It is the key waveform diagram of rated working condition. DETAILED DESCRIPTION

[0099] In order to make the above-mentioned objects, features and advantages of the present invention more obvious and easy to understand, the specific implementation methods of the present invention are described in detail below in conjunction with the accompanying drawings.

[0100] In the following description, many specific details are set forth to facilitate a full understanding of the present invention, but the present invention may also be implemented in other ways different from those described herein, and those skilled in the art may make similar generalizations without violating the connotation of the present invention. Therefore, the present invention is not limited to the specific embodiments disclosed below.

[0101] Secondly, the term "one embodiment" or "embodiment" as used herein refers to a specific feature, structure, or characteristic that may be included in at least one implementation of the present invention. The term "in one embodiment" that appears in different places in this specification does not necessarily refer to the same embodiment, nor does it refer to a separate or selective embodiment that is mutually exclusive with other embodiments.

[0102] Example 1

[0103] Reference Figures 1 to 12 , as one embodiment of the present invention, provides an integral sliding mode current control method for an AC-DC converter for an electric vehicle, comprising:

[0104] S1: Receives AC input through a totem pole bridgeless PFC circuit and reduces current ripple by interleaving in parallel.

[0105] It should be noted that the single-channel totem pole bridgeless PFC structure is as follows Figure 1 Totem pole bridgeless PFC mainly solves the following two problems: ① Boost PFC has many switching devices, large switching losses, and large equipment size; ② Basic bridgeless PFC has high-frequency switching tubes placed in the lower bridge arm, which will cause serious common-mode interference and make filter design difficult.

[0106] from Figure 1 It can be seen that the totem pole bridgeless PFC is mainly composed of two half-bridges, one is a high-frequency half-bridge, including two high-frequency tubes S1 and S2, and the midpoint of its bridge arm is connected to the power transmission inductor L. The bridge arm charges and discharges the inductor through high-frequency switching to achieve the Boost function; by controlling the phase of the inductor current, the power factor correction is completed. The other half-bridge is the power frequency bridge arm, including power frequency tubes S3 and S4. In order to reduce the switching loss caused by the passive diode, the diode is replaced with SiMOSFET. In the positive half cycle of AC, the lower tube S4 is turned on; conversely, in the negative half cycle of AC, the upper tube S3 is turned on, so the main function of this bridge arm is rectification.

[0107] In order to reduce the ripple of the total input current and improve the power of the whole machine, this paper adopts a four-way staggered parallel totem pole bridgeless PFC. The topology is as follows Figure 2 As shown in the figure, Uac is the grid voltage; is is the input current; iL1~iL4 are the four-way inductor currents; Uo is the output capacitor voltage; L1~L4 are the four-way inductors; S1~S8 are the high-frequency bridge arm upper and lower tubes; S9 and S10 are the power frequency bridge arm upper and lower tubes; Co is the output capacitor; RL is the load.

[0108] Furthermore, the totem pole bridgeless PFC includes improving the totem pole bridgeless PFC.

[0109] The improvements include:

[0110] On the basis of the original single-channel totem pole bridgeless PFC, three parallel high-frequency half-bridges are added. Through phase shifting, the adjacent PWMs are shifted by 90°, and the rise and fall of the inductor current are also shifted by 90°. The four inductor currents are superimposed to reduce the current ripple.

[0111] It should be noted that the modes of the totem pole bridgeless PFC are mainly distinguished according to the switch states of the two sets of half-bridges. Although the interleaved parallel technology increases the number of high-frequency bridge arms, its essence is a combination of multiple single-channel topologies. Therefore, the modes of the single-channel totem pole bridgeless PFC can be used to equal the topological modes after interleaved parallel connection. According to the modal analysis, this section will use the input inductor current i L And the output voltage u o is the state variable, and the state equation of the converter in different working modes is listed.

[0112] The working modes of the totem pole bridgeless PFC can be mainly divided into the following four types:

[0113] Working mode 1: Figure 3 As shown in (a), in working mode 1, in the positive half cycle of AC, the high-frequency upper tube S1 is turned off, and the high-frequency lower tube S2 is turned on; the power frequency upper tube S3 is turned off, and the power frequency lower tube S4 is turned on. The AC input Uac charges the inductor L for energy storage; the output filter capacitor Co on the output side supplies power to the load RL. Assuming the input voltage uin = Uac, the state equation of formula (1.1) can be obtained in this working mode.

[0114] Through the modal equivalent of the single-channel totem pole bridgeless PFC, the topological mode after interleaving parallel connection is obtained, with the input inductor current i L And the output voltage u o is the state variable. Taking the positive half cycle as an example, the state equation of the inductor charging is obtained as follows:

[0115]

[0116] Among them, the AC input U ac Charge the inductor L to store energy; output filter capacitor C on the output side o For load R L Power supply; set input voltage u in =U ac ;

[0117] Working mode 2: Figure 3 As shown in (b), working mode 2 is still in the positive half cycle of AC, the high-frequency upper tube S1 is turned on, and the high-frequency lower tube S2 is turned off; the power frequency upper tube S3 is turned off, and the power frequency lower tube S4 is turned on. The AC input Uac and the inductor L simultaneously supply power to the output side capacitor and the load, and the output side capacitor is charged. In working mode 2, the input inductor current iL and the output capacitor voltage uo are shown in formula (1.2).

[0118] The state equation of the inductor current during freewheeling is:

[0119]

[0120] Among them, the AC input U acThe output capacitor and the load are powered together with the inductor L, and the output capacitor is charged; i L is the input inductor current; u o is the output capacitor voltage.

[0121] Working mode 3: Figure 3 As shown in (c), the working mode 3 topology enters the negative half-cycle of AC, so it is necessary to swap the power frequency tubes, the power frequency upper tube S3 is turned on, and the power frequency lower tube S4 is turned off; the high frequency upper tube S1 of the high frequency bridge arm is turned on as the main switch tube, and the high frequency lower tube S2 is turned off as the freewheeling tube; the AC input Uac charges the inductor L to store energy, and the inductor current rises; the output side load is completely powered by the output capacitor. In working mode 3, the converter as a whole is similar to working mode 1, except that the AC input is in the negative half-cycle at this time, and rectification needs to be achieved by adjusting the power frequency bridge arm, so the state equation of the working mode is consistent with equation (1.1).

[0122] Working mode 4: Figure 3 As shown in (d), working mode 4 is in the negative half cycle of AC, the power frequency upper tube S3 is turned on, and the power frequency lower tube S4 is turned off; the high frequency upper tube S1 of the high frequency bridge arm is turned off, and the high frequency lower tube S2 is turned on as a freewheeling tube; the AC input Uac and the inductor L function to the output side, and the inductor current decreases; the output side capacitor is charged by the input voltage and the inductor. Since the inductor is charged in working mode 3, this stage belongs to the boost stage with working mode 2. The state equation under this working mode is consistent with equation (1.2).

[0123] Furthermore, this paper adopts the total current ripple of four-way staggered parallel connection. In order to facilitate the analysis of the principle and effect of staggered parallel connection, a sawtooth carrier is used for analysis, such as Figure 4 shown.

[0124] Depend on Figure 4 It can be seen that the four-way inductor is staggered in parallel, and the PWM modulation carrier is delayed by 90° in sequence. In one switching cycle, the same carrier is divided into four equivalent intervals. By analyzing the current fluctuation in one interval, the current fluctuation of the entire switching cycle can be obtained; the four-way staggered parallel connection increases the total current ripple by four times while ensuring that the branch switching cycle remains unchanged. Therefore, compared with a single-way inductor, the input current frequency is also increased by four times, which is more convenient for filter design. Figure 3 From the analysis, we can know that there are two working modes of the Boost converter, one is inductor charging and the other is inductor freewheeling. Since the totem pole bridgeless PFC is AC input, it needs to be rectified separately, so 4 working modes are generated. Using four-way staggered parallel technology, the working mode of the topology is actually a combination of four single-way totem pole bridgeless PFC working modes. Therefore, there are 16 working modes in half a cycle of the AC voltage. Through the power frequency bridge arm rectification, there are 32 working modes in one power frequency, which will not be repeated.

[0125] Furthermore, the reducing current ripple by interleaving in parallel includes,

[0126] Assume the total current ripple Δi s is the sum of the ripples of the four inductor currents, expressed as:

[0127] Δi s =k(D)Δi L (1.3)

[0128] Assuming that all switches can be switched instantaneously without dead time, when the carrier is less than the duty cycle, the main switch is turned on to charge the boost inductor. According to the volt-second balance principle, Figure 4 It is not difficult to deduce that when the duty cycle D is (0,0.25), there must be one path in the inductor charging state, and the rest are in the freewheeling state, that is, only one path is in mode 1, and the other three paths are in mode 2; when the duty cycle D is (0.25,0.5), two paths will be in mode 1, and the other two paths will always be in mode 2; when the duty cycle D is (0.5,0.75), three paths will be in mode 1, and the remaining path will always be in mode 2; when the duty cycle D is (0.75,1), three paths will remain in mode 1, and the other path will enter mode 2. It is not difficult to solve k(D) as shown in the following formula (1.4):

[0129]

[0130] It should be noted that, from formula (2.4), k(D) is always less than 1. It can be seen that interleaving and paralleling can effectively reduce current ripple and improve power level. However, with each additional interleaving and paralleling, the circuit structure will become more complicated, and the number of devices required will also increase, which will increase the cost. Therefore, the number of interleaving and paralleling should not be too large.

[0131] S2: Use a software phase-locked loop to obtain the phase information of the grid-side voltage to control the phase of the input current and achieve power factor correction.

[0132] Furthermore, the use of a software phase-locked loop to obtain the phase information of the grid-side voltage to control the phase of the input current and achieve power factor correction includes establishing a current loop mathematical model based on a small signal analysis method, and the current loop mathematical model includes:

[0133] The staggered parallel totem pole bridgeless PFC can be simplified to a single-channel totem pole bridgeless PFC for analysis, and the totem pole bridgeless PFC can be simplified to a Boost converter for analysis. Therefore, equations (1.1) and (1.2) can be combined. To facilitate the distinction between the input current and the single-channel inductor current, the output voltage and the output capacitor voltage, the inductor current iL and the output capacitor uc are selected as state variables to obtain the state equation (1.5) of the totem pole bridgeless PFC in CCM mode, which is expressed as:

[0134]

[0135] Among them, u s is the input voltage, d is the main switch duty cycle, R is the output resistor value, C is the output capacitor value, and L is the boost inductor value;

[0136] Formula (1.5) has simplified the interleaved parallel totem pole bridgeless PFC in the positive half cycle of AC into a Boost converter. The mathematical model of the negative half cycle of AC is similar and will not be repeated. Therefore, the input current is the inductor current, and the output voltage is the capacitor voltage as shown in Formula (1.6), which is expressed as:

[0137]

[0138] Assume the drain-source voltage of the main switch is u T (t), the current on the freewheeling switch is i D (t), the large signal model of the converter within one switching cycle is obtained as follows:

[0139]

[0140] Since the switching frequency of the high-frequency bridge arm of the totem pole bridgeless PFC is much higher than the grid power frequency, and the output capacitor is generally large, the changes in the output voltage, input voltage and duty cycle are relatively slow. It can be considered that the totem pole bridgeless PFC operates at a quasi-static operating point, which is conducive to the use of small signal analysis to establish the current loop model.

[0141] It can be assumed that the drain-source voltage u of the main switch tube T (t), the freewheeling secondary switch tube flows through the current i D (t), input current i s (t), output voltage u o The steady-state components of the off duty cycle d′(t) of the main switch are U T ,I D ,I s , U o , D′, and introduce the corresponding small signal perturbations as Adding a small signal perturbation to the above equation yields the following equation:

[0142]

[0143] Since the second-order term of the disturbance is very small, it can be ignored, and equation (1.8) can be simplified to equation (1.9):

[0144]

[0145] The average switch model in the totem pole bridgeless PFC can be obtained from equation (1.8). By connecting the average switch model into the topology, the required current loop small signal model can be obtained as follows: Figure 5 shown.

[0146] The Laplace transform of the small signal model is expressed as:

[0147]

[0148] Eliminate the small output voltage disturbance in equation (1.10) The following formula is obtained:

[0149]

[0150] The transfer function of input current with respect to duty cycle is Is the following formula:

[0151]

[0152] Based on formula (1.12), the frequency response Bode diagram of the current loop under different duty cycles is obtained; Figure 6 It can be seen that different duty cycles mainly affect the low-frequency part of the frequency response, while the high-frequency part remains basically unchanged.

[0153] Since the switching frequency of the high-frequency bridge arm is generally high, the bandwidth of the current loop is generally about 1 / 10 of the switching frequency, which is much higher than the resonant frequency. Therefore, the duty cycle has little effect on the low-frequency part. In addition, in practical applications, the capacitance of the output capacitor is generally large, so the disturbance is very small and can be ignored as zero. At the same time, it is assumed that the disturbance of the off duty cycle is equal to the disturbance of the on duty cycle, and equation (1.12) can be further simplified to equation (1.13). The simplified frequency response of the input current to the duty cycle is as follows: Figure 7 As shown:

[0154]

[0155] Through the small signal analysis method, the mathematical model of the voltage loop is obtained:

[0156]

[0157] Based on the above analysis, this paper controls the entire system through a negative feedback double closed-loop control structure. The outer loop of the double closed-loop is a voltage loop, which is used to control the output voltage of the PFC to achieve output voltage regulation; the inner loop of the double closed-loop is a current loop, which is used to control the input current and control the input current to track the phase of the input voltage to achieve power factor correction.

[0158] Furthermore, in the dual closed-loop control, the voltage outer loop realizes that the output voltage follows the reference given value. The software phase lock loop (SPLL) is used to obtain the phase information of the grid-side voltage, and the phase information is multiplied by the output of the voltage outer loop to obtain a given current value, which is used as the current loop input to complete the control of the entire topology. The overall block diagram of the control system is shown in the figure. Figure 8 shown.

[0159] It should be noted that since this article adopts multi-channel staggered parallel connection, if a single inductor current is still controlled, the chip resources are insufficient. Therefore, the current loop of this article samples and controls the total input current to achieve control of the total current. Compared with controlling a single inductor current, this method greatly saves control chip resources.

[0160] S3: An improved chattering-weakened nonlinear reaching law integral sliding mode control strategy is used to track the input current.

[0161] From the above modal analysis, it can be seen that the function of the high-frequency switch tube of the totem pole bridgeless PFC depends on the positive and negative cycles of the AC voltage. The function switching occurs at the zero crossing point of the AC voltage, so it will produce the following Fig. 9 The zero-crossing current spike problem not only increases the loss of the whole machine during operation and reduces efficiency, but also, as the bus voltage increases, excessive current spikes will also cause overcurrent faults in the whole machine. Therefore, soft zero-crossing measures must be taken to achieve current spike suppression.

[0162] Assume that the zero-crossing process from the negative half cycle to the positive half cycle is positive zero-crossing, and the zero-crossing process from the positive half cycle to the negative half cycle is negative zero-crossing. Since the generation mechanism of the zero-crossing current spike is the same, only the positive zero-crossing is analyzed below.

[0163] like Fig.10 The figure shows the ideal duty cycle waveform of the high-frequency tube. It can be seen that during the switching of the positive and negative half-cycles of the AC, due to the switching of the main and auxiliary switch tube functions, the ideal high-frequency tube duty cycle needs to complete the jump between 0% and 100% to achieve inductor charging; however, there is a delay in digital control itself, and the duty cycle under PI control cannot achieve a sudden change due to the existence of the integral term. This causes the voltage on the DC side of the totem pole PFC to flow back to the AC side when it crosses zero positively, resulting in a negative current spike. Fig.10It can be seen that when the negative half-cycle enters the positive zero crossing, the duty cycle of the high-frequency lower tube is relatively small, because at this time the high-frequency lower tube is used as a freewheeling tube; after entering the positive half-cycle, the high-frequency lower tube needs to be switched to the main switch tube and turned on with a large duty cycle to charge the inductor as much as possible, but because the integral value of PI needs to change gradually, the high-frequency lower tube cannot be turned on with a large duty cycle immediately. At this time, the grid voltage is very small, and the opening of the complementary high-frequency upper tube will directly cause the bus voltage to flow back to the input side, forming a negative current spike.

[0164] Based on the above analysis of the current spike generation mechanism of the totem pole bridgeless PFC at the zero-crossing point, a corresponding suppression strategy will be adopted to solve the problem.

[0165] Taking the positive zero crossing as an example, since the power frequency tube does not have high requirements for the switching frequency, traditional Si MOSFET or IGBT is often used. In order to prevent the bridge arm from being directly connected and reverse recovery, a larger dead zone is usually required between the power frequency bridge arms, during which all switching tubes need to be blocked.

[0166] As mentioned above, one of the reasons for the positive zero-crossing negative current spike is that the integral value of the PI control cannot be changed suddenly, resulting in the duty cycle of the high-frequency lower tube changing during the power frequency bridge arm. After all the switch tubes are unblocked and enter the positive half-cycle, it needs to slowly change from 0% to 100%, which causes the complementary high-frequency upper tube to have a large duty cycle. Therefore, when officially entering the positive half-cycle, this paper processes the phase information obtained by the phase-locked loop, clears the integral value of PI, and uses the ideal open-loop duty cycle to compensate for the integral value of PI and limit the conduction of the complementary tube, helping the PI control to quickly enter the positive half-cycle control, thereby realizing the duty cycle jump on the high-frequency bridge arm and suppressing the generation of current spikes.

[0167] Furthermore, the reason why the totem pole bridgeless PFC has a zero-point current spike during control is the switching of the high-frequency bridge arm switch tube function, and the PI control itself does not require an accurate mathematical model for control, and the control itself has a lag; in addition, the current loop needs to complete the tracking of the phase, so the current loop input is a sine signal, and the PI control has a weak ability to track this signal without static errors. Therefore, if there is an accurate mathematical model for control, without the need for compensation through negative feedback like PI control, it can not only improve the robustness during state switching, but also improve the overall dynamic performance of the system, improve the current tracking effect, and thus improve the quality of the current waveform.

[0168] Compared with PI control, sliding mode control is based on the topological mathematical model, and uses the sliding surface and reaching law as methods to make the points in the state space regress and stabilize. Its duty cycle output is obtained based on the mathematical model, sliding surface and reaching law, so the robustness and rapidity are better than PI control. In summary, this paper uses a current controller based on integral sliding mode to control the input AC current. By introducing the integral component, the integral sliding mode can effectively suppress the chattering problem of the traditional sliding mode and improve the dynamic performance of the system.

[0169] Furthermore, the improved chattering weakening nonlinear reaching law integral sliding mode control strategy is adopted to track and control the input current, including:

[0170] Proportional terms and integral terms of output voltage and input current are introduced into the control system to form an integral sliding surface;

[0171] According to the sliding surface, the corresponding reaching law is designed to make the system state move toward the sliding surface and finally converge.

[0172] It should be noted that the introduction of proportional terms and integral terms of output voltage and input current into the control system to form an integral sliding surface includes:

[0173] The main object of the integral sliding mode control is the input current is. In order to improve the anti-interference performance of the output voltage, the output voltage U is introduced into the design of the sliding mode surface. o , the sliding surface S is composed of the proportional term and integral term of the current and voltage errors, as shown in the following formula:

[0174]

[0175] Among them, α1, α2, α3, and α4 are the parameters of the four components of the sliding surface. and are the given values ​​of output voltage and input current respectively.

[0176] Furthermore, the approach law of the sliding surface design to make the system state move toward the sliding surface and finally converge includes:

[0177] The improved reaching law is shown as follows:

[0178]

[0179] It should be noted that the power approaching law can increase the approaching speed and weaken the chattering; however, the parameters restrict each other in the control effect: when far away from the sliding surface, a larger Increase the approach speed to return to the sliding surface as soon as possible; when the state is near the sliding surface, a smaller This reduces vibration and improves stability.

[0180] The function graph is as follows Fig.11 shown.

[0181] The stability is proved by Lyapunov function, which is expressed as:

[0182]

[0183] After derivation, we have:

[0184]

[0185] Since a>0, the stability condition is satisfied:

[0186]

[0187] According to the sliding surface formula (2.1), taking its derivative we can get:

[0188]

[0189] Among them, the given current of the current tracking controller is given by the voltage loop. Since the voltage loop adopts PI control, it can be obtained:

[0190]

[0191] Among them, K i is the voltage loop integral parameter;

[0192] Based on the topological mathematical model, the simultaneous equations (1.5), (2.6) and (2.7) yield:

[0193]

[0194] Combining the reaching law and the above equation, we can get the duty cycle as:

[0195]

[0196] It should be noted that the control block diagram of ISMC is as follows Fig.12 Compared with PI control, which realizes negative feedback compensation based on error, ISMC uses a certain convergence law to force the system state to return to stability and suppress errors based on the mathematical model of the topology, so the robustness and rapidity are improved.

[0197] In summary, the present invention significantly improves the efficiency of the AC-DC converter, reduces the loss of the system, and enhances the power factor correction capability by adopting GaN HEMT devices and four-way staggered parallel totem pole bridgeless PFC circuits. By improving the input current waveform, the harmonic pollution to the power grid is reduced and the power quality is improved. In addition, the present invention uses an integral sliding mode control strategy and nonlinear convergence law regulation to ensure the accuracy of current control and the robustness of the system, while optimizing the dynamic response of the system and reducing the jitter phenomenon. The PI control strategy based on duty cycle compensation further suppresses the zero-crossing current spike and ensures the smooth operation of the system. Through reasonable parameter design and inductor selection, the present invention has high design flexibility and adaptability, and can meet diverse application requirements.

[0198] Example 2

[0199] Reference Figures 13 to 20 , which is an embodiment of the present invention, provides an integral sliding mode current control method for an AC-DC converter for an electric vehicle. In order to verify the beneficial effects of the present invention, scientific demonstration is carried out through calculation and simulation experiments.

[0200] Simulation and Experiment of Four-way Interleaved Totem Pole Bridgeless PFC

[0201] Simulation Analysis

[0202] like Fig.13 As shown in the figure, under rated full load conditions, the power factor correction results under PI control are compared with the power factor correction under integral sliding mode control. Fig.13 (a) is the power factor correction result under PI control, the power factor is 0.991, and the THD is about 4.72%; Fig.13 (b) is the power factor correction result under integral sliding mode control. The power factor is further improved to 0.998 and the THD is reduced to 3.36%.

[0203] like Fig.14 The FFT comparison of the input current of the integral sliding mode control with different reaching laws. Figure (a) is the integral sliding mode control with the general constant speed reaching law, and the THD is 3.36%; Figure (b) is the integral sliding mode control with the improved vibration weakening nonlinear reaching law, and the THD is reduced to 2.4%, which can be seen that the harmonic pollution is effectively suppressed.

[0204] Further, FFT comparison.

[0205] like Fig.15The output voltage anti-disturbance performance comparison of integral sliding mode control with different reaching laws. Figure (a) shows the integral sliding mode control with a general constant speed reaching law. After the load disturbance is applied, the maximum output voltage reaches 434V and it takes 0.18s to recover. Figure (b) shows the integral sliding mode control with an improved jitter-weakened nonlinear reaching law. After the same disturbance is applied, the maximum voltage is reduced to 421V and the time to recover is also reduced to 0.15s, which effectively improves the rapidity of the system.

[0206] like Fig.16 is the comparison waveform of the four-way inductor current and the total input current. Fig.16 It can be seen that the four single-inductor currents differ by 90° in phase, and the overall waveform is consistent with the total current. The ripple of the single-inductor current peak is about 4A, and the total current ripple is greatly reduced to about 500mA at the peak, which is reduced by about 87.5%. Therefore, it can be seen that after multi-channel staggered parallel connection, the current ripple has been effectively reduced and the power level has been significantly improved.

[0207] Experimental verification

[0208] like Fig.17 As shown in Figure 1, the experimental waveform comparison of the integral sliding mode current control strategy based on the improved nonlinear reaching law for jitter reduction is used. Fig.17 It can be seen that the current waveform based on PI control has obvious current distortion and poor sinusoidality due to its poor speed. After using the integral sliding mode current control strategy based on the improved jitter-weakened nonlinear reaching law, the current distortion is significantly reduced, proving the effectiveness of the strategy.

[0209] At a switching frequency of 200kHz, half-load tests with inductor currents of 3A and 4A were completed. The experimental results are shown in the figure below: Fig.18 As shown. Fig.18 It can be seen that near half load, the power factor has been greatly improved compared with light load. The power factors are 0.992 and 0.996 when the inductor current is 3A and 4A respectively. It is generally hoped that the power factor of PFC under half load needs to be greater than 0.96. Therefore, it can be seen from the half-load experiment that the four-way staggered parallel totem pole bridgeless PFC designed in this paper meets the requirements.

[0210] Due to the limited output capacity of the experimental equipment, the four-way staggered parallel totem pole bridgeless PFC can only complete the half-load experiment. The experimental results are as follows Fig.19 As shown in Figure 1, channels 2-4 are the single inductor current, input voltage, and input current, respectively. After interleaving in parallel, the input current ripple is reduced by nearly 75%.

[0211] At 200kHz switching frequency, the test results of rated load condition are as follows: Fig. 20As shown, Figure (a) is the waveform on the AC side, Channel 2 is the single inductor current waveform, Channel 3 is the input AC voltage waveform, and Figure (b) is the DC side output DC voltage waveform. Fig. 20 It can be seen that under rated operating conditions, the power factor is further improved to 0.9968, ensuring better power quality on the AC side; the output voltage double frequency ripple is about 5% of the output voltage, which meets the requirements.

[0212] It should be understood that the various parts of the present invention can be implemented by hardware, software, firmware or a combination thereof. In the above-mentioned embodiments, multiple steps or methods can be implemented by software or firmware stored in a memory and executed by a suitable instruction execution system. For example, if implemented by hardware, as in another embodiment, it can be implemented by any one of the following technologies known in the art or their combination: a discrete logic circuit having a logic gate circuit for implementing a logic function for a data signal, a dedicated integrated circuit having a suitable combination of logic gate circuits, a programmable gate array (PGA), a field programmable gate array (FPGA), etc. It should be noted that the above embodiments are only used to illustrate the technical solution of the present invention and are not limited. Although the present invention is described in detail with reference to the preferred embodiments, it should be understood by those skilled in the art that the technical solution of the present invention can be modified or replaced by equivalents without departing from the spirit and scope of the technical solution of the present invention, which should be included in the scope of the claims of the present invention.

[0213] It should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention rather than to limit it. Although the present invention has been described in detail with reference to the preferred embodiments, those skilled in the art should understand that the technical solutions of the present invention may be modified or replaced by equivalents without departing from the spirit and scope of the technical solutions of the present invention, which should all be included in the scope of the claims of the present invention.

Claims

1. An integral sliding mode current control method for an AC-DC converter for an electric vehicle, characterized in that: include, The totem pole bridgeless PFC circuit receives AC input and reduces current ripple by interleaving in parallel; Use a software phase-locked loop to obtain the phase information of the grid-side voltage to control the phase of the input current and achieve power factor correction; The chattering-weakened nonlinear reaching law integral sliding mode control strategy is adopted to track the input current; The chattering weakening nonlinear reaching law integral sliding mode control strategy adopted to track the input current includes: Proportional terms and integral terms of output voltage and input current are introduced into the control system to form an integral sliding surface; Design the corresponding convergence law according to the sliding surface, so that the system state moves toward the sliding surface and finally converges; The method of introducing proportional terms and integral terms of output voltage and input current into the control system to form an integral sliding surface includes: The main object of the integral sliding mode control is the input current is. In order to improve the anti-interference performance of the output voltage, the output voltage U is introduced into the design of the sliding mode surface. o , the sliding surface S is composed of the proportional term and integral term of the current and voltage errors, as shown in the following formula: Among them, α1, α2, α3, and α4 are the parameters of the four components of the sliding surface. and are the given values ​​of output voltage and input current respectively; The design of the corresponding convergence law according to the sliding surface so that the system state moves toward the sliding surface and finally converges includes: The reaching law adopted is as follows: The stability is proved by Lyapunov function, which is expressed as: After derivation, we have: Since a>0, the stability condition is satisfied: According to the sliding surface formula, the derivative can be obtained: Among them, the given current of the current tracking controller is given by the voltage loop. Since the voltage loop adopts PI control, it can be obtained: Among them, K i is the voltage loop integral parameter; Based on the topological mathematical model, the simultaneous equation can be obtained: Combining the reaching law and the above equation, we can get the duty cycle as: 。 2. The integral sliding mode current control method of the AC-DC converter for electric vehicles according to claim 1, characterized in that: The totem pole bridgeless PFC includes adding three parallel high-frequency half-bridges on the basis of a single-channel totem pole bridgeless PFC, and making adjacent PWMs 90° apart from each other through phase shifting, and the rise and fall of the inductor current also 90° apart from each other, and the four-channel inductor current is superimposed to reduce the current ripple; Through the modal equivalent of the single-channel totem pole bridgeless PFC, the topological mode after interleaving parallel connection is obtained, with the input inductor current i L And the output voltage u o is the state variable. Taking the positive half cycle as an example, the state equation of the inductor charging is obtained as follows: Among them, the AC input U ac Charge the inductor L to store energy; output filter capacitor C on the output side o For load R L Power supply; set input voltage u in =U ac ; The state equation of the inductor current during freewheeling is: Among them, the AC input U ac The output capacitor and the load are powered together with the inductor L, and the output capacitor is charged; i L is the input inductor current; u o is the output capacitor voltage.

3. The integral sliding mode current control method of the AC-DC converter for electric vehicles as claimed in claim 2, characterized in that: The current ripple reduction by interleaving in parallel includes, Assume the total current ripple Δi s is the sum of the ripples of the four inductor currents, expressed as: Δi s =k(D)Δi L Assume that all switches can be switched instantaneously without dead time. Assume that when the carrier is less than the duty cycle, the main switch is turned on to charge the boost inductor. According to the volt-second balance principle, k(D) is expressed as: Where D is the duty cycle.

4. The integral sliding mode current control method of the AC-DC converter for electric vehicles as claimed in claim 3, characterized in that: The use of a software phase-locked loop to obtain the phase information of the grid-side voltage to control the phase of the input current and achieve power factor correction includes establishing a current loop mathematical model based on a small signal analysis method, wherein the current loop mathematical model includes: The state equation of the totem pole bridgeless PFC in CCM mode is obtained by using the inductor current iL and the output capacitor uc as state variables, which is expressed as: Among them, u s is the input voltage, d is the main switch duty cycle, R is the output resistor value, C is the output capacitor value, and L is the boost inductor value; The interleaved parallel totem pole bridgeless PFC in the positive half cycle of AC is simplified to a Boost converter, where the input current is the inductor current and the output voltage is the capacitor voltage, as shown in the following formula: Assume the drain-source voltage of the main switch is u T (t), the current on the freewheeling switch is i D (t), the large signal model of the converter within one switching cycle is obtained as follows: Assume the drain-source voltage u of the main switch tube T (t), the freewheeling secondary switch tube flows through the current i D (t), input current i s (t), output voltage u o The steady-state components of the off duty cycle d'(t) of the main switch are U T ,I D ,I s , U o , D', and introduce the corresponding small signal perturbations as Adding a small signal perturbation to the above equation yields the following equation: Since the second-order term of the disturbance can be ignored, it is simplified to: Get the average switch model in the totem pole bridgeless PFC, and connect the average switch model to the topology to get the required current loop small signal model; The Laplace transform of the small signal model is expressed as: Eliminate the small disturbance of output voltage in the above formula The following formula is obtained: The transfer function of input current with respect to duty cycle is Is the following formula: Based on the above formula, obtain the frequency response Bode diagram of the current loop under different duty cycles; In practical applications, the output capacitor has a large capacitance, the disturbance can be ignored and is considered to be zero. Duty cycle disturbance Equal, further simplify the above formula and express it as: Through the small signal analysis method, the mathematical model of the voltage loop is obtained: The entire system is controlled by a negative feedback double closed-loop control structure. The outer loop of the double closed-loop is a voltage loop, which is used to control the output voltage of the PFC to achieve output voltage regulation; the inner loop of the double closed-loop is a current loop, which is used to control the input current and control the input current to track the phase of the input voltage to achieve power factor correction.

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