Frequency-doubling soft-switching control method and controller for six-transistor Buck-Boost converter

Abstract

This invention discloses a frequency-doubling soft-switching control method and controller for a six-transistor Buck-Boost converter, belonging to the field of power electronics conversion and control. The method includes: for a six-transistor Buck-Boost converter, using two inductor current cycles as one switching cycle, setting the switching frequency of the converter's two-level bridge arm to twice the switching frequency of the three-level bridge arm. A switching timing sequence is designed to achieve soft switching of all switching devices over a wide input voltage and load range. With minimizing inductor current ripple as the goal and soft switching of each switch as a constraint, the phase shift angle and duty cycle of each switch are calculated. This method can achieve zero-voltage switching of all switching devices at a doubled switching frequency of the six-transistor Buck-Boost two-level bridge arm, thereby improving the converter's operating efficiency.

Description

Technical Field

[0001] The present invention belongs to the field of power electronic conversion and control, and more particularly, relates to a frequency doubling soft switching control method and a controller for a six-tube Buck-Boost converter. Background Art

[0002] In common energy storage systems, bidirectional DC / DC converters are key devices for voltage conversion and power transmission. As the voltage levels of energy storage devices increase, higher voltage-bearing requirements are placed on the bidirectional DC / DC energy storage ports. To facilitate switching device selection and reduce switching voltage stress, a three-level bridge arm replaces the two-level bridge arm in the traditional four-transistor Buck-Boost converter, creating a new six-transistor Buck-Boost converter. Compared to midpoint-clamped three-level and cascaded three-level structures, the flying capacitor three-level six-transistor Buck-Boost converter offers the advantages of a common input and output ground and unrestricted switching timing, making it more suitable for high-voltage, high-power applications.

[0003] Currently, research on six-switch buck-boost converters is focused on hard-switching control strategies. To overcome the high switching losses and low device efficiency associated with traditional hard-switching control methods in high-voltage, high-power operating scenarios, zero-voltage switching (ZVS) can be employed to eliminate losses during switch turn-on. Without adding an additional auxiliary soft-switching circuit, soft-switching all switching devices increases the inductor current ripple and effective value, resulting in increased conduction losses, compared to traditional hard-switching methods. Under fixed-frequency control, if the switching frequency in the two-level bridge arm is the same as that in the three-level bridge arm, a larger proportion of circulating current modes will occur under high input voltage and low power conditions. Simultaneously, the inductor current ripple and effective value will be larger, leading to greater conduction losses.

[0004] Therefore, in order to improve the working efficiency and power density of the six-tube Buck-Boost converter within the full operating range, it is of great research significance to realize soft switching control of the entire device, and at the same time reduce the proportion of circulating current mode and reduce the inductor current ripple and effective value under the conditions of large input voltage and low power. Summary of the Invention

[0005] In response to the defects of the existing technology and the need for improvement, the present invention provides a frequency-doubling soft-switching control method and controller for a six-tube Buck-Boost converter, the purpose of which is to achieve soft-switching control of the entire device, and reduce the proportion of circulating current modes and the ripple and effective value of the inductor current when the input voltage is large and the power is low.

[0006] To achieve the above object, according to one aspect of the present invention, a frequency doubling soft switching control method for a six-tube Buck-Boost converter is provided. The six-tube Buck-Boost converter includes a three-level bridge arm, a two-level bridge arm, an inductor, and a flying capacitor; the three-level bridge arm includes a switching tube Q connected in sequence. A1 , Q A2 , Q A3 and Q A4 The two-level bridge arm includes the switch tubes Q connected in sequence B1 and Q B2 ; One end of the inductor is connected to Q A2 and Q A3 The other end is connected to Q B1 and Q B2 The connection point of the flying capacitor is Q A1 and Q A2 The other end is connected to Q A3 and Q A4 The method includes: taking two inductor current cycles as one switching cycle, and in each switching cycle, Q A4 and Q A1 Complementary conduction, Q A3 and Q A2 Complementary conduction, Q B2 and Q B1 Complementary conduction; in the first inductor current cycle of each switching cycle, the starting time Q A1 It turns on and turns off after the first period of conduction, Q A2 Keep it off and start time Q B1 Shutdown, at Q A1 Turn on Q before turning off B1 , in Q A4 The absolute value of the source current is greater than the zero voltage turn-on current and turns off Q B1 ; In the second inductor current cycle of each switching cycle, the starting time Q A2 turns on and turns off after the second period of conduction, Q A1 Keep it off and start time Q B1 Shutdown, at Q A1 Turn on Q before turning off B1 , in Q A4 The absolute value of the source current is greater than the zero voltage turn-on current and turns off Q B1 ; Wherein, the first time period and the second time period are obtained by the following method: taking the minimum pulsation of the inductor current as the goal and the soft switching of each switch tube as the constraint, and calculating them.

[0007] Furthermore, the calculation method of the first time period and the second time period specifically includes: SA1, determining the working mode of the six-transistor Buck-Boost converter, wherein the working mode includes discontinuous mode and continuous mode; SA2, calculating Q according to the working mode of the six-transistor Buck-Boost converter, with the goal of minimizing the pulsation of the inductor current and the soft switching of each switch tube as a constraint. A1 The initial on-duty cycle D A1 and Q A2 The initial on-duty cycle D A2 ;SA3, based on 0.5V in The difference between the voltage of the flying capacitor and the voltage of the flying capacitor is PI controlled. The result of the PI control is limited to obtain the duty cycle compensation value △D, where V in is the input voltage; SA4, use △D to adjust D A1 and D A2 Compensate and get Q accordingly A1 The final on-duty cycle D A1_1 and Q A2 The final on-duty cycle D A2_2 ;SA5, calculate D A1_1 The corresponding first time period, and D A2_2 The corresponding second time period.

[0008] Furthermore, the SA1 specifically includes: if V in >2V out , when I out >I out1 When the six-tube Buck-Boost converter is in continuous mode, I out ≤I out1 In discontinuous mode; if V in ≤2V out , when I out >I out2 When the six-tube Buck-Boost converter is in continuous mode, I out ≤I out2 I out1 and I out2 They are:

[0009]

[0010] Among them, V out is the output voltage, I out is the output current reference value, I out1 is the first boundary current, I out2 is the second boundary current, k is the ratio of output voltage to input voltage, I ZVS is the zero voltage turn-on current, L is the inductance value, Ts is the inductor current period.

[0011] Furthermore, the SA2 specifically includes: according to the working mode of the six-transistor Buck-Boost converter, with the goal of minimizing the pulsation of the inductor current and the soft switching of each switch tube as a constraint, calculating the proportion of each mode in an inductor current cycle; D A1 and D A2 They are:

[0012] D A1 =D1+D2

[0013] D A2 =D1+D2

[0014] Wherein, D1 is the proportion of the first mode in one inductor current cycle, and D2 is the proportion of the second mode in one inductor current cycle. The first mode refers to V A =0.5V in And V B =0 mode, the second mode refers to V A =0.5V in And V B =V out The mode, V A is the output voltage of the three-level bridge arm, V B is the output voltage of the two-level bridge arm, V out is the output voltage.

[0015] Furthermore, the method further comprises: calculating Q B1 Duty cycle D B1 and conduction phase shift angle φ B1 , according to D B1 and Control Q B1 On-off, D B1 and They are:

[0016] D B1 =D2+D3

[0017]

[0018] Wherein, D3 is the proportion of the third mode in one inductor current cycle, and the third mode refers to V A =0 and V B =V out modality.

[0019] Furthermore, V in >2V out And I out >I out1When D1, D2 and D3 are:

[0020]

[0021] V in >2V out And I out ≤I out1 When D1, D2 and D3 are:

[0022]

[0023] V in ≤2V out And I out >I out2 When D1, D2 and D3 are:

[0024]

[0025] V in ≤2V out And I out ≤I out2 When D1, D2 and D3 are:

[0026]

[0027] Where W = 4I zvs 2 L 2 -LT s V in (16k 2 I out +(8k+4)(I out +I zvs ))+2T s 2 V in 2 k, W are intermediate parameters, D3 is the proportion of the third mode in one inductor current cycle, and the third mode refers to V A =0 and V B =V out The modality, I out is the output current reference value, I out1 is the first boundary current, I out2 is the second boundary current, k is the ratio of output voltage to input voltage, I ZVS is the zero voltage turn-on current, L is the inductance value, T s is the inductor current period.

[0028] Furthermore, D A1_1 =D A1 +△D,D A2_2 =DA2 -△D.

[0029] Furthermore, the control switch Q A1 , Q A2 , Q A3 , Q A4 , Q B1 and Q B2 During the on-off process, a dead time is set.

[0030] According to another aspect of the present invention, a frequency doubling soft switching controller for a six-transistor Buck-Boost converter is provided, which is used to execute the frequency doubling soft switching control method for a six-transistor Buck-Boost converter as described above.

[0031] In general, the above technical solutions conceived by the present invention can achieve the following beneficial effects:

[0032] (1) A frequency doubling soft switching control method for a six-tube Buck-Boost converter is provided. For the six-tube Buck-Boost converter, the method takes the realization of zero voltage conduction of each switch tube as a constraint, utilizes the degree of freedom of the duty cycle and phase shift angle of the switch device, optimizes the inductor current ripple to minimize it, and controls the on and off of each switch tube according to the method. It can reduce the inductor current ripple while achieving full-range soft switching, effectively reduce the conduction loss of the inductor and the switch device, reduce the inductor volume, and improve the power density;

[0033] (2) A specific calculation method for the duty cycle of a driving signal is provided. The input voltage and output power reference values ​​are used to determine the converter working mode. The input voltage, output power and working area signal are used to calculate the duty cycle and phase shift angle of each switch tube. The switch duty cycle is compensated through the flying capacitor voltage closed-loop control to achieve flying capacitor voltage following. According to the compensated switch tube duty cycle and phase shift angle, a carrier and a modulation wave are generated to control the on and off of each switch tube, thereby improving the working efficiency of the converter. BRIEF DESCRIPTION OF THE DRAWINGS

[0034] Figure 1 A topological diagram of a six-transistor Buck-Boost converter provided in an embodiment of the present invention;

[0035] Figure 2 A control block diagram of a six-transistor Buck-Boost converter provided in an embodiment of the present invention;

[0036] Figure 3 A schematic diagram of the division of the continuous mode and the intermittent mode provided in an embodiment of the present invention;

[0037] Figure 4A switching timing diagram of a switching cycle provided by an embodiment of the present invention;

[0038] Figure 5 A switching timing diagram of an inductor current cycle provided by an embodiment of the present invention;

[0039] Figure 6 The control method provided by the embodiment of the present invention is V in >2V out Schematic diagram of inductor current change when load size changes;

[0040] Figure 7 The control method provided by the embodiment of the present invention is V in ≤2V out Schematic diagram of inductor current change when load size changes;

[0041] Figure 8 When the control method provided by the embodiment of the present invention is adopted, V in >2V out Working waveform when working in discontinuous mode;

[0042] Figure 9 When the control method provided by the embodiment of the present invention is adopted, V in >2V out Working waveform when working in continuous mode;

[0043] Figure 10 When the control method provided by the embodiment of the present invention is adopted, V in ≤2V out Working waveform when working in discontinuous mode;

[0044] Figure 11 When the control method provided by the embodiment of the present invention is adopted, V in ≤2V out Working waveform when working in continuous mode. DETAILED DESCRIPTION

[0045] In order to make the objectives, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely for the purpose of explaining the present invention and are not intended to limit the present invention. In addition, the technical features involved in the various embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.

[0046] In the present invention, the terms "first", "second", etc. (if any) in the present invention and the drawings are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence.

[0047] Example 1

[0048] A frequency-doubling soft-switching control method for a six-transistor Buck-Boost converter, see Figures 1-11 , the frequency doubling soft switching control method for the six-transistor Buck-Boost converter in this embodiment is described in detail.

[0049] The six-tube Buck-Boost converter includes a three-level bridge arm, a two-level bridge arm, an inductor and a flying capacitor. The three-level bridge arm includes a switch tube Q connected in sequence. A1 , Q A2 , Q A3 and Q A4 The two-level bridge arm includes a switch tube Q connected in sequence B1 and Q B2 One end of the inductor is connected to Q A2 and Q A3 The other end is connected to Q B1 and Q B2 One end of the flying capacitor is connected to Q A1 and Q A2 The other end is connected to Q A3 and Q A4 connection point.

[0050] See Figure 1 The three-level bridge arm includes the upper bridge arm outer tube Q A1 、Upper arm inner tube Q A2 、Lower bridge arm inner tube Q A3 、Lower bridge arm outer tube Q A4 And the flying capacitor C fly Composition, among which C fly One end and Q A1 Source, Q A2 The drain is connected to the node, and the other end is connected to the Q A3 Source, Q A4 The node connected to the drain is connected, and the output port is Q A2 Source and Q A3 The connection point of the drain. The two-level bridge arm consists of the upper bridge arm switch tube Q B1 With the lower bridge arm switch tube Q B2 The output port is Q B1 With Q B2 The two ends of the inductor L are connected to the output port of the three-level bridge arm and the output port of the two-level bridge arm respectively. Output filter capacitor C o The two ends are connected to the switch tube Q B1 The drain is connected to ground. Figure 2 The control block diagram shown implements this method.

[0051] The frequency doubling soft switching control method for the six-transistor Buck-Boost converter is as follows. Two inductor current cycles constitute one switching cycle. When operating in steady state, the inductor currents in the two inductor current cycles are exactly the same. In each switching cycle, Q A4 and Q A1 Complementary conduction, Q A3 and Q A2 Complementary conduction, Q B2 and Q B1 Complementary conduction. In the first inductor current cycle of each switching cycle, the starting time Q A1 It turns on and turns off after the first period of conduction, Q A2 Keep it off and start time Q B1 Shutdown, at Q A1 Turn on Q before turning off B1 , in Q A4 The absolute value of the source current is greater than the zero voltage turn-on current and turns off Q B1 , Q B1 The turn-off time lags behind Q A1 Off and ahead of Q A3 During the second inductor current cycle of each switching cycle, the starting time Q A2 turns on and turns off after the second period of conduction, Q A1 Keep it off and start time Q B1 Shutdown, at Q A1 Turn on Q before turning off B1 , in Q A4 The absolute value of the source current is greater than the zero voltage turn-on current and turns off Q B1 , Q B1 The turn-off time lags behind Q A2 Off and ahead of Q A4 The first time period and the second time period are obtained by the following method: taking the minimum pulsation of the inductor current as the goal and the soft switching of each switch tube as the constraint, the timing of each switch tube in a switching cycle is as follows: Figure 4 shown.

[0052] Preferably, the calculation method of the first time period and the second time period specifically includes the following operations SA1 to SA5.

[0053] Operation SA1 is performed to determine the operating mode of the six-transistor Buck-Boost converter. The operating modes include discontinuous mode and continuous mode.

[0054] See Figure 2 , sample output voltage V out With the reference output voltage V out_ref After comparison, the output voltage error value is obtained, which is input into the PI controller PI1, and the power reference value P is output through PI1.ref According to P ref Calculate the output current reference value I out =P ref / V out .

[0055] V in >2V out When the load changes, the inductor current is as follows: Figure 6 As shown, V in ≤2V out When the load changes, the inductor current is as follows: Figure 7 See Figure 6 and Figure 7 , it can be seen that there is Q in the discontinuous mode A3 , Q A4 With Q B2 The switching mode of simultaneous conduction does not exist in continuous mode. Figure 3 The specific method for determining the working mode of the six-tube Buck-Boost converter is as follows: If V in >2V out , when I out >I out1 When the six-tube Buck-Boost converter is in continuous mode, I out ≤I out1 In discontinuous mode; if V in ≤2V out , when I out >I out2 When the six-tube Buck-Boost converter is in continuous mode, I out ≤I out2 is in intermittent mode.

[0056] I out1 and I out2 They are:

[0057]

[0058]

[0059] Among them, V out is the output voltage, I out is the output current reference value, I out1 is the first boundary current, I out2 is the second boundary current, k is the ratio of output voltage to input voltage, I ZVS is the zero voltage turn-on current, L is the inductance value, T s is the inductor current period.

[0060] Operate SA2, according to the working mode of the six-transistor Buck-Boost converter, with the goal of minimizing the pulsation of the inductor current and the soft switching of each switch tube as the constraint, calculate Q A1 The initial on-duty cycle D A1 and Q A2 The initial on-duty cycle D A2 .

[0061] According to an embodiment of the present invention, operation SA2 specifically includes: calculating the proportion of each mode within an inductor current cycle based on the operating mode of the six-transistor Buck-Boost converter, with the goal of minimizing the pulsation of the inductor current and the soft switching of each switch tube as a constraint; A1 and D A2 They are:

[0062] D A1 =D1+D2

[0063] D A2 =D1+D2

[0064] Among them, D1 is the proportion of the first mode in one inductor current cycle, and D2 is the proportion of the second mode in one inductor current cycle. The first mode refers to V A =0.5V in And V B = 0 mode, the second mode refers to V A =0.5V in And V B =V out The mode, V A is the output voltage of the three-level bridge arm, V B is the output voltage of the two-level bridge arm, V out is the output voltage.

[0065] See Figure 5 In one inductor current cycle, the converter has the following four modes: V A =0.5V in And V B =0 first mode, V A =0.5V in And V B =V out The second mode, V A =0 and V B =V out The third mode, V A =0 and V B = 0. The proportion of each mode is calculated as follows.

[0066] V in >2V out And Iout >I out1 When D1, D2, D3 and D4 are:

[0067]

[0068] D4=0

[0069] V in >2V out And I out ≤I out1 When D1, D2, D3 and D4 are:

[0070]

[0071] V in ≤2V out And I out >I out2 When D1, D2, D3 and D4 are:

[0072]

[0073] D4=0

[0074] V in ≤2V out And I out ≤I out2 When D1, D2, D3 and D4 are:

[0075]

[0076] Where W = 4I zvs 2 L 2 -LT s V in (16k 2 I out +(8k+4)(I out +I zvs ))+2T s 2 V in 2 k, W are intermediate parameters, D3 is the proportion of the third mode in one inductor current cycle, and the third mode refers to V A =0 and V B =V out The modality, I out is the output current reference value, I out1 is the first boundary current, I out2 is the second boundary current, k is the ratio of output voltage to input voltage, I ZVS is the zero voltage turn-on current, L is the inductance value, T sis the inductor current cycle, D3 is the proportion of the third mode in one inductor current cycle, and the third mode refers to V A =0 and V B =V out mode, and D4 is the proportion of the fourth mode in one inductor current cycle.

[0077] Preferably, the method further comprises: calculating Q B1 Duty cycle D B1 and conduction phase shift angle According to D B1 and Control Q B1 On-off, D B1 and They are:

[0078] D B1 =D2+D3

[0079]

[0080] Operate SA3, according to 0.5V in The difference between the voltage of the flying capacitor and the voltage of the flying capacitor is PI controlled. The result of the PI control is limited to obtain the duty cycle compensation value △D, where V in is the input voltage.

[0081] Operation SA4, use △D to adjust D A1 and D A2 Compensate and get Q accordingly A1 The final on-duty cycle D A1_1 and Q A2 The final on-duty cycle D A2_2 .

[0082] D A1_1 =D A1 +△D

[0083] D A2_2 =D A2 -△D

[0084] Operation SA5, calculate D A1_1 The corresponding first time period, and D A2_2 The corresponding second time period. Specifically, the first time period is equal to D A1_1 The product of the inductor current period, the second time period is equal to D A2_2 The product of the inductor current period.

[0085] Preferably, in this embodiment, the control switch tube Q A1 , Q A2 , Q A3 , Q A4 , QB1 and Q B2 In the switching process, a dead time is set. Specifically, the PWM duty cycle and phase shift angle of each switch tube are modulated, and the dead time is set to obtain the driving signal of each switch tube.

[0086] Under steady-state conditions, in two adjacent cycles, Q in the latter inductor current cycle is A1 , Q A4 Respectively with Q in the previous inductor current cycle A2 , Q A3 The driving signals are exactly the same; the Q A2 , Q A3 The drive signal is respectively the same as the Q in the previous inductor current cycle A1 , Q A4 The drive signals are exactly the same.

[0087] by Figure 1 、 Figure 2 The main parameters in the test example shown are set as follows to verify the effectiveness of this method: Rated capacity P N =6.6kW, the switching frequency of the three-level bridge arm is 100kHz, the switching frequency of the two-level bridge arm is 200kHz, and the minimum DC voltage on the input side is V inmin =500V, maximum input DC voltage V inmax =2000V, output voltage value V out =750V, flying capacitor value C fly =120μF, inductance value L = 75μH, output filter capacitor capacitance value C f =80μF, DC input capacitor value C in =200μF, output capacitance of the switching device C oss =200pF, zero voltage turn-on current I zvs =2A, dead time t dead =200ns.

[0088] With the above parameter settings, the six-tube Buck-Boost converter operates at V in >2V out When working in discontinuous mode and continuous mode, the key waveforms are as follows: Figure 8 、 Figure 9 As shown; working at V in ≤2V out When working in discontinuous mode and continuous mode, the key waveforms are as follows: Figure 10 、 Figure 11 See Figures 8-11It can be seen that this method can effectively achieve zero-voltage turn-on of all switching devices, verifying the effectiveness of this method. This method can reduce switching losses while reducing inductor conduction losses, thereby improving the operating efficiency and power density of the six-transistor Buck-Boost converter.

[0089] Example 2

[0090] A frequency doubling soft switching controller for a six-transistor Buck-Boost converter is used to implement the above-mentioned frequency doubling soft switching control method for a six-transistor Buck-Boost converter. The related technical solutions are the same as those in the first embodiment and will not be repeated here.

[0091] It will be easily understood by those skilled in the art that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A frequency doubling soft switching control method for a six-transistor Buck-Boost converter, wherein the six-transistor Buck-Boost converter comprises a three-level bridge arm, a two-level bridge arm, an inductor, and a flying capacitor; the three-level bridge arm comprises a switching tube Q connected in sequence; A1 , Q A2 , Q A3 and Q A4 The two-level bridge arm includes the switch tubes Q connected in sequence B1 and Q B2 ; One end of the inductor is connected to Q A2 and Q A3 The other end is connected to Q B1 and Q B2 The connection point of the flying capacitor is Q A1 and Q A2 The other end is connected to Q A3 and Q A4 connection point; characterized in that, Methods include: Two inductor current cycles are considered as one switching cycle. In each switching cycle, Q A4 and Q A1 Complementary conduction, Q A3 and Q A2 Complementary conduction, Q B2 and Q B1 complementary conduction; In the first inductor current cycle of each switching cycle, the starting time Q A1 It turns on and turns off after the first period of conduction, Q A2 Keep it off and start time Q B1 Shutdown, at Q A1 Turn on Q before turning off B1 , in Q A4 The absolute value of the source current is greater than the zero voltage turn-on current and turns off Q B1 ; In the second inductor current cycle of each switching cycle, the starting time Q A2 turns on and turns off after the second period of conduction, Q A1 Keep it off and start time Q B1 Shutdown, at Q A1 Turn on Q before turning off B1 , in Q A4 The absolute value of the source current is greater than the zero voltage turn-on current and turns off Q B1 ; The first time period and the second time period are obtained by calculation with the goal of minimizing the pulsation of the inductor current and the soft switching of each switch tube as a constraint.

2. The frequency doubling soft switching control method for a six-transistor Buck-Boost converter according to claim 1, wherein: The calculation method of the first time period and the second time period specifically includes: SA1, determining the operating mode of the six-transistor Buck-Boost converter, wherein the operating mode includes a discontinuous mode and a continuous mode; SA2, based on the operating mode of the six-transistor Buck-Boost converter, with the goal of minimizing the pulsation of the inductor current and the soft switching of each switch tube as the constraint, calculate Q A1 The initial on-duty cycle D A1 and Q A2 The initial on-duty cycle D A2 ; SA3, based on 0.5V in The difference between the voltage of the flying capacitor and the voltage of the flying capacitor is PI controlled. The result of the PI control is limited to obtain the duty cycle compensation value △D, where V in is the input voltage; SA4, use △D to calculate D A1 and D A2 Compensate and get Q accordingly A1 The final on-duty cycle D A1_1 and Q A2 The final on-duty cycle D A2_2 ; SA5, calculate D A1_1 The corresponding first time period, and D A2_2 The corresponding second time period.

3. The frequency doubling soft switching control method for a six-transistor Buck-Boost converter according to claim 2, wherein: The SA1 specifically includes: If V in >2V out , when I out >I out1 When the six-tube Buck-Boost converter is in continuous mode, I out ≤I out1 In discontinuous mode; if V in ≤2V out , when I out >I out2 When the six-tube Buck-Boost converter is in continuous mode, I out ≤I out2 When in intermittent mode; I out1 and I out2 They are: Among them, V out is the output voltage, I out is the output current reference value, I out1 is the first boundary current, I out2 is the second boundary current, k is the ratio of output voltage to input voltage, I ZVS is the zero voltage turn-on current, L is the inductance value, T s is the inductor current period.

4. The frequency doubling soft switching control method for a six-transistor Buck-Boost converter according to claim 2, wherein: The SA2 specifically includes: Based on the operating mode of the six-transistor Buck-Boost converter, with the goal of minimizing the inductor current ripple and the soft switching of each switch as a constraint, the proportion of each mode within an inductor current cycle is calculated. D A1 and D A2 They are: D A1 =D1+D2 D A2 =D1+D2 Wherein, D1 is the proportion of the first mode in one inductor current cycle, and D2 is the proportion of the second mode in one inductor current cycle. The first mode refers to V A =0.5V in And V B =0 mode, the second mode refers to V A =0.5V in And V B =V out The mode, V A is the output voltage of the three-level bridge arm, V B is the output voltage of the two-level bridge arm, V out is the output voltage.

5. The frequency doubling soft switching control method for a six-transistor Buck-Boost converter according to claim 4, wherein: The method further includes: calculating Q B1 Duty cycle D B1 and conduction phase shift angle According to D B1 and Control Q B1 On-off, D B1 and They are: D B1 =D2+D3 Wherein, D3 is the proportion of the third mode in one inductor current cycle, and the third mode refers to V A =0 and V B =V out modality.

6. The frequency doubling soft switching control method for a six-transistor Buck-Boost converter according to claim 4 or 5, characterized in that: V in >2V out And I out >I out1 When D1, D2 and D3 are: V in >2V out And I out ≤I out1 When D1, D2 and D3 are: V in ≤2V out And I out >I out2 When D1, D2 and D3 are: V in ≤2V out And I out ≤I out2 When D1, D2 and D3 are: Where W = 4I zvs 2 L 2 -LT s V in (16k 2 I out +(8k+4)(I out +I zvs ))+2T s 2 V in 2 k, W are intermediate parameters, D3 is the proportion of the third mode in one inductor current cycle, and the third mode refers to V A =0 and V B =V out The modality, I out is the output current reference value, I out1 is the first boundary current, I out2 is the second boundary current, k is the ratio of output voltage to input voltage, I ZVS is the zero voltage turn-on current, L is the inductance value, T s is the inductor current period.

7. The frequency doubling soft switching control method for a six-transistor Buck-Boost converter according to claim 2, wherein: D A1_1 =D A1 +△D,D A2_2 =D A2 -△D。 8. The frequency doubling soft switching control method for a six-transistor Buck-Boost converter according to claim 1, wherein: Control switch tube Q A1 , Q A2 , Q A3 , Q A4 , Q B1 and Q B2 During the on-off process, a dead time is set.

9. A frequency-doubling soft-switching controller for a six-transistor Buck-Boost converter, characterized in that: Used to execute the frequency doubling soft switching control method for a six-transistor Buck-Boost converter as described in any one of claims 1 to 8.

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