A LPST air gap magnetic field modeling method and inductance parameter acquisition method

Abstract

The present invention relates to the field of transformers, and more particularly to a method for modeling the air gap magnetic field of an LPST (linear phase-shifting transformer) and obtaining its inductance parameters. The method comprises: analyzing the relationship between the LPST primary and secondary sides and the air gap to construct an analytical air gap magnetic field model that accounts for ferromagnetic material saturation; analyzing the relationship between the LPST slots and the air gap to construct a correction air gap magnetic field model that accounts for the slot effect; and analyzing the relationship between the LPST edges and the air gap to construct a correction air gap magnetic field model that accounts for core disconnection. The LPST air gap magnetic field model constructed by the present invention simultaneously accounts for the effects of ferromagnetic material saturation, the slot effect, and the core disconnection. Based on the LPST air gap magnetic field model of the present invention, the calculation of various inductance parameters is highly accurate, has a small computational dimension, and is quick to calculate.

Description

Technical Field

[0001] The present invention belongs to the field of transformers, and in particular relates to an air gap magnetic field modeling method of an LPST (linear phase-shifting transformer) and a method for obtaining inductance parameters thereof. Background Art

[0002] like Figure 1 As shown in Figure 1, the linear phase-shifting transformer (LPST) is a novel phase-shifting transformer designed based on the structure and operating principles of linear motors. It is a key component of multiple superposition inverter systems. The LPST has essentially the same structure as a linear motor, differing in that the primary and secondary cores of the LPST are of equal length and symmetrical about the air gap. Unlike conventional phase-shifting transformers, the LPST primarily achieves energy conversion through a traveling wave magnetic field across the air gap. When the primary winding of the LPST is energized, a traveling wave magnetic field is generated within the core, inducing three-phase AC power on the secondary side.

[0003] Compared with phase-shifting transformers of other structures, LPST has the advantages of a wide phase-shifting angle range, no need for a special winding turns ratio, high core and winding utilization, and the ability to achieve electrical isolation. When used in rectifier / inverter equipment, it can effectively eliminate low-order harmonics, improve output waveform quality, and reduce grid harmonic pollution. It can be used in microgrids such as ship integrated power systems and electric vehicles.

[0004] The air gap magnetic field is the medium for LPST energy conversion, and the accuracy of its calculation directly impacts the calculation of electromagnetic characteristics such as transformer impedance, efficiency, losses, and vibration noise. However, interrupting the transformer core leads to three-phase impedance asymmetry and generates end-face magnetic flux at the interruption point, causing air gap magnetic field distortion. Furthermore, the large number of slots on the primary and secondary sides of the transformer increases the harmonic content of the air gap magnetic field due to the cogging effect, complicating the accurate calculation of the air gap magnetic flux density. Due to the special operating conditions of ships, which require transformers to be as lightweight and compact as possible, the air gap magnetic flux density is relatively high, and the influence of ferromagnetic material saturation on the transformer air gap magnetic field cannot be ignored.

[0005] The existing LPST air gap magnetic field model still has the following shortcomings: First, it cannot simultaneously consider the influence of multiple factors such as ferromagnetic material saturation, cogging effect and core breaking; Second, the model is relatively complex, the calculation process is cumbersome, and the amount of calculation is large. Summary of the Invention

[0006] The purpose of the present invention is to solve the technical problems existing in the background technology. To this end, a LPST air gap magnetic field modeling method and an inductance parameter acquisition method thereof are provided.

[0007] In order to achieve the above object, the technical solution adopted by the present invention is as follows:

[0008] A method for modeling an LPST air gap magnetic field comprises the following steps:

[0009] Step S1: Analyze the relationship between the primary and secondary sides of the LPST and the air gap, and build an analytical model of the air gap magnetic field considering the saturation of ferromagnetic materials;

[0010] Step S101: Divide the LPST into N blocks at equal intervals along the length of the air gap, primary tooth, primary yoke, secondary tooth, and secondary yoke. Within the LPST length, divide the air gap centerline into N blocks at equal intervals, with a total of N+1 nodes.

[0011] Obtain the total magnetic voltage drop of the circuit, the normal magnetic voltage drop of the air gap at the mth and m+1th nodes, the longitudinal magnetic voltage drop of the yoke at the mth node of the primary core, the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the primary core, the longitudinal magnetic voltage drop of the yoke at the mth node of the secondary core, and the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the secondary core;

[0012] Step S102: fitting the BH curve of the nonlinear ferromagnetic material to correct the actual normal flux density of the tooth when the ferromagnetic material is saturated;

[0013] Step S103: According to the flux continuity theorem, the yoke transverse magnetic flux at the mth node is equal to the total longitudinal magnetic flux of the air gap from the first node to the mth node, and the yoke magnetic flux is obtained;

[0014] Step S104: Simplify the magnetomotive force of each node in the air gap based on the distribution characteristics of the LPST winding, and obtain the air gap flux density at the mth node;

[0015] Step S105: Based on the simplified magnetomotive force of each node in the air gap, iterative judgment is performed until the judgment condition is met. If the condition is met, the air gap magnetic flux density is output; if the condition is not met, the air gap magnetic flux density is corrected;

[0016] Step S2: Analyze the relationship between the LPST cogging and the air gap, and build an air gap magnetic field correction model that takes the cogging effect into account;

[0017] Step S201: determining the relative magnetic permeability of the air gap considering the influence of the cogging effect;

[0018] Step S3: Analyze the relationship between the LPST edge and the air gap, and build an air gap magnetic field correction model considering the core breaking;

[0019] Step S301: simplify the edge model and regularize the magnetic field distribution through Hook transformation;

[0020] Step S302: taking into account the end flux density of the core being broken;

[0021] Step S303: using the flux continuity theorem, introducing the core breaking flux density change value to represent the lost flux density;

[0022] Step S4: The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of the slot and core disconnection factors.

[0023] The following is a technical solution further defined by the modeling method of the present invention: the primary and secondary sides of the LPST are symmetrical about the center line of the air gap.

[0024] The following is a technical solution further defined by the modeling method of the present invention, where the total magnetic voltage drop of the loop is:

[0025] F ∑ (m) = F δ_n (m+1)+F t1_n (m+1)-F δ_n (m)-F j1_l (m)-F t1_n (m)

[0026] Where, F δ_n (m), F δ_n (m+1) is the normal magnetic pressure drop of the air gap at the mth and m+1th nodes, F j1_l (m) is the longitudinal magnetic voltage drop of the yoke at the mth node of the primary core, F t1_n (m), F t1_n (m+1) is the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the primary core. The positive direction of the coordinate axis is the positive direction of the magnetic voltage drop of each segment. According to Ampere's loop theorem:

[0027]

[0028] Where H δ_n is the magnetic field strength along the normal direction of the air gap; B t1_n 、H t1_n is the normal magnetic flux density and magnetic flux density of the virtual tooth of the primary core; B j1_l 、H j1_l is the longitudinal magnetic flux density and magnetic flux density of the primary core yoke; δ is the air gap length; h t1 is the virtual tooth height of the primary core; μ0 is the magnetic permeability of air; μ j1 is the magnetic permeability of each node of the primary core yoke.

[0029] The following is a technical solution further defined by the modeling method of the present invention, where the actual normal magnetic flux density of the tooth is:

[0030] B t_n (m) = B t_n′(m)-μ0H t_n (m)k δ

[0031] Where B′ t_n (m) is the apparent normal flux density of the tooth at the mth node, which means the flux density when all the magnetic flux enters the tooth; B t_n (m) is the actual normal flux density of the tooth at the mth node; H t_n (m) is the actual normal magnetic field intensity of the tooth at the mth node; k δ is the slot coefficient, k δ =(H·b s ) / (K Fe ·L·b t ).

[0032] The following is a technical solution further defined by the modeling method in the present invention, where the yoke magnetic flux density is:

[0033]

[0034] The following is a technical solution further defined by the modeling method of the present invention, which simplifies the magnetomotive force of each node in the air gap to:

[0035]

[0036] Where,

[0037]

[0038] The air gap magnetic flux density at the mth node is:

[0039]

[0040] Where K B It is a saturation coefficient preset according to the saturation degree of the primary side core.

[0041] The following is a technical solution further limited by the modeling method in the present invention, and the judgment conditions are:

[0042]

[0043] If the judgment conditions are not met, the air gap magnetic flux density is corrected:

[0044]

[0045] Where k s is the iteration coefficient.

[0046] The following is a technical solution further defined by the modeling method of the present invention, wherein the relative magnetic permeability of the air gap is:

[0047]

[0048] Where t1 is the tooth pitch and T is the extension period, then:

[0049]

[0050] The nonlinear function β(y) is:

[0051]

[0052] Wherein, the expression of v is obtained by software fitting.

[0053] The following is a technical solution further defined by the modeling method of the present invention, taking into account the end magnetic flux density of the broken core:

[0054]

[0055] Using the flux continuity theorem, the core breaking flux change value Q is introduced to represent the lost flux density:

[0056]

[0057] The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of the tooth slot and core disconnection factors:

[0058] B _x (x) = (1-Q)B δ_sl_n (x)·λ YB_c_n ·λ FB_c_n

[0059] Where B δ0_sl_n (x) is the expression of the air gap flux density considering the saturation of ferromagnetic materials; λ YB_c_n ,λ FB_c_n These are the relative permeability functions of the air gap when slots are opened on the primary and secondary sides, respectively.

[0060] A method for obtaining inductance parameters, which obtains inductance parameters by using a model built using the above-mentioned LPST air gap magnetic field modeling method, includes:

[0061] In the LPST effective region, the energy stored in the air gap magnetic field is:

[0062]

[0063] Solve the self-inductance and mutual inductance parameters of each LPST winding, including:

[0064]

[0065]

[0066] When solving the self-inductance of the i-th winding, a small current perturbation ΔI is made to the i-th winding. i , the other winding currents remain unchanged, and the magnetic field energy W(I i +ΔI i ) and W(I i -ΔI i ), solve for self-inductance When solving the mutual inductance of the i-th winding and the j-th winding, a small current perturbation ΔI is made to the i-th and j-th windings respectively. i , ΔI j , we get the magnetic field energy W(I i +ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j ) and W(I i -ΔI i ,I j -ΔI j ), solve for self-inductance

[0067] Compared with the prior art, the present invention has the following technical effects:

[0068] The LPST air-gap magnetic field model constructed in the present invention can simultaneously consider the effects of ferromagnetic material saturation, cogging, and core disconnection. Based on the LPST air-gap magnetic field model of the present invention, the calculation accuracy of various inductance parameters is high, the calculation dimension is small, and the calculation time is short.

[0069] The present invention will be further described below with reference to the accompanying drawings and embodiments. BRIEF DESCRIPTION OF THE DRAWINGS

[0070] In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the following briefly introduces the drawings required for use in the embodiments or the prior art. Obviously, the drawings described below are only some embodiments of the present invention. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying any creative work.

[0071] Figure 1 It is a structural diagram of a linear phase-shifting transformer;

[0072] Figure 2 It is a simplified diagram of the model of the present invention;

[0073] Figure 3 is an iterative flow chart of the present invention;

[0074] Figure 4 It is the end Hook transformation analysis diagram of the present invention;

[0075] Figure 5 1 is a comparison diagram of the air gap magnetic field model of the present invention and the finite element simulation results when the DC bus voltage of LPST is 100V;

[0076] Figure 6 1 is a comparison chart of the calculated values ​​of each winding of the LPST of the present invention and the finite element simulation values ​​when the DC bus voltage of the LPST is 100V;

[0077] Figure 7 This is a schematic diagram of the LPST winding structure. DETAILED DESCRIPTION

[0078] To make the above-mentioned objects, features, and advantages of the present invention more readily apparent, specific embodiments of the present invention are described in detail below with reference to the accompanying drawings. The following description sets forth numerous specific details to facilitate a full understanding of the present invention. However, the present invention can be implemented in many other ways than those described herein, and those skilled in the art may make similar modifications without departing from the scope of the present invention. Therefore, the present invention is not limited to the specific embodiments disclosed below.

[0079] like Figure 2-7 As shown, this embodiment provides a LPST air gap magnetic field modeling method and an inductance parameter acquisition method thereof.

[0080] A LPST air gap magnetic field modeling method is used to build an LPST air gap magnetic field model based on electromagnetic field theory. The LPST air gap magnetic field modeling method includes:

[0081] Step 1: Analyze the relationship between the primary and secondary sides of the LPST and the air gap, and build an analytical model of the air gap magnetic field considering the saturation of ferromagnetic materials.

[0082] Step 2: Analyze the relationship between the LPST cogging and the air gap, and build an air gap magnetic field correction model that takes the cogging effect into account.

[0083] Step 3: Analyze the relationship between the LPST edge and the air gap, and build an air gap magnetic field correction model considering the core breaking.

[0084] To further elaborate on step one, the specific process is as follows:

[0085] (1) The five regions along the length of the LPST (air gap (I), primary side teeth (II), primary side yoke (III), secondary side teeth (IV), and secondary side yoke (V)) are divided into blocks at equal intervals. Within the length of the LPST, the air gap centerline is divided into N blocks at equal intervals, with a total of N+1 nodes. Since the primary and secondary sides of the LPST are symmetrical about the air gap centerline, only one side of the core is analyzed. The model is simplified as follows: Figure 1 shown.

[0086] Figure 1 In the figure, 1 represents the air gap, 2 represents the primary side tooth part, and 3 represents the primary side yoke part.

[0087] At this time, the total magnetic voltage drop of the circuit is:

[0088] F ∑ (m) = F δ_n (m+1)+F t1_n (m+1)-F δ_n (m)-F j1_l (m)-F t1_n (m) (1)

[0089] Where, F δ_n (m), F δ_n (m+1) is the normal magnetic pressure drop of the air gap at the mth and m+1th nodes, F j1_l (m) is the longitudinal magnetic voltage drop of the yoke at the mth node of the primary core, F t1_n (m), F t1_n (m+1) is the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the primary core, and the positive direction of the coordinate axis is the positive direction of the magnetic voltage drop of each segment.

[0090] According to Ampere's circuit theorem:

[0091]

[0092] Where H δ_n is the magnetic field strength along the normal direction of the air gap; B t1_n 、H t1_n is the normal magnetic flux density and magnetic flux density of the virtual tooth of the primary core; B j1_l 、H j1_l is the longitudinal magnetic flux density and magnetic flux density of the primary core yoke; δ is the air gap length; h t1 is the virtual tooth height of the primary core; μ0 is the magnetic permeability of air; μ j1 is the magnetic permeability of each node of the primary core yoke.

[0093] (2) The BH curve of the nonlinear ferromagnetic material (DW465-50) is fitted to correct the actual magnetic flux expression of the tooth when the ferromagnetic saturation occurs, as shown in formula (3):

[0094] B t_n (m) = B t_n ′(m)-μ0H t_n (m)k δ (3)

[0095] Where B′ t_n (m) is the apparent normal flux density of the tooth at the mth node, which means the flux density when all the magnetic flux enters the tooth; B t_n (m) is the actual normal flux density of the tooth at the mth node; H t_n (m) is the actual normal magnetic field intensity of the tooth at the mth node; k δ is the slot coefficient, k δ =(H·b s ) / (K Fe ·L·b t ).

[0096] (3) According to the flux continuity theorem, the transverse magnetic flux density of the yoke at the mth node is equal to the total longitudinal magnetic flux density of the air gap from the first node to the mth node. The yoke magnetic flux density is calculated as:

[0097]

[0098] (4) Combined with the distribution characteristics of the LPST winding, the magnetomotive force of each node in the air gap is simplified to formula (5):

[0099]

[0100] Where,

[0101]

[0102] The air gap magnetic flux density at the mth node is:

[0103]

[0104] Where K B It is a saturation coefficient preset according to the saturation degree of the primary side core.

[0105] (5) According to the above solution process, it is iterated until the judgment condition is met. The specific judgment condition is as follows:

[0106]

[0107] If the judgment conditions are not met, the air gap magnetic flux density is corrected:

[0108]

[0109] Where k s is the iteration coefficient.

[0110] Perform multiple iterations until the judgment conditions are met, at which point the air gap magnetic flux density distribution within the LPST length range can be obtained. The specific iteration flow chart is as follows: Figure 3 The specific iterative process includes:

[0111] Set the DC bus voltage;

[0112] Calculate the effective value of primary side current, air gap magnetomotive force, and initial air gap magnetic flux density;

[0113] Selection and calculation: Calculate the primary side yoke magnetic flux density and the primary side tooth magnetic flux density; correct the tooth magnetic flux density; calculate the total magnetic voltage drop of each node in the circuit;

[0114] Determine whether the conditions are met;

[0115] If the conditions are met, the air gap magnetic flux density is output;

[0116] If the conditions are not met, reselect the magnetic flux density of each node in the air gap and then perform the selection and calculation again.

[0117] To further elaborate on step 2, the specific process is as follows:

[0118] Determine the relative permeability of the air gap (per unit value) considering the influence of the slot effect:

[0119]

[0120] Where t1 is the tooth pitch, T is the extension period (extended with the tooth pitch as the period), so:

[0121]

[0122] The nonlinear function β(y) is:

[0123]

[0124] Wherein, the expression of v is obtained by software fitting.

[0125] To further elaborate on step three, the specific process is as follows:

[0126] Simplify the edge model and regularize the magnetic field distribution through Hook transformation, such as Figure 4 shown.

[0127] The end magnetic flux density taking into account the core breaking is:

[0128]

[0129] Using the flux continuity theorem, the core breaking flux change value Q is introduced to represent the lost flux density:

[0130]

[0131] The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of the tooth slot and core disconnection factors:

[0132] B _x (x) = (1-Q)B δ_sl_n (x)·λ YB_c_n ·λ FB_c_n (14)

[0133] Where B δ0_sl_n (x) is the air gap flux density expression considering the saturation of ferromagnetic materials. YB_c_n ,λ FB_c_n These are the relative permeability functions of the air gap when slots are opened on the primary and secondary sides, respectively.

[0134] Based on the above model construction method, an LPST air gap magnetic field model based on electromagnetic field theory is constructed to obtain the main inductance parameters of the LPST based on the air gap magnetic field distribution.

[0135] (1) Modeling the inductance of each LPST winding using the above analysis and based on the energy perturbation method;

[0136] (2) Obtain the inductance parameters of each winding based on the LPST air gap magnetic field distribution;

[0137] Furthermore, in the LPST effective region, the energy storage of the air gap magnetic field is given by Equation (15), where the magnetic flux density B _x (m) From the above analysis, we can see that:

[0138]

[0139] Furthermore, the self-inductance and mutual inductance parameters of each LPST winding are solved, including:

[0140]

[0141] When solving the self-inductance of the i-th winding, a small current perturbation ΔI is made to the i-th winding. i , the other winding currents remain unchanged, and the magnetic field energy W(I i +ΔI i ) and W(I i -ΔI i ), and then solve the self-inductance according to formula (16) When solving the mutual inductance of the i-th winding and the j-th winding, a small current perturbation ΔI is made to the i-th and j-th windings respectively. i , ΔI j , we get the magnetic field energy W(I i +ΔIi ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j ) and W(I i -ΔI i ,I j -ΔI j ), and then solve the self-inductance according to formula (17)

[0142] The LPST air gap magnetic field model calculation program was written based on MATLAB software, and the LPST simulation model was built based on finite element software. The accuracy of the LPST air gap magnetic field model in the present invention can be verified by the operating characteristics of the transformer under different working conditions of the finite element model. Figure 5 、 Figure 6 .

[0143] Compared with the finite element model, the LPST air gap magnetic field distribution error calculated by the present invention is less than 2%. The self-inductance parameter error of the 12 phases on the primary side is 0.812%, and the mutual inductance error (taking phase a1 as an example) is 2.43%. The self-inductance parameter error of the three phases on the secondary side is 4.002%, and the mutual inductance error is 0.352%. The mutual inductance error between the primary and secondary sides (taking phase a1 as an example) is 1.499%.

[0144] Figure 5 The figure shows the comparison between the air gap magnetic field model proposed in the present invention and the finite element simulation results when the DC bus voltage of LPST is 100V.

[0145] Figure 6 Comparison between the calculated values ​​and finite element simulation values ​​of each LPST winding when the DC bus voltage is 100V.

[0146] In summary, if Figure 7 As shown, a LPST air gap magnetic field model can be derived that simultaneously accounts for the effects of ferromagnetic material saturation, cogging, and core disconnection. This reduces the computational dimension and shortens the calculation time while maintaining accuracy. This magnetic field modeling method meets the preliminary design requirements of the LPST and is versatile. The inductance parameters obtained through precise magnetic field analysis are more accurate.

[0147] The above description is merely a preferred embodiment of the present invention and does not constitute any form of limitation to the present invention. Any person skilled in the art can utilize the methods and technical contents disclosed above to make many possible variations and modifications to the technical solutions of the present invention without departing from the scope of the technical solutions of the present invention, or modify them into equivalent embodiments with equivalent variations. Therefore, any equivalent variations made in accordance with the shape, structure, and principles of the present invention without departing from the content of the technical solutions of the present invention should be included in the scope of protection of the present invention.

Claims

1. A LPST air gap magnetic field modeling method, characterized in that: The following steps are involved: Step S1: Analyze the relationship between the primary and secondary sides of the LPST and the air gap, and build an analytical model of the air gap magnetic field considering the saturation of ferromagnetic materials; Step S101: Divide the LPST into N blocks at equal intervals along the length of the air gap, primary tooth, primary yoke, secondary tooth, and secondary yoke. Within the LPST length, divide the air gap centerline into N blocks at equal intervals, with a total of N+1 nodes. Obtain the total magnetic voltage drop of the circuit, the normal magnetic voltage drop of the air gap at the mth and m+1th nodes, the longitudinal magnetic voltage drop of the yoke at the mth node of the primary core, the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the primary core, the longitudinal magnetic voltage drop of the yoke at the mth node of the secondary core, and the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the secondary core; Step S102: fitting the BH curve of the nonlinear ferromagnetic material to correct the actual normal flux density of the tooth when the ferromagnetic material is saturated; Step S103: According to the flux continuity theorem, the yoke transverse magnetic flux at the mth node is equal to the total longitudinal magnetic flux of the air gap from the first node to the mth node, and the yoke magnetic flux is obtained; Step S104: Simplify the magnetomotive force of each node in the air gap based on the distribution characteristics of the LPST winding, and obtain the air gap flux density at the mth node; Step S105: Based on the simplified magnetomotive force of each node in the air gap, iterative judgment is performed until the judgment condition is met. If the condition is met, the air gap magnetic flux density is output; if the condition is not met, the air gap magnetic flux density is corrected; Step S2: Analyze the relationship between the LPST cogging and the air gap, and build an air gap magnetic field correction model that takes the cogging effect into account; Step S201: determining the relative magnetic permeability of the air gap considering the influence of the cogging effect; Step S3: Analyze the relationship between the LPST edge and the air gap, and build an air gap magnetic field correction model considering the core breaking; Step S301: simplify the edge model and regularize the magnetic field distribution through Hook transformation; Step S302: taking into account the end magnetic flux density of the core being broken; Step S303: using the flux continuity theorem, introducing the core breaking flux density change value to represent the lost flux density; Step S4: The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of the slot and core disconnection factors.

2. The LPST air gap magnetic field modeling method according to claim 1, wherein: The primary and secondary sides of the LPST are symmetrical about the center line of the air gap.

3. The LPST air gap magnetic field modeling method according to claim 2, wherein: The total magnetic voltage drop in the circuit is: F ∑ (m)=F δ_n (m+1)+F t1_n (m+1)-F δ_n (m)-F j1_l (m)-F t1_n (m) Where, F δ_n (m), F δ_n (m+1) is the normal magnetic pressure drop of the air gap at the mth and m+1th nodes, F j1_l (m) is the longitudinal magnetic voltage drop of the yoke at the mth node of the primary core, F t1_n (m), F t1_n (m+1) is the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the primary core. The positive direction of the coordinate axis is the positive direction of the magnetic voltage drop of each segment. According to Ampere's loop theorem: Where H δ_n is the magnetic field strength along the normal direction of the air gap; B t1_n 、H t1_n is the normal magnetic flux density and magnetic flux density of the virtual tooth of the primary core; B j1_l 、H j1_l is the longitudinal magnetic flux density and magnetic flux density of the primary core yoke; δ is the air gap length; h t1 is the virtual tooth height of the primary core; μ0 is the magnetic permeability of air; μ j1 is the magnetic permeability of each node of the primary core yoke.

4. The LPST air gap magnetic field modeling method according to claim 3, wherein: The actual normal magnetic flux density of the tooth is: B t_n (m)=B t_n ′(m)-μ0H t_n (m)k δ Where B′ t_n (m) is the apparent normal flux density of the tooth at the mth node, which means the flux density when all the magnetic flux enters the tooth; B t_n (m) is the actual normal flux density of the tooth at the mth node; H t_n (m) is the actual normal magnetic field intensity of the tooth at the mth node; k δ is the slot coefficient, k δ =(H·b s ) / (K Fe ·L·b t ).

5. The LPST air gap magnetic field modeling method according to claim 4, characterized in that: The magnetic flux density of the yoke is:

6. The LPST air gap magnetic field modeling method according to claim 5, characterized in that: The simplified magnetomotive force of each node in the air gap is: Where, The air gap magnetic flux density at the mth node is: Where K B It is a saturation coefficient preset according to the saturation degree of the primary side core.

7. The LPST air gap magnetic field modeling method according to claim 6, characterized in that: The judgment conditions are: If the judgment conditions are not met, the air gap magnetic flux density is corrected: Where k s is the iteration coefficient.

8. The LPST air gap magnetic field modeling method according to claim 1, wherein: The relative magnetic permeability of the air gap is: Where t1 is the tooth pitch and T is the extension period, then: The nonlinear function β(y) is: Wherein, the expression of v is obtained by software fitting.

9. The LPST air gap magnetic field modeling method according to claim 1, wherein: The end magnetic flux density taking into account the core breaking is: Using the flux continuity theorem, the core breaking flux change value Q is introduced to represent the lost flux density: The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of the slot and core disconnection factors: B _x (x)=(1-Q)B δ_sl_n (x)·λ YB_c_n ·l FB_c_n Where B δ0_sl_n (x) is the expression of the air gap flux density considering the saturation of ferromagnetic materials; λ YB_c_n ,λ FB_c_n These are the relative permeability functions of the air gap when slots are opened on the primary and secondary sides, respectively.

10. A method for obtaining inductance parameters, wherein the inductance parameters are obtained by using a model constructed by the LPST air gap magnetic field modeling method according to any one of claims 1 to 9, wherein: include: In the LPST effective region, the energy stored in the air gap magnetic field is: Solve the self-inductance and mutual inductance parameters of each LPST winding, including: When solving the self-inductance of the i-th winding, a small current perturbation ΔI is made to the i-th winding. i , the other winding currents remain unchanged, and the magnetic field energy W(I i +ΔI i ) and W(I i -ΔI i ), solve for self-inductance When solving the mutual inductance of the i-th winding and the j-th winding, a small current perturbation ΔI is made to the i-th and j-th windings respectively. i , ΔI j , we get the magnetic field energy W(I i +ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j ) and W(I i -ΔI i ,I j -ΔI j ), solve for self-inductance

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