WPT system ZVS control method and system based on DC bias variable capacitor

Abstract

The invention relates to the technical field of wireless power transmission (WPT), and particularly discloses a ZVS control method and system for a WPT system based on a direct current bias variable capacitor, which utilizes asymmetric voltage control (AVC) to realize bias of direct current voltage, directly generates square wave signals with asymmetric positive and negative half cycles at an alternating current side of a primary side inversion full bridge, and realizes zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero zero-zero and through Fourier decomposition, a direct current voltage acting on the direct current bias variable capacitor can be obtained. And additional auxiliary devices do not need to be controlled, so that the control is relatively simple. The direct-current voltage generated by the AVC is used for changing the capacitance value of the ceramic capacitor, and hard switching loss caused by system parameter changes can be easily reduced. On the basis of the bilateral LCC system, phase difference control is carried out under the condition that coil parameters change, and the purpose of flexibly controlling the phase difference between the voltage and the current is achieved.

Description

Technical Field

[0001] The present invention relates to the technical field of wireless power transmission (WPT), and in particular to a ZVS control method and system for a WPT system based on a DC bias variable capacitor. Background Art

[0002] Wireless power transfer (WPT) technology can transfer electrical energy from a power source to a load without electrical contact. Over the years, WPT technology has attracted widespread attention and has been widely used in consumer electronics, rotating equipment, electric vehicles, biomedical implants, and underwater detection equipment. However, due to system parameter deviations and temperature changes, the actual resonant frequency often changes, causing the system's resonant frequency to deviate from the operating frequency. This will cause the current to advance or lag relative to the voltage, making the system unable to operate in a soft switching state, increasing system losses, reducing the actual efficiency of the system, and causing certain electromagnetic interference (EMI), affecting the reliability of the system.

[0003] In order to avoid hard switching, make the phase of the primary voltage and current of the system reach the ideal range, and achieve zero voltage switching (ZVS), there are many methods to achieve this goal. Variable parameter components are one of the important methods. Using controllable passive devices such as variable capacitors (VC), the equivalent impedance of the system can be flexibly adjusted. In the research on variable parameter components, variable capacitors have the characteristics of low parasitic resistance. At present, the research on variable capacitors is mainly divided into two types. One is the switching capacitor: it is generally composed of a series or parallel connection of a switch tube and a capacitor. By controlling the conduction angle of the switch, the equivalent capacitance value is changed to achieve the effect of the variable parameter. However, this method relies on the precise detection of the current. When the detected current is distorted, the control will not meet expectations; the second is the DC bias variable capacitor, which uses the second type of ceramic capacitor. Under the application of DC bias voltage, due to material properties and manufacturing process, its capacitance value will change to a certain extent, such as Figure 1 The curve of the capacitance value of capacitor GRM31CR72A105KA01 changing with the DC bias voltage when the DC bias voltage is applied. However, the current research on DC bias variable capacitors often requires additional auxiliary devices to generate the DC voltage when applying the DC voltage. Similar to the switched capacitor, the additional auxiliary devices often require additional control, which increases the complexity of the system. Summary of the invention

[0004] The present invention provides a WPT system ZVS control method and system based on a DC bias variable capacitor, and solves the technical problem that the existing scheme for realizing WPT system ZVS based on a DC bias variable capacitor requires additional auxiliary devices to realize the generation of DC voltage, requires additional control, and increases the complexity of the system.

[0005] In order to solve the above technical problems, the present invention provides a WPT system ZVS control method based on DC bias variable capacitor, comprising the steps of:

[0006] S1. Determine the circuit structure and design parameters of the WPT system, wherein the circuit structure adopts an LCC-LCC compensation network, the series resonant capacitor at the transmitting end adopts a DC bias variable capacitor, and the full-bridge inverter at the transmitting end is driven by an asymmetric square wave signal;

[0007] S2. Set the parameter α of the asymmetric square wave signal at power-on + =α=0°, β=180°, α refers to α + The value of α + It is defined as the angle at which the voltage waveform changes from positive to 0 in one cycle, and β is defined as the angle at which the voltage waveform changes from 0 positive to 0 negative in one cycle;

[0008] S3: After power-on, sample the voltage and current of the primary AC measurement and calculate the current phase difference between the voltage and current of the primary AC measurement.

[0009] S4. Determine the current phase difference Phase difference with reference If the current phase difference Greater than the reference phase difference Keep β unchanged, according to the current phase difference Phase difference with reference The difference between them calculates the new α; if the phase difference Smaller than the reference phase difference Keep α unchanged, according to the current phase difference Phase difference with reference The new β is calculated by the difference between Equal to the reference phase difference Then return to step S3;

[0010] S5. Change α or β to the newly calculated α or β and return to step S3.

[0011] Furthermore, in step S4, the step of calculating a new α is:

[0012] The current phase difference Phase difference with reference Make a difference and get the phase deviation e1(k) at the current moment;

[0013] According to the obtained phase deviation e1(k), the first PID module is used to calculate α=K P1 *e1(k)+K I1*Σe1(i)+K D1 *[e1(k)-e1(k-1)] / T calculates the new α, where k represents the current sampling moment, T is the sampling period, Σe1(i) represents the cumulative sum of phase deviations from the initial moment to the current moment when calculating the new α, and K P1 , K I1 , K D1 They represent the proportional coefficient, integral coefficient and differential coefficient when calculating α respectively.

[0014] Further, in step S4, the steps of calculating the new β are:

[0015] The current phase difference Phase difference with reference Make a difference and get the phase deviation e2(k) at the current moment;

[0016] According to the obtained phase deviation e2(k), the second PID module is used according to β=K P2 *e2(k)+K I2 *Σe2(i)+K D2 *[e2(k)-e2(k-1)] / T calculates the new β, K P2 , K I2 , K D2 They respectively represent the proportional coefficient, integral coefficient and differential coefficient (PID parameters) when calculating the new β, and Σe2(i) represents the accumulated phase deviation from the initial moment to the current moment when calculating the new β.

[0017] Furthermore, in step S5, the phase difference φ of the first-order harmonic component of the inverter output voltage relative to the asymmetric square wave signal is calculated according to the newly calculated α or β. v1 In step S3, the phase difference φ is obtained v1 After that, the current phase difference is the phase difference between the voltage and current measured by the primary side AC calculated by sampling minus the phase difference φ v1 .

[0018] Furthermore, the phase difference φ v1 equal a1 and b1 are defined as:

[0019] a1=sin(β-α + )+sin(β)+sin(α - ), b1=1-cos(β-α + )-cos(β)+cos(α - ), where α - definition

[0020] It is the angle between when the voltage changes from negative voltage to 0 voltage in one cycle and when the next cycle begins.

[0021] Furthermore, α - Always equal to 0°.

[0022] Furthermore, the reference phase difference Determined by the output capacitance of the MOSFET of the full-bridge inverter.

[0023] The present invention also provides a WPT system ZVS control system based on a DC bias variable capacitor, the key of which is that it includes a primary side sampling module and a primary side control module, the primary side sampling module is used to sample the voltage and current of the primary side AC measurement and send it to the primary side control module; the primary side control module is used to execute the remaining contents of the WPT system ZVS control method based on a DC bias variable capacitor except sampling the voltage and current of the primary side AC measurement.

[0024] The present invention provides a WPT system ZVS control method and system based on DC bias variable capacitor, which realizes DC voltage bias by using asymmetric voltage control (AVC), directly generates asymmetric square wave signals of positive and negative half cycles on the AC side of the primary inverter full bridge, and obtains the DC voltage acting on the DC bias variable capacitor in the primary resonant circuit through Fourier decomposition. Since no additional auxiliary devices need to be controlled, the control is relatively simple. By using the DC voltage generated by AVC to change the capacitance of the ceramic capacitor, the hard switching loss caused by the change of system parameters can be easily reduced. The present invention is based on a double-sided LCC system, and performs phase difference control when the coil parameters change, thereby achieving the purpose of flexibly controlling the phase difference between voltage and current. BRIEF DESCRIPTION OF THE DRAWINGS

[0025] Figure 1 is a curve diagram showing the effect of a DC bias voltage on a capacitor provided by an embodiment of the present invention;

[0026] Figure 2 is a waveform diagram of an asymmetric voltage-controlled output voltage provided by an embodiment of the present invention;

[0027] Figure 3 is a typical waveform diagram of asymmetric voltage control output provided by an embodiment of the present invention;

[0028] Figure 4 is a circuit diagram of a WPT system with bilateral LCC resonance compensation provided by an embodiment of the present invention;

[0029] Figure 5 The embodiment of the present invention provides Figure 4 The mutual inductance equivalent circuit diagram;

[0030] Figure 6 It is a separate control α provided by an embodiment of the present invention +=α(0° to 180°), β=180° when the fundamental voltage (V ab1 ) relative to the maximum fundamental amplitude (V ab1max ) percentage and phase influence diagram;

[0031] Figure 7 It is a separate control α provided by an embodiment of the present invention + =α=0°, β is a variable (0° to 360°) when the fundamental voltage (V ab1 ) relative to the maximum fundamental amplitude (V ab1max ) percentage and phase influence diagram;

[0032] Figure 8 It is a separate control α provided by an embodiment of the present invention + =α(0° to 180°), β=180° for fundamental voltage (V ab1 ) influence diagram;

[0033] Fig. 9 It is a separate control α provided by an embodiment of the present invention + =α=0°, β is variable (0° to 360°) for the fundamental voltage (V ab1 ) influence diagram;

[0034] Fig.10 It is α provided by the embodiment of the present invention + The effect of AVC on the voltage fundamental phase when and β change simultaneously;

[0035] Fig.11 The primary side series capacitor C provided in the embodiment of the present invention p The change diagram of the input impedance angle of the primary side when it changes between 30% and 130% and the other parameters remain unchanged;

[0036] Fig.12 is a flow chart of a ZVS control method provided by an embodiment of the present invention;

[0037] Fig.13 The data provided in the embodiment of the present invention is fitted to show the change of capacitance with the DC bias voltage;

[0038] Fig.14 is a waveform diagram of square wave voltage and current at the primary inverter port when the system provided by the embodiment of the present invention is in full resonance;

[0039] Fig.15 It is a waveform diagram of square wave voltage and current at the primary inverter port when the system provided by the embodiment of the present invention is in a capacitive detuning state and is not controlled;

[0040] Fig.16It is a waveform diagram of square wave voltage and current at the primary inverter port when the system provided by the embodiment of the present invention is in capacitive detuning control. DETAILED DESCRIPTION

[0041] The following specifically illustrates the implementation mode of the present invention in conjunction with the accompanying drawings. The embodiments are provided for illustrative purposes only and cannot be understood as limiting the present invention. The accompanying drawings are provided for reference and illustration only and do not constitute a limitation on the scope of patent protection of the present invention, because many changes can be made to the present invention without departing from the spirit and scope of the present invention.

[0042] For the full-bridge resonant converter of the WPT system, Figure 2 A generalized quasi-square wave output voltage is defined. The output voltage is controlled by up to four variables: three angle variables (β, α + and α - , where β is defined as the angle from the beginning of a cycle until the voltage changes from 0 voltage to negative voltage, α + and α - They are defined as the angle between when the voltage changes from positive voltage to 0 voltage and when it changes from 0 voltage to negative voltage in one cycle, and the angle between when the voltage changes from negative voltage to 0 voltage and when the next cycle begins. A period variable T s , in general fixed frequency control, the operating frequency is constant, so T s Constant, only three variables need to be considered for control.

[0043] Figure 2 In the case of α + =α - When β is a variable and β = 180°, it becomes a commonly used phase shift control. + =α - =0°, and β is a variable, it becomes asymmetric duty cycle control (ADC). Figure 3 The figure is a typical output waveform of asymmetric voltage control. Using the fundamental wave analysis method, the phase relationship between the fundamental wave of the output voltage and the current can be obtained. Figure 3 Medium ab With v ab1 are square wave voltage and fundamental wave voltage respectively, i o It is the current obtained from the fundamental voltage. Using Fourier analysis, we can get the expression formula of each component of the output voltage and the phase of each component relative to the square wave:

[0044]

[0045] Where V abh is the h-order harmonic component of the inverter output voltage, φ vh Yes V abhRelative to the phase difference of the asymmetric square wave signal, V ab0 is the DC component, V i is the input DC voltage. h and b h They are:

[0046] a h = sin(β-α + )+sin(hβ)+sin(hα - ) (4)

[0047] b h =1-cos(h(β-α + ))-cos(hβ)+cos(hα - ) (5)

[0048] Figure 3 Where φ1 is the phase difference of the fundamental current relative to the fundamental voltage. The phase difference of the current relative to the voltage is only related to the circuit topology and the actual parameters of the circuit, and has nothing to do with other things. Δφ is defined as the phase difference of the current relative to the square wave voltage.

[0049] Depend on Figure 1 It can be seen that under the influence of DC bias voltage, the capacitance of the capacitor will decrease as the voltage increases, which will lead to an increase in capacitive reactance. For the SS topology, its sensitivity to parameters requires more precise capacitance control, which increases the difficulty of control. Therefore, a double-sided LCC is selected in the topology structure. For LCC, the reduction in capacitance in the coil resonant cavity will make the overall circuit present an inductive characteristic, which will be more conducive to achieving ZVS. Therefore, in this example, a double LCC is selected as the analysis object.

[0050] Figure 4 This is the circuit structure diagram of the dual LCC topology wireless power transmission system. The primary side of the system adopts a full-bridge inverter circuit, which is composed of four MOS tubes from S1 to S4, to convert DC power into high-frequency AC power and transmit it through a resonant network. The secondary side adopts a diode uncontrolled rectifier, which is composed of four diodes from D1 to D4, to convert the output AC power into DC power to supply energy to the load. in Indicates the DC input power of the system, U R is the DC voltage output by the secondary side, U AB and U ab are the primary and secondary AC voltages, and the primary and secondary coil self-inductances are L p and L s , the compensation inductances are L f1 and L f2 , the series compensation capacitor is C pv and C s , where C pvis a variable capacitor, and the parallel compensation capacitors are C f1 and C f2 , M is the mutual inductance between the primary and secondary coils, R L For load.

[0051] Figure 5 This is the equivalent circuit diagram of the mutual inductance of the bilateral LCC topology.

[0052] The mutual inductance M between the coils can be expressed by the coupling coefficient k:

[0053]

[0054] Define the primary impedance as Z p , reflected impedance Z r , the secondary impedance is Z s , ignoring the compensation inductance, capacitance and coil internal resistance, the following formula can be obtained:

[0055]

[0056] Among them, R eq is the equivalent resistance of the load to AC measurement, and ω represents the operating angular frequency of the system.

[0057] Where R eq The equivalent resistance of the load to AC measurement.

[0058]

[0059] The resonance condition of the double-sided LCC is:

[0060]

[0061] When the system is in a resonant state, ignoring the parasitic resistance of the coil resistance and the inductor and capacitor, the current parameters of the primary side of the system can be obtained:

[0062]

[0063] When the system is in a resonant state, the system current is the largest. Formula (12) can be used to obtain the system current under resonant conditions. The variable capacitor is selected accordingly and the voltage and current are balanced according to the maximum current to ensure system reliability.

[0064] Due to the dominant position of the fundamental wave component of the system, the phase difference between the system voltage fundamental wave and the current fundamental wave can be expressed as follows using the fundamental wave analysis method combined with formulas (7) and (8):

[0065]

[0066] When the system is in resonance, φ1 = 0°, and the phase relationship between the current and the square wave is the same as the phase relationship between the voltage and the square wave. Figure 3 You can get:

[0067] Δφ=φ1-φ v1 (14)

[0068] In order to meet the ZVS condition, the current needs to lag the square wave voltage by a certain angle, so Δφ needs to be greater than 0. For AVC, control will change the phase of the voltage fundamental wave relative to the square wave to a certain extent. From formulas (2), (4), and (5), it can be seen that the effects of α+ and α- on control are basically similar. The difference is that the positive and negative DC bias voltages obtained by Fourier decomposition have the same effect on the phase. In order to simplify the control, only α is considered later. + and β have an influence on the control, during which α is always 0°.

[0069] Through the formula mentioned above, we can get different α + The influence of α and β on the phase of the system can be seen. + When the control is performed, the phase angle brought by the control will make the phase of the fundamental voltage of the system ahead of the square wave voltage, and at the same time + When the control of β and β is used, the voltage fundamental wave will lead or lag the system according to different values. Therefore, in the case of large inductance, due to the large turn-off current, the inductance of the circuit needs to be reduced to reduce the current when the switch is turned off. At this time, the voltage fundamental wave phase of the system can be advanced to the square wave through AVC, so that the phase difference between the current and the square wave voltage is reduced to reduce the turn-off loss; when the system is capacitive, it can be controlled by C pv Reduce Z p To achieve ZVS, this method is more suitable for achieving ZVS and has a better control effect on ZVS. If the circuit has very large inductance, although this method has a certain effect in reducing the turn-off current at the switching moment, it cannot fully achieve the expected control goal.

[0070] According to formula (2), the influence of AVC on the system voltage fundamental wave can be obtained.

[0071] When no control is performed, the amplitude of the voltage fundamental wave is the largest, and the voltage fundamental wave has no phase difference with respect to the square wave. The maximum amplitude of the fundamental wave is:

[0072]

[0073] When controlling α alone + When β = 180°, the influence of AVC on the voltage fundamental wave can be obtained from formulas (2), (4), and (5):

[0074]

[0075] When controlling β, α + =0°, the influence of AVC on the fundamental voltage is:

[0076]

[0077] Individual control of α + = α (0° to 180°), β = 180° and single control α + =α=0°, β is a variable (0° to 360°), the fundamental voltage (V ab1 ) relative to the maximum fundamental amplitude (V ab1max ) and the influence of phase are as follows: Figure 6 , Figure 7 As shown. Figure 6 , Figure 7 It can be seen that when performing AVC control, α is controlled separately + The impact on the fundamental amplitude will be smaller; when β is controlled, its effect on the voltage fundamental amplitude decreases significantly as the control increases. Comparing the two different control methods, the latter control obviously needs to be limited to a certain range, otherwise the reduction of the fundamental amplitude will significantly change the size of the transmission power; while for the former, at the maximum control angle, the voltage fundamental amplitude is half of the maximum fundamental amplitude, and its impact on power is minimal.

[0078] Individual control of α + = α (0° to 180°), β = 180° and single control α + =α=0°, β is a variable (0° to 360°), the impact on the voltage fundamental phase is as follows: Figure 8 and Fig. 9 As shown. Figure 8 You can see α + When the phase changes from 0 to 180° and β=180°, the influence of AVC on the voltage fundamental phase is leading. Fig. 9 Represents α + =0°, when β changes from 0 to 360°, the influence of AVC on the voltage fundamental phase changes from leading to lagging as β increases. Given that it generally starts from β=180°, it can be considered that the fundamental lags as the β angle increases.

[0079] Fig.10 is α + and β change simultaneously (α + From 0° to 180°, β From 0° to 360°, α +The influence of AVC on the voltage fundamental phase when the voltage does not exceed β. Fig.10 It can be seen that the influence of control on phase is relatively complicated at this time. In order to make the control simpler, only the first two separate control methods are considered for AVC control.

[0080] For the two separate control methods, it can be obtained from formula (9) that when the capacitance of the capacitor connected in series with the coil in the LCC decreases, for the impedance characteristics of the inverter port, the decrease in capacitance will cause the circuit current to lag behind the voltage.

[0081] Fig.11 The primary series capacitor C p When the input impedance angle of the primary side changes between 30% and 130% and the other parameters remain unchanged, it can be seen that the input impedance angle of the primary side changes between 30% and 130% and ... p When it decreases, the impedance angle is positive, the system is inductive and as the capacitance decreases, the inductance gradually increases; otherwise, the system is capacitive.

[0082] Combination Figures 8 to 10 The influence of AVC on the fundamental wave of system voltage can be concluded as follows:

[0083] 1) For systems with high inductance, it is desirable to reduce the phase difference between the current and the square wave, but in controlling α + When the capacitance is reduced, the AVC's influence on the voltage and the capacitance's influence on the current are opposite in direction, which may cause the current phase difference relative to the square wave to decrease in an insignificant amount, or even increase, which is related to the circuit parameters.

[0084] 2) For capacitive systems, controlling β will cause the voltage signal to lag behind the square wave. At the same time, the reduction in capacitance will cause the current signal to lag behind the voltage fundamental signal. Since the impact of both is consistent with the expected target, only a relatively small control angle is often required to achieve the control effect. At this time, the impact of control on the fundamental voltage amplitude is also at a relatively small stage, and the overall effect will be relatively good.

[0085] According to the above conclusions, the following ZVS control method can be implemented, as shown in 12, which includes the steps:

[0086] S1. Determine the circuit structure and design parameters of the WPT system, wherein the circuit structure adopts an LCC-LCC compensation network, the series resonant capacitor at the transmitting end adopts a DC bias variable capacitor, and the full-bridge inverter at the transmitting end is driven by an asymmetric square wave signal;

[0087] S2, the system defaults to α when powered on + =α=0°, β is 180°, α- is always equal to 0°;

[0088] S3: After power-on, sample the voltage and current of the primary AC measurement and calculate the current phase difference between the voltage and current of the primary AC measurement.

[0089] S4. Determine the current phase difference Phase difference with reference If the current phase difference Greater than the reference phase difference Keep β unchanged, according to the current phase difference Phase difference with reference The difference between them calculates the new α; if the phase difference Smaller than the reference phase difference Keep α unchanged, according to the current phase difference Phase difference with reference The new β is calculated by the difference between Equal to the reference phase difference Then return to step S3;

[0090] S5. Change α or β to the newly calculated α or β and return to step S3.

[0091] In step S3, the reference phase difference According to the output capacitance of the inverter MOSFET (C oss ) OK, C oss is the gate-drain capacitance (C gd ) and drain-source capacitance (C ds ) (When MOSFETGS is short-circuited, C gd and C ds In parallel), according to different C oss The inductance required to achieve ZVS is also different, and the required reference phase difference needs to be calculated based on the corresponding switching tube.

[0092] In step S4, the steps for calculating the new α are:

[0093] S41, the current phase difference Phase difference with reference Make a difference and get the phase deviation e1(k) at the current moment;

[0094] S42, according to the obtained phase deviation e1(k), use the first PID module to calculate α=K P1 *e1(k)+K I1 *

[0095] Σe1(i)+K D1*[e1(k)-e1(k-1)] / T calculates the new α, where k represents the current sampling moment, T is the sampling period, and Σe1(i) represents the cumulative sum of the phase deviations from the initial moment to the current moment when calculating the new α.

[0096] K P1 , K I1 , K D1 They respectively represent the proportional coefficient, integral coefficient and differential coefficient (PID parameters) when calculating α.

[0097] In step S4, the steps for calculating the new β are:

[0098] S43, the current phase difference Phase difference with reference Make a difference and get the phase deviation e2(k) at the current moment;

[0099] S44, the same as calculating α, according to the obtained phase deviation e2(k), the second PID module is used according to β=K P2 *e2(k)+K I2 *Σe2(i)+K D2 *[e2(k)-e2(k-1)] / T calculates the new β, K P2 , K I2 , K D2 They respectively represent the proportional coefficient, integral coefficient and differential coefficient (PID parameters) when calculating the new β, and Σe2(i) represents the accumulated phase deviation from the initial moment to the current moment when calculating the new β.

[0100] However, it should be noted that when calculating α and β, the PID parameters of PID control are different and need to be set separately according to actual conditions.

[0101] S5. Calculate the phase error based on the newly calculated α or β Then return to step S2.

[0102] In step S5, the phase error The calculation steps are:

[0103] According to the control angle (α or β) output at this time, the phase difference φ of the first-order harmonic component caused by the control relative to the square wave is calculated by formulas (2), (4) and (5): v1 ;

[0104] In step S5, the phase error φ is calculated v1 After that, when the next control is performed, the current phase difference obtained by PQ detection needs to be subtracted from the phase error φ v1 , get the correct current phase difference.

[0105] Based on the above-mentioned ZVS control method, this embodiment also provides a ZVS control system, including a primary side sampling module and a primary side control module, wherein the primary side sampling module is used to sample the voltage and current of the primary side AC measurement and send them to the primary side control module; the primary side control module is used to execute the remaining contents of the above-mentioned control method except sampling the voltage and current of the primary side AC measurement.

[0106] The embodiment of the present invention provides a WPT system ZVS control method and system based on DC bias variable capacitor, which uses asymmetric voltage control (AVC) to realize DC voltage bias, directly generates asymmetric square wave signals of positive and negative half cycles on the AC side of the primary inverter full bridge, and the specific value of the DC component can be calculated by Fourier analysis formula according to the control angle (α and β). Since there is no need to control additional auxiliary devices, the control is relatively simple. By using the DC voltage generated by AVC to change the capacitance of the ceramic capacitor, the hard switching loss caused by the change of system parameters can be easily reduced. Based on the double-sided LCC system, the present invention performs phase difference control when the coil parameters change, thereby achieving the purpose of flexibly controlling the phase difference between voltage and current.

[0107] The following is a simulation analysis.

[0108] 1. Establishment of variable capacitor simulation model

[0109] Since the DC bias characteristics of a capacitor are determined by the physical and chemical properties of the capacitor's internal materials, which causes charge accumulation or polarization at both ends of the capacitor under DC voltage bias conditions, additional modeling of the capacitor is required before simulation.

[0110] According to the previous analysis, we first need to model the DC bias variable capacitor. Figure 1 The percentage of capacitance change under each DC bias voltage is fitted from the data to obtain a curve showing the change of capacitance with DC bias voltage. Fig.13 As shown. After obtaining the curve equation of the capacitance changing with the DC bias voltage, in the simulation model, write a function of the capacitance changing with voltage. Through the VC module, the capacitance of the capacitor can be changed in real time during the simulation process. The DC component of the inverter port voltage is detected through the Fourier module (in the simulation model, this module is not needed in practice, only the formula needs to be calculated on the DSP), which is used as the dependent variable, and the real-time capacitance value is calculated through the capacitance change function.

[0111] 2. Simulation parameter setting

[0112] Since the withstand voltage and current of a single multilayer ceramic capacitor (MLCC) are limited, this requirement must be taken into account when designing system parameters. This article selects the "GRM31CR72A105KA01" capacitor for simulation. The withstand voltage and current values ​​of "GRM31CR72A105KA01" are obtained from the data sheet. According to the withstand voltage and current requirements, the maximum current at resonance is calculated in combination with formula (12). According to the maximum current and voltage of each branch on the primary side of the system, the data sheet is consulted to select the appropriate capacitor, and the corresponding capacitor array is designed with appropriate voltage and current balancing. The design is shown in Table 1 for simulation parameters, where f s refers to the self-resonant frequency of the system, and f0 refers to the operating frequency of the system.

[0113] Table 1

[0114]

[0115] 3. Simulation results

[0116] According to the simulation circuit model built with the above parameters, the waveforms of the square wave voltage and current at the primary inverter port are as follows: Fig.14 , Fig.15 and Fig.16 As shown. Fig.14 It can be seen from the figure that when the system coil parameters are completely resonant, there is a certain lag angle between the current and the square wave voltage, and the system is in the ZVS state. After the system changes the coil parameters according to the parameters in Table 1, the simulation results are as follows Fig.15 As shown, it can be seen that the phase difference between current and voltage is negative, the current leads the voltage, the system exhibits significant capacitance, and does not meet the ZVS condition. Fig.16 The control effect of AVC combined with DC bias variable capacitor under the parameters in Table 1 is shown. It can be seen that after the control, the phase difference of the current lagging behind the voltage is the same as that in the complete resonance, but compared with the complete resonance state, during the process of the voltage positive cycle converting to the negative cycle, the current meets the positive and negative requirements of ZVS, but the current value is slightly smaller, and ZVS may not be fully achieved. Although the small current at the moment of voltage zero crossing will slow down the discharge process of the switch tube and may cause some oscillation, it is relatively Fig.15 In the case where the current significantly leads the voltage, the switching losses and oscillations will still be smaller.

[0117] However, when implementing AVC control, Figure 7It can be known that the amplitude of the fundamental voltage will decrease accordingly, which will slightly reduce the power output of the system. Therefore, when performing control, it is necessary to limit the maximum control amount in order to reduce the change of the control on the power output.

[0118] By adjusting the variable capacitor value through AVC and combining the influence of AVC on the system, the ZVS of the switch tube is realized under the configuration of capacitive system parameters through simulation, and the influence of AVC on the fundamental voltage amplitude is analyzed. The final results show that this method can adjust the capacitive system to achieve ZVS to a certain extent, but it will slightly reduce the system power output.

[0119] The above embodiments are preferred implementation modes of the present invention, but the implementation modes of the present invention are not limited to the above embodiments. Any other changes, modifications, substitutions, combinations, and simplifications that do not deviate from the spirit and principles of the present invention should be equivalent replacement methods and are included in the protection scope of the present invention.

Claims

1. A ZVS control method for a WPT system based on a DC bias variable capacitor, characterized in that: Includes steps: S1. Determine the circuit structure and design parameters of the WPT system, wherein the circuit structure adopts an LCC-LCC compensation network, the series resonant capacitor at the transmitting end adopts a DC bias variable capacitor, and the full-bridge inverter at the transmitting end is driven by an asymmetric square wave signal; S2. Set the parameter α of the asymmetric square wave signal at power-on + =α=0°, β=180°, α refers to α + The value of α + It is defined as the angle at which the voltage waveform changes from positive to 0 in one cycle, and β is defined as the angle at which the voltage waveform changes from 0 positive to 0 negative in one cycle; S3: After power-on, sample the voltage and current of the primary AC measurement and calculate the current phase difference between the voltage and current of the primary AC measurement. S4. Determine the current phase difference Phase difference with reference If the current phase difference Greater than the reference phase difference Keep β unchanged, according to the current phase difference Phase difference with reference The difference between them calculates the new α; if the phase difference Smaller than the reference phase difference Keep α unchanged, according to the current phase difference Phase difference with reference The difference between them is used to calculate the new β; if the phase difference Equal to the reference phase difference Then return to step S3; S5. Change α or β to the newly calculated α or β and return to step S3.

2. The ZVS control method of a WPT system based on a DC bias variable capacitor according to claim 1, characterized in that: In step S4, the steps of calculating a new α are: The current phase difference Phase difference with reference Make a difference and get the phase deviation e1(k) at the current moment; According to the obtained phase deviation e1(k), the first PID module is used to calculate α=K P1 *e1(k)+K I1 *Σe1(i)+K D1 *[e1(k)-e1(k-1)] / T calculates the new α, where k represents the current sampling moment, T is the sampling period, Σe1(i) represents the cumulative sum of phase deviations from the initial moment to the current moment when calculating the new α, and K P1 , K I1 , K D1 They represent the proportional coefficient, integral coefficient and differential coefficient when calculating α respectively.

3. The ZVS control method and system of the WPT system based on DC bias variable capacitor according to claim 2 is characterized in that: In step S4, the steps for calculating the new β are: The current phase difference Phase difference with reference Make a difference and get the phase deviation e2(k) at the current moment; According to the obtained phase deviation e2(k), the second PID module is used according to β=K P2 *e2(k)+K I2 *Σe2(i)+K D2 *[e2(k)-e2(k-1)] / T calculates the new β, K P2 , K I2 , K D2 They respectively represent the proportional coefficient, integral coefficient and differential coefficient (PID parameters) when calculating the new β, and Σe2(i) represents the accumulated phase deviation from the initial moment to the current moment when calculating the new β.

4. The ZVS control method and system of the WPT system based on DC bias variable capacitor according to claim 3 is characterized in that: In step S5, the phase difference φ of the first-order harmonic component of the inverter output voltage relative to the asymmetric square wave signal is calculated according to the newly calculated α or β. v1 In step S3, the phase difference φ is obtained v1 After that, the current phase difference is the phase difference between the voltage and current measured by the primary side AC calculated by sampling minus the phase difference φ v1 .

5. The ZVS control method and system of the WPT system based on DC bias variable capacitor according to claim 4 is characterized in that: Phase difference φ v1 equal a1 and b1 are defined as: a1=sin(β-α) + )+sin(β)+sin(α - ), b1=1-cos(β-α + )-cos(β)+cos(α - ), among them - definition It is the angle between when the voltage changes from negative voltage to 0 voltage in one cycle and when the next cycle begins.

6. The ZVS control method and system of the WPT system based on DC bias variable capacitor according to claim 5, characterized in that: α - Always equal to 0°.

7. The ZVS control method and system of the WPT system based on DC bias variable capacitor according to claim 6, characterized in that: Reference phase difference Determined by the output capacitance of the MOSFET of the full-bridge inverter.

8. A ZVS control system for a WPT system based on a DC bias variable capacitor, characterized in that: It includes a primary side sampling module and a primary side control module, wherein the primary side sampling module is used to sample the voltage and current of the primary side AC measurement and send them to the primary side control module; the primary side control module is used to execute the remaining contents of the ZVS control method of the WPT system based on DC bias variable capacitor described in any one of claims 1 to 7 except sampling the voltage and current of the primary side AC measurement.

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