Permanent magnet synchronous motor control method of discrete type adaptive complex coefficient observer

Abstract

The invention discloses a permanent magnet synchronous motor control method of a discrete self-adaptive complex coefficient observer, and the method comprises the following steps: S10) taking collected motor stator current and motor stator voltage as the input of a sliding-mode observer; on this basis, a sliding mode area integral parameter, a sliding mode integral gain parameter, a sliding mode integral gain correction parameter and a sliding mode switching boundary parameter are set, a discrete adaptive complex coefficient sliding mode observer is established, and an extended counter electromotive force observation value is obtained; s20) taking the extended back electromotive force observation value as the input of a self-adaptive vector phase-locked loop, and extracting the estimated values of the electric angular speed and angle of the motor rotor; s30) taking the estimated value of the electrical angular velocity as a model correction variable of the sliding-mode observer, and feeding back and inputting the estimated value into the sliding-mode observer for closed-loop observation; high-rotating-speed non-inductive closed-loop control of the motor under the working condition that the carrier wave ratio is smaller than 10 can be achieved, and the requirement for the computing power of a processor in a high-rotating-speed operation occasion is effectively lowered.

Description

Technical Field

[0001] The invention relates to the technical field of low-carrier ratio sensorless control of permanent magnet synchronous motors, and in particular to a permanent magnet synchronous motor control method using a discrete adaptive complex coefficient observer. Background Art

[0002] In recent years, high-speed permanent magnet synchronous motors have been widely used in aerospace, flywheel energy storage, chip manufacturing and other high-tech fields due to their advantages such as fast dynamic response, small torque ripple coefficient and high power density. Since position sensors are difficult to install under high-speed and complex working conditions and are susceptible to large interference, traditional position sensors cannot be reliably applied to the field of high-speed permanent magnet synchronous motor control. However, the use of magnetic field oriented control for high-speed permanent magnet synchronous motors inevitably requires accurate rotor position acquisition, so research on position sensorless control of high-speed permanent magnet synchronous motors is of great significance.

[0003] At present, after theoretical analysis and experimental demonstration by a large number of experts and scholars, it is concluded that the main reasons for the poor robustness of high-speed permanent magnet synchronous motor position sensorless control are:

[0004] (1) During high-speed operation, any signal processing delay and phase error will be amplified, resulting in delayed response and instability of the control system.

[0005] (2) Although the motor back-EMF amplitude increases when running at high speed, it is helpful for the observation of the back-EMF observer, but it may also lead to the amplification of the estimation error, thereby reducing the speed estimation accuracy.

[0006] (3) In practice, after the algorithm design is completed, the designed algorithm is usually executed in a shorter time by increasing the processor computing power. However, due to external factors such as power consumption, the processor computing power that can be increased is limited, making it difficult to completely solve the above technical problems.

[0007] In the prior art, when the motor is running in the high-speed area, the sensorless algorithm usually used is to extract the motor electrical angular velocity and electrical angle by observing the extended back electromotive force. However, when the motor speed is getting faster and faster, the observed motor extended back electromotive force will lag as the carrier ratio decreases, resulting in large errors in the estimated motor electrical angular velocity and electrical angle, making it difficult for the control system to achieve accurate magnetic field oriented control, causing the motor to lose control. Summary of the invention

[0008] In view of this, the purpose of the present invention is to provide a permanent magnet synchronous motor control method based on a discrete adaptive complex coefficient observer, so as to realize high-speed sensorless closed-loop control of the motor under the working condition that the carrier ratio is less than 10, and effectively reduce the requirements on the processor computing power in high-speed operation occasions.

[0009] In order to achieve the above-mentioned invention object, the present invention specifically provides a technical implementation scheme of a permanent magnet synchronous motor control method of a discrete adaptive complex coefficient observer, and the permanent magnet synchronous motor control method of a discrete adaptive complex coefficient observer comprises the following steps:

[0010] S10) using the collected motor stator current and motor stator voltage as inputs of the sliding mode observer, setting sliding mode surface integral parameters, sliding mode integral gain parameters, sliding mode integral gain correction parameters, and sliding mode switching boundary parameters on this basis, establishing a discrete adaptive complex coefficient sliding mode observer, and obtaining an extended back electromotive force observation value;

[0011] S20) using the extended back electromotive force observation value as the input of the adaptive vector phase-locked loop to extract the estimated value of the motor rotor electrical angular velocity and angle;

[0012] S30) The estimated value of the electrical angular velocity is used as a model correction variable of the sliding mode observer and fed back to the sliding mode observer for closed-loop observation.

[0013] Further, the step S10) includes the following process:

[0014] S101) after obtaining the current observation error by subtracting the estimated current of the previous cycle and the electronic stator current collected in the current cycle, an integral sliding surface with the current observation error as the state variable is constructed using the preset sliding surface integral parameters;

[0015] S102) taking the absolute value of the calculated value of the integral sliding mode surface of the previous cycle as input, combining the preset sliding mode integral gain parameter and the sliding mode integral gain correction parameter to construct an adaptive sliding mode gain, on this basis taking the adaptive sliding mode gain calculated in the current cycle and the estimated value of the electrical angular velocity of the previous cycle as input, and multiplying them with the preset sliding mode switching boundary parameter to obtain the sliding mode switching function boundary value;

[0016] S103) Based on the output value of the sliding mode switching function, comprehensively considering the influence of the motor stator resistance and inductance to obtain an extended back electromotive force observation value that compensates for the motor characteristics;

[0017] S104) Using the motor stator voltage and the estimated value of the electrical angular velocity of the previous cycle as input, the motor equation modeled in a complex vector manner is introduced to obtain an estimated current that more precisely considers the coupling relationship of the motor's three-phase windings.

[0018] Furthermore, in the step S101), an integral sliding mode surface with the current observation error as the state variable is constructed according to the following formula:

[0019]

[0020] Where s(k) is the calculated value of the integral sliding surface of the current cycle, Δi s (k) is the current observation error of the current cycle, M is the preset sliding mode surface integral parameter, T s To control the execution cycle of the method, is the sum of the current observation errors accumulated to the previous cycle.

[0021] Furthermore, in the step S102), the adaptive sliding mode gain and the sliding mode switching function boundary value are calculated according to the following formula:

[0022] h asmo (k) = h asmo (k-1)+KT s (|s(k-1)|-ηh asmo (k-1))

[0023]

[0024] In the formula, h asmo (k) is the adaptive sliding mode gain of the current cycle, h asmo (k-1) is the adaptive sliding mode gain of the previous cycle, K is the preset sliding mode integral gain parameter, T s is the execution cycle of the control method, s(k-1) is the integral sliding surface calculation value of the previous cycle, η is the preset sliding mode integral gain correction parameter, Z0(k) is the boundary value of the sliding mode switching function, k z is the sliding mode switching boundary parameter, is the estimated value of the electrical angular velocity of the previous cycle, ψ f is the permanent magnet flux of the motor rotor, R s is the motor stator phase resistance, and L is the motor stator synchronous inductance.

[0025] Furthermore, in the step S103), a sliding mode switching function is constructed according to the following formula to calculate the extended back electromotive force observation value:

[0026]

[0027]

[0028] Where Z(k) is the output value of the sliding mode switching function in the current cycle, Δi s (k) is the current observation error of the current cycle, M is the preset sliding mode surface integral parameter, T s To control the execution cycle of the method, is the sum of the current observation errors accumulated to the previous cycle, is the extended back EMF observation value of the current cycle, R sis the motor stator phase resistance, j is an imaginary unit, is the estimated value of the electrical angular velocity of the previous cycle, L is the synchronous inductance of the motor stator, h asmo (k) is the adaptive sliding mode gain of the current cycle.

[0029] Furthermore, in step S104), the current is estimated according to the following formula:

[0030]

[0031]

[0032] In the formula, is the estimated current value of the next cycle of the α axis, is the estimated current value of the α axis in the current cycle, e is the natural logarithm, T s is the execution cycle of the control method, R s is the motor stator phase resistance, L is the motor stator synchronous inductance, Z α (k) is the output value of the sliding mode switching function of the current cycle of the α axis, Z β (k) is the output value of the sliding mode switching function of the current cycle of the β axis, cos() is the cosine function, sin() is the sine function, is the estimated value of the electrical angular velocity of the previous cycle, u α (k-1) is the calculated value of the motor stator α-axis voltage in the previous cycle, is the estimated current value of the next cycle of the β axis, is the estimated current value of the β-axis in the current cycle.

[0033] Furthermore, in step S20), the estimated values ​​of the motor rotor electrical angular velocity and angle are extracted according to the following formula:

[0034]

[0035]

[0036] In the formula, is the estimated value of the electrical angular velocity of the current cycle, T s To control the execution cycle of the method, is the estimated value of the electrical angular velocity of the previous cycle, is the estimated value of the electrical angular velocity of the previous cycle, k P is the proportional parameter of the adaptive vector phase-locked loop, k I is the integral parameter of the adaptive vector phase-locked loop, is the extended back EMF observation value of the current cycle of the α-axis, is the extended back EMF observation value of the current cycle of the β axis, is the estimated value of the electrical angle of the current cycle, It is the estimated value of the electrical angle of the previous cycle.

[0037] Furthermore, the sliding mode surface integral parameter M is 0 to R s L, sliding mode integral gain parameter K is 0 to 1, sliding mode integral gain correction parameter η is 0 to 0.1, sliding mode switching boundary parameter k z > 1. The proportional parameter k of the adaptive vector phase-locked loop P Take 10% to 20% of the motor frequency, the integral parameter k of the adaptive vector phase-locked loop I Take the square of the designed cutoff frequency.

[0038] Furthermore, the closed-loop observation part of the control method includes:

[0039] A discrete adaptive complex coefficient sliding mode observer is used to use the collected motor stator current and motor stator voltage as the input of the sliding mode observer, and on this basis, set the sliding mode surface integral parameter, sliding mode integral gain parameter, sliding mode integral gain correction parameter, and sliding mode switching boundary parameter, establish a discrete adaptive complex coefficient sliding mode observer, and obtain the extended back electromotive force observation value;

[0040] The adaptive vector phase-locked loop module is used to use the extended back-electromotive force observation value as the input of the adaptive vector phase-locked loop, extract the estimated value of the motor rotor electrical angular velocity and angle, use the estimated value of the electrical angular velocity as the model correction variable of the sliding mode observer, and feed it back to the sliding mode observer for closed-loop observation.

[0041] Furthermore, the execution cycle T of the control method s Under high speed conditions, the maximum value can be 0.0002 seconds.

[0042] By implementing the technical solution of the permanent magnet synchronous motor control method of the discrete adaptive complex coefficient observer provided by the present invention, the following beneficial effects are achieved:

[0043] (1) The present invention adopts a sensorless control method for a permanent magnet synchronous motor based on a discrete adaptive complex coefficient sliding mode observer, and achieves accurate acquisition of extended back electromotive force observation values ​​under low-carrier conditions by targeted setting of sliding mode surface integral parameters, sliding mode integral gain parameters, sliding mode integral gain correction parameters, and sliding mode switching boundary parameters, thereby effectively reducing the requirements for processor computing power for sensorless estimation in high-speed operation occasions, thereby well alleviating the technical problem of difficulty in controlling low-carrier conditions when the permanent magnet synchronous motor is running at high speed;

[0044] (2) The present invention can accurately estimate the rotor position and speed of the permanent magnet synchronous motor directly based on the sampled current and the voltage calculated in the previous cycle without adding position sensors and other electrical signal sampling structures. It not only solves the technical problem of poor position estimation accuracy of the permanent magnet synchronous motor under high-speed conditions, but also reduces the requirements on processor computing power, thereby reducing costs and improving the system's adaptability under complex working conditions. It has important theoretical significance and engineering application value. BRIEF DESCRIPTION OF THE DRAWINGS

[0045] In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the following briefly introduces the drawings required for use in the embodiments or the prior art descriptions. Obviously, the drawings described below are only some embodiments of the present invention, and for ordinary technicians in this field, other embodiments can be obtained based on these drawings without creative work.

[0046] Figure 1 is a flow chart of a permanent magnet synchronous motor control method using a discrete adaptive complex coefficient observer according to an embodiment of the present invention;

[0047] Figure 2 is a system schematic diagram of a permanent magnet synchronous motor position sensorless control system based on a discrete adaptive complex coefficient sliding mode observer provided in an embodiment of the present invention;

[0048] Figure 3 is a control block diagram of a discrete adaptive complex coefficient sliding mode observer provided according to an embodiment of the present invention;

[0049] Figure 4 is a control block diagram of an adaptive vector phase-locked loop provided according to an embodiment of the present invention;

[0050] Figure 5 It is a structural block diagram of the closed-loop observation part of the permanent magnet synchronous motor control method of the discrete adaptive complex coefficient observer provided in an embodiment of the present invention;

[0051] Figure 6 is the execution period T of the control method provided by the embodiment of the present invention s Take the schematic diagram of the high speed dynamic response simulation results at 0.0002 seconds;

[0052] Figure 7 is the execution period T of the control method provided by the embodiment of the present invention s Schematic diagram of simulation results of continuous operation at the limit speed of 0.0002 seconds;

[0053] Figure 8 is the execution period T of the control method provided by the embodiment of the present invention sSchematic diagram of simulation results of actual and estimated values ​​of the motor electrical angle at the limit speed state at 0.0002 seconds;

[0054] In the figure: 51 - discrete adaptive complex coefficient sliding mode observer, 52 - adaptive vector phase-locked loop. DETAILED DESCRIPTION

[0055] In order to make the purpose, technical solution and advantages of the embodiments of the present invention clearer, the technical solution in the embodiments of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0056] As attached Figure 1 To Attachment Figure 8 As shown, a specific embodiment of the permanent magnet synchronous motor control method of the discrete adaptive complex coefficient observer of the present invention is given, and the present invention is further described below in conjunction with the drawings and specific embodiments.

[0057] As attached Figure 1 As shown, an embodiment of the present invention provides a permanent magnet synchronous motor control method using a discrete adaptive complex coefficient observer, the method comprising the following steps:

[0058] S10) using the collected motor stator current and motor stator voltage as inputs of the sliding mode observer, setting sliding mode surface integral parameters, sliding mode integral gain parameters, sliding mode integral gain correction parameters, and sliding mode switching boundary parameters on this basis, establishing a discrete adaptive complex coefficient sliding mode observer, and obtaining an extended back electromotive force observation value;

[0059] S20) using the extended back electromotive force observation value as the input of the adaptive vector phase-locked loop to extract the estimated value of the motor rotor electrical angular velocity and angle;

[0060] S30) The estimated value of the electrical angular velocity is used as a model correction variable of the sliding mode observer and fed back to the sliding mode observer for closed-loop observation.

[0061] The present invention uses the stator current of the motor as a reference, combines the complex vector model of the motor, and builds a discrete adaptive complex coefficient sliding mode observer. By presetting the sliding mode surface integral parameter, sliding mode integral gain parameter, sliding mode integral gain correction parameter, and sliding mode switching boundary parameter, it is used to accurately obtain the observation value of the motor's extended back electromotive force under low carrier ratio working conditions. The obtained extended back electromotive force observation value is used as the input of the adaptive vector phase-locked loop, and the estimated value of the motor rotor's electrical angular velocity and electrical angle is extracted through the adaptive phase-locked loop. The motor's electrical angular velocity estimate is further used as the model correction variable of the sliding mode observer, and fed back to the sliding mode observer to realize closed-loop observation. This method effectively reduces the requirements for processor computing power for sensorless estimation in high-speed operation occasions, thereby well alleviating the technical problem of low-carrier working condition control difficulties when the permanent magnet synchronous motor is running at high speed.

[0062] Step S10) further comprises:

[0063] S101) after obtaining the current observation error by subtracting the estimated current of the previous cycle and the electronic stator current collected in the current cycle, an integral sliding surface with the current observation error as the state variable is constructed using the preset sliding surface integral parameters;

[0064] S102) taking the absolute value of the calculated value of the integral sliding mode surface of the previous cycle as input, combining the preset sliding mode integral gain parameter and the sliding mode integral gain correction parameter to construct an adaptive sliding mode gain, on this basis taking the adaptive sliding mode gain calculated in the current cycle and the estimated value of the electrical angular velocity of the previous cycle as input, and multiplying them with the preset sliding mode switching boundary parameter to obtain the sliding mode switching function boundary value;

[0065] S103) Based on the output value of the sliding mode switching function, comprehensively considering the influence of the motor stator resistance and inductance to obtain an extended back electromotive force observation value that compensates for the motor characteristics;

[0066] S104) Using the motor stator voltage and the estimated value of the electrical angular velocity of the previous cycle as input, the motor equation modeled in a complex vector manner is introduced to obtain an estimated current that more precisely considers the coupling relationship of the motor's three-phase windings.

[0067] As attached Figure 2 The block diagram of the permanent magnet synchronous motor position sensorless control system based on discrete adaptive complex coefficient sliding mode observer is shown in FIG. When the motor is in high-speed operation, the three-phase stator current i of the motor is obtained by current sampling. a 、i b 、i c , and then the two-phase stationary coordinate system current i is obtained through Clarke transformation α and i βOn this basis, the motor electrical angle estimation value extracted by the adaptive phase-locked loop according to the present invention is used to calculate the two-phase stationary coordinate system current i α and i β Perform Park transformation to obtain the two-phase rotating coordinate system current i d and i q Then, after subtracting the estimated value of the motor electrical angular velocity extracted by the adaptive phase-locked loop according to the present invention from the given speed, the q-axis current given value is obtained through speed PI control, and then u is obtained by subtracting the currents of the two-phase rotating coordinate system respectively and then controlling the current PI. d and u q , and then use the motor electrical angle estimation value extracted by the adaptive phase-locked loop of the present invention to obtain u d and u q Perform Park inverse transformation to obtain the voltage given value u of the two-phase stationary coordinate system α and u β Finally, the SVPWM module is input to adjust the duty cycle of each bridge arm of the three-phase inverter, and then the three-phase voltage is output to realize the sensorless closed-loop control of the motor.

[0068] As attached Figure 3 The control block diagram of the discrete adaptive complex coefficient sliding mode observer is shown in FIG. The extended back electromotive force observation value obtained by the discrete adaptive complex coefficient sliding mode observer and It is used as the input of the adaptive vector phase-locked loop to extract the estimated values ​​of the motor rotor electrical angular velocity and angle.

[0069] As a specific embodiment of the present invention, an integral sliding mode surface with the current observation error as the state variable is constructed according to the following formula:

[0070]

[0071] Where s(k) is the calculated value of the integral sliding surface of the current cycle, Δi s (k) is the current observation error of the current cycle, M is the preset sliding mode surface integral parameter, T s To control the execution cycle of the method, is the sum of the current observation errors accumulated to the previous cycle.

[0072] The adaptive sliding mode gain and sliding mode switching function boundary value are further calculated according to the following formula:

[0073] h asmo (k) = h asmo (k-1)+KT s (|s(k-1)|-ηh asmo (k-1))

[0074]

[0075] In the formula, h asmo (k) is the adaptive sliding mode gain of the current cycle, h asmo (k-1) is the adaptive sliding mode gain of the previous cycle, K is the preset sliding mode integral gain parameter, T s is the execution cycle of the control method, s(k-1) is the integral sliding surface calculation value of the previous cycle, η is the preset sliding mode integral gain correction parameter, Z0(k) is the boundary value of the sliding mode switching function, k z is the sliding mode switching boundary parameter, is the estimated value of the electrical angular velocity of the previous cycle, ψ f is the permanent magnet flux of the motor rotor, R s is the motor stator phase resistance, and L is the motor stator synchronous inductance.

[0076] The sliding mode switching function is further constructed according to the following formula to calculate the extended back electromotive force observation value:

[0077]

[0078]

[0079] Where Z(k) is the output value of the sliding mode switching function in the current cycle, Δi s (k) is the current observation error of the current cycle, M is the preset sliding mode surface integral parameter, T s To control the execution cycle of the method, is the sum of the current observation errors accumulated to the previous cycle, is the extended back EMF observation value of the current cycle, R s is the motor stator phase resistance, j is the imaginary unit, is the estimated value of the electrical angular velocity of the previous cycle, L is the synchronous inductance of the motor stator, h asmo (k) is the adaptive sliding mode gain of the current cycle.

[0080] It should be noted that in the formula representing the relationship between the output value of the sliding mode switching function of the current cycle and the extended back-electromotive force observation value of the current cycle, parameter items related to the motor stator phase resistance and the motor stator synchronous inductance are introduced to compensate for the voltage drop caused by the motor stator phase resistance and the motor stator synchronous inductance, thereby improving the accuracy of the extended back-electromotive force observation.

[0081] The current is further estimated according to the following formula:

[0082]

[0083]

[0084] In the formula, is the estimated current value of the next cycle of the α axis, is the estimated current value of the α axis in the current cycle, e is the natural logarithm, T s is the execution cycle of the control method, R s is the motor stator phase resistance, L is the motor stator synchronous inductance, Z α (k) is the output value of the sliding mode switching function of the current cycle of the α axis, Z β (k) is the output value of the sliding mode switching function of the current cycle of the β axis, cos() is the cosine function, sin() is the sine function, is the estimated value of the electrical angular velocity of the previous cycle, u α (k-1) is the calculated value of the motor stator α-axis voltage in the previous cycle, is the estimated current value of the next cycle of the β axis, is the estimated current value of the β-axis in the current cycle.

[0085] In this embodiment, the motor model modeled by the complex vector method is simplified by taking the α axis in the two-phase stationary coordinate system as the real axis and the β axis as the imaginary axis, so as to achieve more accurate current estimation. It should be noted that although the above formula represents the relationship between the current estimation value of the current cycle and the current estimation value of the next cycle, when the actual control algorithm is running, the current estimation value of the previous cycle is used to obtain the current estimation value of the current cycle.

[0086] As attached Figure 4 The control block diagram of the adaptive vector phase-locked loop is shown in Figure 1. The extended back electromotive force obtained by the discrete adaptive complex coefficient sliding mode observer and As input, the estimated value of the motor rotor electrical angular velocity is extracted through the adaptive vector proportional integral link it contains and electrical angle estimate

[0087] As a specific embodiment of the present invention, the estimated values ​​of the electrical angular velocity and angle of the motor rotor are extracted according to the following formula:

[0088]

[0089]

[0090] In the formula, is the estimated value of the electrical angular velocity of the current cycle, T s To control the execution cycle of the method, is the estimated value of the electrical angular velocity of the previous cycle, is the estimated value of the electrical angular velocity of the previous cycle, k Pis the proportional parameter of the adaptive vector phase-locked loop, k I is the integral parameter of the adaptive vector phase-locked loop, is the extended back EMF observation value of the current cycle of the α-axis, is the extended back EMF observation value of the current cycle of the β axis, is the estimated value of the electrical angle of the current cycle, It is the estimated value of the electrical angle of the previous cycle.

[0091] As attached Figure 5 As shown, the present invention provides a closed-loop observation part of a permanent magnet synchronous motor control method of a discrete adaptive complex coefficient observer according to an embodiment, and the closed-loop observation part includes:

[0092] A discrete adaptive complex coefficient sliding mode observer is used to use the collected motor stator current and motor stator voltage as the input of the sliding mode observer, and on this basis, set the sliding mode surface integral parameter, sliding mode integral gain parameter, sliding mode integral gain correction parameter, and sliding mode switching boundary parameter, establish a discrete adaptive complex coefficient sliding mode observer, and obtain the extended back electromotive force observation value;

[0093] The adaptive vector phase-locked loop module is used to use the extended back-electromotive force observation value as the input of the adaptive vector phase-locked loop, extract the estimated value of the motor rotor electrical angular velocity and angle, use the estimated value of the electrical angular velocity as the model correction variable of the sliding mode observer, and feed it back to the sliding mode observer for closed-loop observation.

[0094] Furthermore, the sliding mode surface integral parameter M is 0 to R s L, sliding mode integral gain parameter K is 0 to 1, sliding mode integral gain correction parameter η is 0 to 0.1, sliding mode switching boundary parameter k z > 1. The proportional parameter k of the adaptive vector phase-locked loop P Take 10% to 20% of the motor frequency, the integral parameter k of the adaptive vector phase-locked loop I Take the square of the designed cutoff frequency.

[0095] In a specific embodiment of the present invention, the parameters of the permanent magnet synchronous motor used are as shown in the following table:

[0096] parameter Parameter Value <![CDATA[DC supply voltage V dc (V)]]> 48 Pole pair number p 1 <![CDATA[Phase resistance R s (Ω)]]> 0.0255 Synchronous inductance (500Hz) L (μH) 18.48 <![CDATA[Rotor permanent magnet flux linkage ψ f (Wb)]]> 0.0011

[0097] Furthermore, the execution cycle T of the control method s Under high speed conditions, the maximum value can be 0.0002 seconds.

[0098] As attached Figure 6 The execution cycle T is shown sThe schematic diagram of the dynamic response experiment results of the motor running without induction at high speed at 0.0002 seconds shows that when the speed command changes continuously between 15000 and 30000, the estimated speed of the motor can quickly follow the speed setting value, and still has a good dynamic response effect under low frequency ratio conditions. Figure 7 is the execution cycle T s The schematic diagram of the continuous running experiment results of the motor's sensorless operation at the limit speed at 0.0002 seconds. It can be seen that under the limit state of the speed given as 33000, the discrete adaptive complex coefficient sliding mode observer disclosed in the present invention can still accurately obtain the estimated value of the motor speed and realize stable sensorless closed-loop control. Figure 8 is the execution cycle T s The schematic diagram of the experimental results of the motor electrical angle estimation when the motor is running without sensor at the limit speed at 0.0002 seconds. It can be seen that when the control algorithm is executed only 9 times in one electrical cycle of the motor, a relatively accurate motor electrical angle estimation value can still be extracted.

[0099] By implementing the technical solution of the permanent magnet synchronous motor control method of the discrete adaptive complex coefficient observer described in the specific embodiment of the present invention, the following technical effects can be produced:

[0100] (1) The permanent magnet synchronous motor control method of the discrete adaptive complex coefficient observer described in the specific embodiment of the present invention adopts the permanent magnet synchronous motor sensorless control method based on the discrete adaptive complex coefficient sliding mode observer, and achieves accurate acquisition of the extended back electromotive force observation value under low carrier working conditions by setting the sliding mode surface integral parameter, sliding mode integral gain parameter, sliding mode integral gain correction parameter, and sliding mode switching boundary parameter in a targeted manner, thereby effectively reducing the requirements of the processor computing power for the sensorless estimation in high-speed operation occasions, thereby well alleviating the technical problem of the difficulty of controlling the low carrier working condition when the permanent magnet synchronous motor is running at high speed;

[0101] (2) The permanent magnet synchronous motor control method of the discrete adaptive complex coefficient observer described in the specific embodiment of the present invention can accurately estimate the rotor position and speed of the permanent magnet synchronous motor directly based on the sampled current and the voltage calculated in the previous cycle without adding position sensors and other electrical signal sampling structures. It not only solves the technical problem of poor position estimation accuracy of the permanent magnet synchronous motor under high-speed conditions, but also reduces the requirements on processor computing power, thereby reducing costs and improving the adaptability of the system under complex working conditions. It has important theoretical significance and engineering application value.

[0102] The various embodiments in this specification are described in a progressive manner, and each embodiment focuses on the differences from other embodiments. The same or similar parts between the various embodiments can be referenced to each other.

[0103] The above is only a preferred embodiment of the present invention and does not limit the present invention in any form. Although the present invention has been disclosed as above in the preferred embodiment, it is not used to limit the present invention. Any technician familiar with the art can make many possible changes and modifications to the technical solution of the present invention by using the methods and technical contents disclosed above without departing from the spirit and technical solution of the present invention, or modify it into an equivalent embodiment of equivalent changes. Therefore, any simple modification, equivalent replacement, equivalent change and modification made to the above embodiments according to the technical essence of the present invention without departing from the content of the technical solution of the present invention still falls within the scope of protection of the technical solution of the present invention.

Claims

1. A method for controlling a permanent magnet synchronous motor using a discrete adaptive complex coefficient observer, characterized in that: The following steps are involved: S10) using the collected motor stator current and motor stator voltage as inputs of the sliding mode observer, setting sliding mode surface integral parameters, sliding mode integral gain parameters, sliding mode integral gain correction parameters, and sliding mode switching boundary parameters on this basis, establishing a discrete adaptive complex coefficient sliding mode observer, and obtaining an extended back electromotive force observation value; S20) using the extended back electromotive force observation value as the input of the adaptive vector phase-locked loop to extract the estimated value of the motor rotor electrical angular velocity and angle; S30) The estimated value of the electrical angular velocity is used as a model correction variable of the sliding mode observer and fed back to the sliding mode observer for closed-loop observation.

2. The permanent magnet synchronous motor control method of discrete adaptive complex coefficient observer according to claim 1, characterized in that: Step S10) The process includes: S101) after obtaining the current observation error by subtracting the estimated current of the previous cycle and the electronic stator current collected in the current cycle, an integral sliding surface with the current observation error as the state variable is constructed using the preset sliding surface integral parameters; S102) taking the absolute value of the calculated value of the integral sliding mode surface of the previous cycle as input, combining the preset sliding mode integral gain parameter and the sliding mode integral gain correction parameter to construct an adaptive sliding mode gain, on this basis taking the adaptive sliding mode gain calculated in the current cycle and the estimated value of the electrical angular velocity of the previous cycle as input, and multiplying them with the preset sliding mode switching boundary parameter to obtain the sliding mode switching function boundary value; S103) Based on the output value of the sliding mode switching function, comprehensively considering the influence of the motor stator resistance and inductance to obtain an extended back electromotive force observation value that compensates for the motor characteristics; S104) Using the motor stator voltage and the estimated value of the electrical angular velocity of the previous cycle as input, the motor equation modeled in a complex vector manner is introduced to obtain an estimated current that more precisely considers the coupling relationship of the motor's three-phase windings.

3. The permanent magnet synchronous motor control method of discrete adaptive complex coefficient observer according to claim 1, characterized in that: In the step S101), an integral sliding mode surface with the current observation error as the state variable is constructed according to the following formula: Where s(k) is the calculated value of the integral sliding surface of the current cycle, Δi s (k) is the current observation error of the current cycle, M is the preset sliding mode surface integral parameter, T s To control the execution cycle of the method, is the total current observation error accumulated to the previous cycle.

4. The permanent magnet synchronous motor control method of discrete adaptive complex coefficient observer according to claim 1, characterized in that: In the step S102), the adaptive sliding mode gain and the sliding mode switching function boundary value are calculated according to the following formula: h asmo (k)=h asmo (k-1)+KT s (|s(k-1)|-ηh asmo (k-1)) In the formula, h asmo (k) is the adaptive sliding mode gain of the current cycle, h asmo (k-1) is the adaptive sliding mode gain of the previous cycle, K is the preset sliding mode integral gain parameter, T s is the execution cycle of the control method, s(k-1) is the integral sliding surface calculation value of the previous cycle, η is the preset sliding mode integral gain correction parameter, Z0(k) is the boundary value of the sliding mode switching function, k z is the sliding mode switching boundary parameter, is the estimated value of the electrical angular velocity of the previous cycle, ψ f is the permanent magnet flux of the motor rotor, R s is the motor stator phase resistance, and L is the motor stator synchronous inductance.

5. The permanent magnet synchronous motor control method of discrete adaptive complex coefficient observer according to claim 1, characterized in that: In the step S103), a sliding mode switching function is constructed according to the following formula to calculate the extended back electromotive force observation value: Where Z(k) is the output value of the sliding mode switching function in the current cycle, Δi s (k) is the current observation error of the current cycle, M is the preset sliding mode surface integral parameter, T s To control the execution cycle of the method, is the sum of the current observation errors accumulated to the previous cycle, is the extended back EMF observation value of the current cycle, R s is the motor stator phase resistance, j is the imaginary unit, is the estimated value of the electrical angular velocity of the previous cycle, L is the synchronous inductance of the motor stator, h asmo (k) is the adaptive sliding mode gain of the current cycle.

6. The method for controlling a permanent magnet synchronous motor using a discrete adaptive complex coefficient observer according to claim 1, characterized in that: In the step S104), the current is estimated according to the following formula: In the formula, is the estimated current value of the next cycle of the α axis, is the estimated current value of the α axis in the current cycle, e is the natural logarithm, T s is the execution cycle of the control method, R s is the motor stator phase resistance, L is the motor stator synchronous inductance, Z α (k) is the output value of the sliding mode switching function of the current cycle of the α axis, Z β (k) is the output value of the sliding mode switching function of the current cycle of the β axis, cos() is the cosine function, sin() is the sine function, is the estimated value of the electrical angular velocity of the previous cycle, u α (k-1) is the calculated value of the motor stator α-axis voltage in the previous cycle, is the estimated current value of the next cycle of the β axis, is the estimated current value of the β-axis in the current cycle.

7. The permanent magnet synchronous motor control method of discrete adaptive complex coefficient observer according to claim 1, characterized in that: In step S20), the estimated values ​​of the motor rotor electrical angular velocity and angle are extracted according to the following formula: In the formula, is the estimated value of the electrical angular velocity of the current cycle, T s To control the execution cycle of the method, is the estimated value of the electrical angular velocity of the previous cycle, is the estimated value of the electrical angular velocity of the previous cycle, k P is the proportional parameter of the adaptive vector phase-locked loop, k I is the integral parameter of the adaptive vector phase-locked loop, is the extended back EMF observation value of the current cycle of the α-axis, is the extended back EMF observation value of the current cycle of the β axis, is the estimated value of the electrical angle of the current cycle, It is the estimated value of the electrical angle of the previous cycle.

8. The permanent magnet synchronous motor control method using a discrete adaptive complex coefficient observer according to claim 1, characterized in that: The sliding mode surface integral parameter M ranges from 0 to R s L, sliding mode integral gain parameter K is 0 to 1, sliding mode integral gain correction parameter η is 0 to 0.1, sliding mode switching boundary parameter k z > 1. The proportional parameter k of the adaptive vector phase-locked loop P Take 10% to 20% of the motor frequency, the integral parameter k of the adaptive vector phase-locked loop I Take the square of the designed cutoff frequency.

9. The method for controlling a permanent magnet synchronous motor using a discrete adaptive complex coefficient observer according to claim 1, characterized in that: The closed-loop observation part of the control method includes: A discrete adaptive complex coefficient sliding mode observer is used to use the collected motor stator current and motor stator voltage as the input of the sliding mode observer, and on this basis, set the sliding mode surface integral parameter, sliding mode integral gain parameter, sliding mode integral gain correction parameter, and sliding mode switching boundary parameter, establish a discrete adaptive complex coefficient sliding mode observer, and obtain the extended back electromotive force observation value; The adaptive vector phase-locked loop module is used to use the extended back-electromotive force observation value as the input of the adaptive vector phase-locked loop, extract the estimated value of the motor rotor electrical angular velocity and angle, use the estimated value of the electrical angular velocity as the model correction variable of the sliding mode observer, and feed it back to the sliding mode observer for closed-loop observation.

10. The method for controlling a permanent magnet synchronous motor using a discrete adaptive complex coefficient observer according to claim 1, characterized in that: The execution period T of the control method s Under high speed conditions, the maximum value can be 0.0002 seconds.

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