Direct-current-bias-free DAB converter phase shift control method

Abstract

The invention discloses a DAB converter phase shift control method without DC bias, which introduces a new control variable D4 to control a driving signal, and is suitable for various phase shift control strategies such as single phase shift, expanded phase shift, double phase shift, three phase shift and the like. According to the method, only input voltage and output voltage values need to be sampled, and an additional hardware circuit does not need to be added. The initial value and the final value of the auxiliary inductive current in a single switching period are both 0, and the inductive current in each switching period is completely symmetrical, so that the problem of power shortage of the first half switching period in a common direct current bias elimination method is solved, and good dynamic response performance is ensured from control. The method is simple in implementation process, and meanwhile good dynamic response performance is guaranteed.

Description

Technical Field

[0001] The invention relates to the field of power electronic control, in particular to a phase shift control method of a DAB converter without direct current bias. Background Art

[0002] DAB converters have been widely used in energy storage systems, electric vehicle charging, DC transformers and other fields due to their excellent soft switching characteristics, wide range voltage regulation capability, electrical isolation, and large transmission power level (L.Li, G.Xu, D.Sha, Y.Liu, Y.Sun and M.Su, "Review of Dual-Active-Bridge Converters With Topological Modifications," in IEEE Transactions on Power Electronics, vol.38, no.7, pp.9046-9076, July 2023, doi:10.1109 / TPEL.2023.3258418.). In terms of control, DAB converters widely adopt single phase shift (SPS), extended phase shift (EPS), double phase shift (DPS), three phase shift (TPS) and other phase shift control strategies to change the direction and size of the converter transmission power by controlling the phase shift angle. However, during the process of converter startup or load switching, the phase shift angle of adjacent switching cycles will change suddenly, which will cause DC bias current to appear in the high-frequency transformer and auxiliary inductor (S.Mu, Z.Guo and Y.Luo, "Universal Modulation Scheme to Suppress TransientDC Bias Current in Dual Active Bridge Converters," in IEEE Transactions on Power Electronics, vol. 37, no. 2, pp. 1322-1333, Feb. 2022, doi: 10.1109 / TPEL.2021.3104628.). This DC bias increases the current stress of the switching device, reduces the converter efficiency, and shortens the service life of the switching device. Excessive DC bias current may even cause the magnetic circuit of the transformer and auxiliary inductor to saturate, and in severe cases, directly burn the circuit.

[0003] At present, there are two main methods for suppressing and eliminating DC bias current in control: one is to correct the inductor current in the first half of the switching cycle (Chen Yandong, Wang Zili, Liang Yunfei, etc. DAB converter inductor current DC bias elimination method and control system [P]. Zhejiang Province: CN115995981A, 2023-04-21.); the other is to add an additional control time between two adjacent switching cycles to correct the inductor current (Yang Ping, Wu Wenrong, Wang Mixin, etc. A DAB four-degree-of-freedom asymmetric modulation transient bias suppression method [P]. Sichuan Province: CN117013793A, 2023-11-07.). Method 1 will cause the transmission power of the first half of the switching cycle and the second half of the switching cycle to be unbalanced, resulting in a power shortage; Method 2 has high control complexity and cannot guarantee the size of the transmission power during the transient process. Although both existing control methods suppress the amplitude of the current peak, they both reduce the dynamic response speed of the system. Summary of the invention

[0004] The technical problem to be solved by the present invention is to provide a phase shift control method for a DAB converter without DC bias in view of the shortcomings of the prior art, so as to completely eliminate the generation of DC bias.

[0005] In order to solve the above technical problems, the technical solution adopted by the present invention is: a phase shift control method of a DAB converter without DC bias, comprising the following steps:

[0006] S1, sampling DAB converter input voltage V in With the output voltage V o ;

[0007] S2, output reference voltage V oref With the output voltage V o The voltage error is obtained by performing the difference, and the voltage error is used as the input of the voltage controller to obtain the output power reference value P oref ;

[0008] S3. Define the current reference power P b =N*V in *V o / (2*f s *L r ), where N is the transformer ratio, f s is the switching frequency, L r is the auxiliary inductor value, and the per-unit power is defined as P n =P oref / P b , define the voltage conversion ratio d = N * V o / V in; Define D1 and D2 as the duty cycle of the primary and secondary full-bridge voltage waveforms, D3 as the phase shift angle of the rising edge of the primary and secondary full-bridge voltage waveforms, and D4 as the phase shift angle of the forward shift of the drive waveform;

[0009] S4, based on the standard power P n The value of the voltage conversion ratio d determines the working mode of the DAB converter, and calculates the values ​​of D1, D2, and D3 corresponding to the current mode. in 、V o And the phase shift angle D4 of the driving waveform forward is obtained by calculating the D1, D2, and D3 sub-modal modes;

[0010] S5. Generate a PWM signal according to the obtained values ​​of D1, D2, D3, and D4, and use the PWM signal as a driving signal for the DAB converter.

[0011] The present invention introduces a new control degree of freedom D4 to control the drive signal, so that the initial value and the final value of the inductor current in a single switching cycle are both 0, realizing the decoupling of the inductor current in adjacent switching cycles, and being able to completely eliminate the generation of DC bias in control, thereby ensuring the smooth transition of the DAB converter in transient processes such as load jump; the initial value and the final value of the inductor current in each switching cycle can be made 0, realizing the decoupling of the inductor current in adjacent switching cycles, completely eliminating the DC bias current of the inductor from the control, and solving the problem of power shortage in the first half of the switching cycle at the same time, with good dynamic response performance.

[0012] The operating modes of the DAB converter include: when d < 1 and 0 ≤ P n ≤d*(1-d) / 2, the DAB converter operates in mode 1f; when d<1 and d*(1-d) / 2 <P n ≤0.25, the DAB converter operates in mode 2f; when d>1 and 0≤P n ≤(d-1) / 2 / d 2 When d>1 and (d-1) / 2 / d 2 <P n ≤0.25, the DAB converter operates in mode 4f; when d<1 and -d*(1-d) / 2≤P n <0, the DAB converter operates in mode 1r; when d<1 and -0.25≤P n <-d*(1-d) / 2, the DAB converter operates in mode 2r; when d>1 and -(d-1) / 2 / d 2 ≤P n <0, the DAB converter operates in mode 3r; when d>1 and -0.25≤P n <(d-1) / 2 / d 2When , the DAB converter works in mode 4r; among them, mode 1f, mode 2f, mode 3f, and mode 4f work in the forward power transmission mode; mode 1r, mode 2r, mode 3r, and mode 4r work in the reverse power transmission mode.

[0013] The expressions of D1, D2, and D3 are as follows:

[0014]

[0015]

[0016]

[0017]

[0018]

[0019]

[0020]

[0021]

[0022] The calculation formula of the phase shift angle D4 of the driving waveform is:

[0023]

[0024] According to the phase shift angle D4 of the driving waveform, the driving waveform generated by the duty ratios D1, D2, and D3 is corrected.

[0025] Compared with the prior art, the beneficial effects of the present invention are as follows: in order to eliminate the DC bias current that occurs during the transient processes of the DAB converter such as startup and load switching, the present invention proposes a phase shift control method without DC bias. The proposed control method introduces a new control variable D4 to control the drive signal, and is applicable to a variety of phase shift control strategies such as single phase shift, extended phase shift, double phase shift, and triple phase shift. The proposed method only needs to sample the input voltage and output voltage values, without adding additional hardware circuits. The initial value and final value of the auxiliary inductor current are both 0 in a single switching cycle, and the inductor current is completely symmetrical in each switching cycle, which solves the problem of power shortage in the first half of the switching cycle in the commonly used DC bias elimination method, and ensures good dynamic response performance from the control perspective. The implementation process of the present invention is simple, and at the same time, good dynamic response performance is guaranteed. BRIEF DESCRIPTION OF THE DRAWINGS

[0026] Figure 1 It is the topological structure diagram of DAB converter;

[0027] Figure 2The schematic diagram of the three-phase shift modulation strategy and inductor current waveform of the DAB converter;

[0028] Figure 3 The driving and current waveforms without and with DC bias optimization during forward power transmission;

[0029] Figure 4 The driving and current waveforms without and with DC bias optimization during reverse power transmission;

[0030] Figure 5 A control block diagram of a control method according to an embodiment of the present invention;

[0031] Figure 6 The inductor current waveform is not optimized for DC bias when the load changes;

[0032] Figure 7 Inductor current waveform optimized for adding DC bias when load changes. DETAILED DESCRIPTION

[0033] In order to make the purpose, technical solution and advantages of the embodiments of the present invention clearer, the technical solution in the embodiments of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. Obviously, the described embodiments are part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0034] like Figure 1 The topological structure of the DAB converter is shown in Figure 1. The converter consists of eight switching tubes and a high-frequency transformer. P1, P2, P3, and P4 are the primary switching devices of the converter, S1, S2, S3, and S4 are the secondary switching devices of the converter, and T r With L r They are high frequency transformer and auxiliary inductor respectively, and the transformer ratio is N:1. Its input end is connected to a DC power supply, and the input voltage is V in , C i is the input filter capacitor; the output end is connected to the load R, and the output voltage is V o , C ois the output filter capacitor. The source of the switch tube P1 is connected to the drain of P2 and connected to the same-name end of the primary winding of the transformer; the source of the switch tube P3 is connected to the drain of P4 and connected to the non-same-name end of the primary winding of the transformer; the drains of the switch tubes P1 and P3 are both connected to the positive pole of the voltage source, and the sources of the switch tubes P2 and P4 are both connected to the negative pole of the voltage source. The source of the switch tube S1 is connected to the drain of S2 and connected to the same-name end of the secondary winding of the transformer; the source of the switch tube S3 is connected to the drain of S4 and connected to the non-same-name end of the secondary winding of the transformer; the drains of the switch tubes S1 and S3 are both connected to the positive end of the load, and the sources of the switch tubes S2 and S4 are both connected to the negative end of the load.

[0035] like Figure 2 The figure shows the schematic diagram of the three-phase shift modulation strategy and the inductor current waveform of the DAB converter. D1 and D2 are the duty cycles of the primary and secondary full-bridge voltage waveforms, respectively, and D3 is the phase shift angle of the rising edge of the primary and secondary full-bridge voltage waveforms. D1′, D2′, and D3′ are the corresponding values ​​of the three phase shift angles in the second switching cycle. When the load suddenly changes and causes the phase shift angle of two adjacent switching cycles to change, the inductor current will not change suddenly, which will cause a DC bias in the next switching cycle, such as Figure 2 The ideal inductor current in the second switching cycle does not have a DC bias, as shown in the dashed line. Figure 2 As shown by the gray solid line. In the actual circuit, the DC bias current of the inductor will gradually approach 0 due to the damping in the circuit. However, the DC bias in the transient process will produce a large current peak, which seriously threatens the normal and stable operation of the DAB converter.

[0036] The basic principle of the phase shift control method for a DAB converter without DC bias proposed in the embodiment of the present invention is as follows: Figure 3 and Figure 4 shown. Figure 3 (a) shows the driving signal and inductor current waveform during forward power transmission without DC bias optimization; Figure 3 (b) shows the driving signal and inductor current waveform after DC bias optimization during forward power transmission; Figure 4 (a) shows the driving signal and inductor current waveform when reverse power transmission is not optimized for DC bias; Figure 4 (b) shows the driving signal and inductor current waveform after DC bias optimization during reverse power transmission.

[0037] For forward power transfer, Figure 3 As shown, by sampling the input voltage V in , output voltage V o, and the calculated D1, D2, D3, the phase shift angle D4 at the zero-crossing point of the inductor current can be calculated. At this time, D4>0, and the generation of the drive signal can be adjusted by D4 to obtain the optimized drive signal as follows Figure 3 As shown in (b) in .

[0038] For reverse power transfer, such as Figure 4 As shown, a similar method can be used to calculate the phase shift angle D4 at the zero-crossing point of the inductor current. At this time, D4<0. By adjusting the generation of the drive signal through D4, the optimized drive signal can be obtained as shown in Figure 4 As shown in (b) in .

[0039] By introducing a new control variable D4 to adjust the drive waveform, the initial and final values ​​of the inductor current in each switching cycle can be made 0, achieving the decoupling of the inductor current in adjacent switching cycles, completely eliminating the DC bias current of the inductor from the control point of view, and solving the problem of power shortage in the first half of the switching cycle, with good dynamic response performance. This method is applicable to a variety of phase shift control strategies such as single phase shift, extended phase shift, double phase shift, and triple phase shift.

[0040] Figure 6 and Figure 7 The inductor current waveforms are shown respectively when the load changes without DC bias optimization and with DC bias optimization.

[0041] The specific implementation methods are as follows:

[0042] Figure 5 The figure shows the implementation process of the proposed control method in the DAB converter.

[0043] First, sample the input voltage V in With output voltage V o , calculate the voltage conversion ratio of the converter d = N * V o / V in With reference power P b =N*V in *V o / (2*f s *L r ). Where N is the transformer ratio, f s is the switching frequency of the DAB converter, L r is the inductance of the auxiliary inductor. oref The output voltage sampling value V o The voltage error obtained by the difference is output as the output power reference value P through the voltage controller. oref , calculate the unit power as P n =P oref / P b .

[0044] The calculation of the primary and secondary duty ratios and phase shift angles D1, D2, and D3 is determined by the primary and secondary voltages of the converter and its processing power. Currently, a large number of researchers have optimized the calculation methods of D1, D2, and D3 to optimize the reactive circulating current of the DAB converter. The focus of the present invention is to introduce a new control variable D4 to eliminate the generation of the converter DC bias, rather than the calculation method of D1, D2, and D3. Therefore, the following analysis takes the calculation method of the primary and secondary duty ratios and phase shift angles D1, D2, and D3 given by the maximum current stress optimization modulation strategy proposed in the literature (S.Shao, M.Jiang, W.Ye, Y.Li, J.Zhang and K.Sheng, "Optimal Phase-Shift Control to Minimize Reactive Power for a Dual ActiveBridge DC-DC Converter," in IEEE Transactions on Power Electronics, vol.34, no.10, pp.10193-10205, Oct.2019, doi:10.1109 / TPEL.2018.2890292.) as an example. On the basis of obtaining D1, D2, and D3, the calculation method of the phase shift angle D4 of the driving waveform forward is analyzed.

[0045] According to d and P n The circuit can be divided into 8 working modes. When d<1 and 0≤P n ≤d*(1-d) / 2, the converter operates in mode 1f; when d<1 and d*(1-d) / 2 <P n ≤0.25, the converter operates in mode 2f; when d>1 and 0≤P n ≤(d-1) / 2 / d 2 When d>1 and (d-1) / 2 / d 2 <P n ≤0.25, the converter operates in mode 4f; when d<1 and -d*(1-d) / 2≤P n <0, the converter operates in mode 1r; when d<1 and -0.25≤P n <-d*(1-d) / 2, the converter operates in mode 2r; when d>1 and -(d-1) / 2 / d 2 ≤P n <0, the converter operates in mode 3r; when d>1 and -0.25≤P n <(d-1) / 2 / d 2When , the converter works in mode 4r. For the above eight working modes, the expressions for calculating D1, D2, and D3 in the sub-mode are as follows:

[0046]

[0047]

[0048]

[0049]

[0050]

[0051]

[0052]

[0053]

[0054] Based on D1, D2, D3, V in With V o The value of the new control variable D4 introduced in the embodiment of the present invention is calculated.

[0055] For mode 1f and mode 3f, the initial value and the final value in the switching cycle are both 0, and no optimization is required, that is, D4=0.

[0056] For mode 2f and mode 4f, the power transmission direction is forward. The drive control waveforms without DC bias optimization and with DC bias optimization are as follows: Figure 3 As shown. Based on the volt-ampere characteristics of the inductor, the expression of the inductor current changing with time can be obtained as:

[0057]

[0058] Among them, t1, t2, t3, t4 can be expressed as:

[0059]

[0060] From the volt-second balance characteristics of inductance, we can get:

[0061] i L4 =-i L0

[0062] Combining the above three equations, we can get the inductor current values ​​at each moment t1, t2, t3, and t4 respectively:

[0063]

[0064] In order to eliminate the DC bias, a new control variable D4 is introduced to adjust the drive signal so that the initial and final values ​​of the inductor current in a single switching cycle are both 0, thus:

[0065]

[0066] The calculation expression of D4 under mode 2f and mode 4f is solved as follows:

[0067]

[0068] For mode 1r and mode 3r, the driving signal and current waveform are symmetrical with mode 1f and mode 3f, and the rising edge of the secondary bridge driving signal leads the primary bridge driving signal by a phase shift angle of D3, so D4=D3 in this mode.

[0069] For mode 2r and mode 4r, the power transmission direction is reverse. The drive control waveforms without DC bias optimization and with DC bias optimization are as follows: Figure 4 As shown. The current at each segment point is solved by the volt-second balance characteristic of the inductor:

[0070]

[0071] In order to eliminate the DC bias, a new control variable D4 is introduced to adjust the drive signal so that the initial and final values ​​of the inductor current in a single switching cycle are both 0, thus:

[0072]

[0073] The calculation expression of D4 under mode 2r and mode 4r is obtained as follows:

[0074]

[0075] According to the calculated D4, the driving waveform generated by D1, D2, and D3 is corrected. The optimized driving waveform is as follows: Figure 3 (b) and Figure 4 This control method can make the initial value and final value of the inductor current in a single switching cycle both 0, realize the decoupling of the inductor current in adjacent switching cycles, and thus eliminate the generation of inductor bias current.

[0076] The core idea of ​​the control method proposed in the embodiment of the present invention is to introduce a new control variable D4 to adjust the drive signal so that the initial value and the final value of the inductor current in a single switching cycle are both 0. This method is applicable to multiple phase shift control strategies such as single phase shift, extended phase shift, double phase shift, and triple phase shift.

[0077] When the embodiment of the present invention generates a PWM signal based on the obtained values ​​of D1, D2, D3, and D4, it is not necessary to change the duty cycle of the primary and secondary full-bridge voltage waveforms and the phase shift angle of the rising edge of the primary and secondary full-bridge voltage waveforms. It is only necessary to perform an overall translation of the drive signal based on the calculated forward phase shift angle D4 of the drive waveform. When the control method is used to drive the DAB converter to work, the initial value and the final value of the auxiliary inductor current are both 0 in a single switching cycle, and the inductor current is completely symmetrical in the switching cycle, which solves the problem of power shortage in the first half of the switching cycle of the transient process.

[0078] Although the preferred embodiments of the present application have been described, those skilled in the art may make other changes and modifications to these embodiments once they have learned the basic creative concept. Therefore, the appended claims are intended to be interpreted as including the preferred embodiments and all changes and modifications falling within the scope of the present application.

[0079] Obviously, those skilled in the art can make various changes and modifications to the present application without departing from the spirit and scope of the present application. Thus, if these modifications and variations of the present application fall within the scope of the claims of the present application and their equivalents, the present application is also intended to include these modifications and variations.

Claims

1. A phase shift control method for a DAB converter without DC bias, characterized in that: The following steps are involved: S1, sampling DAB converter input voltage V in With output voltage V o ; S2, output reference voltage V oref With output voltage V o The voltage error is obtained by performing the difference, and the voltage error is used as the input of the voltage controller to obtain the output power reference value P oref ; S3. Define the current reference power P b =N*V in *V o / (2*f s *L r ), where N is the transformer ratio, f s is the switching frequency, L r is the auxiliary inductor value, and the per-unit power is defined as P n =P oref / P b , define the voltage conversion ratio d = N * V o / V in ; Define D1 and D2 as the duty cycle of the primary and secondary full-bridge voltage waveforms, D3 as the phase shift angle of the rising edge of the primary and secondary full-bridge voltage waveforms, and D4 as the phase shift angle of the forward shift of the drive waveform; S4, based on the standard power P n The value of the voltage conversion ratio d determines the working mode of the DAB converter, and calculates the values ​​of D1, D2, and D3 corresponding to the current mode. in 、V o And the phase shift angle D4 of the driving waveform forward is obtained by calculating the D1, D2, and D3 sub-modal modes; S5. Generate a PWM signal according to the obtained values ​​of D1, D2, D3, and D4, and use the PWM signal as a driving signal for the DAB converter.

2. The phase shift control method for a DAB converter without DC bias according to claim 1, characterized in that: The operating modes of the DAB converter include: when d < 1 and 0 ≤ P n ≤d*(1-d) / 2, the DAB converter operates in mode 1f; when d<1 and d*(1-d) / 2 <P n ≤0.25, the DAB converter operates in mode 2f; when d>1 and 0≤P n ≤(d-1) / 2 / d 2 When d>1 and (d-1) / 2 / d 2 <P n ≤0.25, the DAB converter operates in mode 4f; when d<1 and -d*(1-d) / 2≤P n <0, the DAB converter operates in mode 1r; when d<1 and -0.25≤P n <-d*(1-d) / 2, the DAB converter operates in mode 2r; when d>1 and -(d-1) / 2 / d 2 ≤P n <0, the DAB converter operates in mode 3r; when d>1 and -0.25≤P n <(d-1) / 2 / d 2 When , the DAB converter works in mode 4r; among them, mode 1f, mode 2f, mode 3f, and mode 4f work in the forward power transmission mode; mode 1r, mode 2r, mode 3r, and mode 4r work in the reverse power transmission mode.

3. The phase shift control method for a DAB converter without DC bias according to claim 2, characterized in that: The expressions of D1, D2, and D3 are as follows: Mode 1f: Mode 2f: Mode 3f: Mode 4f: Mode 1r: Mode 2r: Mode 3r: Mode 4r:

4. The phase shift control method for a DAB converter without DC bias according to claim 2 or 3, characterized in that: The calculation formula of the phase shift angle D4 of the driving waveform is:

5. The phase shift control method for a DAB converter without DC bias according to claim 4, characterized in that: According to the phase shift angle D4 of the driving waveform, the driving waveform generated by the duty ratios D1, D2, and D3 is corrected.

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