A resonance compensation control method for three-phase inverter

Abstract

The present invention belongs to the field of inverter technology, and specifically relates to a three-phase inverter resonance compensation control method, including a power module and a resonance compensation module, wherein the power module is the power conversion circuit, LC output filter and protection circuit of the three-phase inverter; the resonance compensation module is mainly a proportional resonant controller (PR controller), which is used to generate harmonic feedback compensation. The control architecture of the entire inverter is a dual closed-loop control, with voltage control as the outer loop, current control as the inner loop, and the PR controller as feedback control. The outer loop of the voltage adopts proportional and integral control, and the inner loop of the current uses only proportional control, which avoids the phase delay introduced by the PI of the inner loop and the PI of the outer loop. By adjusting the control parameters of the voltage control loop and the current control loop, multiple harmonics can be effectively suppressed.

Description

Technical Field

[0001] The present invention belongs to the technical field of inverters, and in particular relates to a resonance compensation control method for a three-phase inverter. Background Art

[0002] With the shortage of traditional energy sources and increasingly serious environmental concerns, distributed generation technologies such as photovoltaics, wind power, and fuel cells are rapidly developing. Microgrids, as systems integrating distributed generation, energy storage, and local loads, are becoming increasingly widespread. In isolated microgrid operation, nonlinear loads cause output current distortion, which in turn leads to output voltage distortion, reducing voltage output quality. According to national and IEEE standards, the total harmonic distortion (THD) of grid voltage should be controlled within 5%. When PCC (point of common connection) voltage harmonics in a microgrid are severe, they can disrupt the normal operation of electrical equipment within the microgrid. Therefore, measures must be taken to suppress PCC voltage harmonics caused by nonlinear loads to ensure the power quality of the microgrid and minimize grid interference when integrated into the grid.

[0003] The traditional control strategy often uses a voltage and current double closed loop plus a PR controller. The current inner loop uses the capacitor current i on the inverter side. C Feedback control often involves multiple capacitors or complex filtering circuits, requiring more sensors to monitor the capacitor currents at different nodes. The presence of multiple capacitors or current paths in the circuit requires individual current monitoring for each capacitor, resulting in a system with many sensors and high costs. When the PR controller is on the main line, it assumes the responsibility of leading the system's dynamic response, directly affecting the system's control output. In this case, the PR controller must process all frequency components in the system. However, because the PR controller responds slowly to frequency components beyond the fundamental frequency, its dynamic response becomes sluggish. Especially when faced with rapid changes or disturbances, the PR controller on the main line is in the outer voltage loop, which has a bandwidth approximately 1 / 10 of the inner current loop. This results in a slower response, slower adjustment of the system output, and weaker dynamic performance. Summary of the Invention

[0004] The present invention aims to provide a three-phase inverter resonance compensation control method to reduce the impact of harmonics on the stability of power grid operation, enhance system robustness and dynamic response characteristics, and reduce costs; adjust the control parameters of the voltage loop and current loop to effectively suppress harmonics and improve system stability.

[0005] In order to achieve the above object, the present invention provides a method for controlling resonance compensation of a three-phase inverter, comprising the following steps:

[0006] Step 1: Detect the inductor current i on the inverter side Labc , detect output voltage voabc , detect the LC filter output current i oabc ;

[0007] Step 2: The frequency-locked loop (FLL) controls the system frequency to the reference frequency and converts the inverter frequency into an angle using the formula ph = 2π × ω.

[0008] Step 3: Adjust the current loop control parameter K, where the transfer function of the current loop control is:

[0009]

[0010] Step 4: Adjust the voltage loop control parameters KP and KI, where the transfer function of the voltage loop control is:

[0011]

[0012] Step 5: Tune the PR controller parameter K h , the transfer function of the proportional resonant controller is:

[0013]

[0014] Step 6: v oabc The coordinate transformation is transformed from the abc coordinate system to the dq axis coordinate system, and then the control parameters of the voltage outer loop are optimized and the quasi-proportional resonant controller is used to eliminate the influence of low-order harmonics to obtain the inverter port reference current i ref abc ;

[0015] Step 7: Inverter port reference current i ref abc , inverter side inductor current i Labc 、Output voltage v oabc Transform from the abc coordinate system to the dq axis coordinate system, input it into the current loop, and then obtain the inverter port voltage u in the αβ coordinate system through coordinate transformation. αβ ;

[0016] Step 8: Inverter port voltage u αβ The SVPWM modulation method is used to output the PWM drive signal of the inverter switching device, driving the inverter switch to generate stable three-phase AC power.

[0017] The working principle and beneficial effect of this solution are: by detecting the inductor current i Labc 、Output voltage v oabc , LC filter output current i oabc , and then convert the abc coordinate system into the dq coordinate system to obtain the i required for control Ldq 、v odq After the voltage outer loop and proportional resonant controller, the inverter port current reference value i is obtained. ref abc, the current reference value and i Ldq 、v odq Input to the current inner loop controller, and the inverter port voltage u in the αβ coordinate system is obtained through the current inner loop controller αβ Finally, SVPWM modulation is used to output PWM drive signals for the inverter's switching devices. This PWM drive signal enables the inverter to output three-phase AC power. This reduces the impact of harmonics on grid stability, enhances system robustness and dynamic response, and reduces costs. Adjusting the control parameters of the voltage and current loops effectively suppresses harmonics and improves system stability.

[0018] Optionally, a filter inductor current sensor is used to detect the inductor current i Labc , use the filter capacitor voltage sensor to detect the output voltage v oabc , use the filter output current sensor to detect the LC filter output current i oabc .

[0019] Optionally, in step 1, the inductor current i on the inverter side is detected. Labc . Reduce the number of sensors.

[0020] Optionally, in step 4, a multi-resonant harmonic compensator is added, and the transfer function is:

[0021]

[0022] Harmonic voltage distortion is further reduced, the resonance peak is further effectively suppressed, and the attenuation effect of high-amplitude low-order harmonics is enhanced.

[0023] Optionally, in step 4, an FLL frequency-locked loop is added to keep the reference frequency ω of the inverter always at ω ref Ensure that the power output by the inverter matches the voltage and frequency of the grid and achieves synchronization with the grid voltage.

[0024] Optionally, in step 5, a proportional resonant controller is added to the voltage outer loop for feedback control.

[0025] Optionally, voltage feedforward is used in the voltage outer loop in step 6 to enhance the dynamic performance of the system. BRIEF DESCRIPTION OF THE DRAWINGS

[0026] Figure 1 Schematic diagram of a three-phase inverter resonance compensation control method according to an embodiment of the present invention;

[0027] Figure 2 This is a voltage inner loop control diagram in an embodiment of the present invention;

[0028] Figure 3 1 is a Bode diagram of the system under different current inner loop gains in an embodiment of the present invention;

[0029] Figure 4 This is a block diagram of the voltage outer loop control in an embodiment of the present invention;

[0030] Figure 5 This is a simplified block diagram of the voltage outer loop control in an embodiment of the present invention;

[0031] Figure 6 1 is the Bode diagram of the system in the embodiment of the present invention when K=15.4, KP=0.15 and loads Z=10, Z=50, and Z=100;

[0032] Figure 7 1 is the Bode diagram of the system in the embodiment of the present invention when K=15.4, KP=0.143 and loads Z=10, Z=50, and Z=100;

[0033] Figure 8 : The open-loop Bode diagram of the system in the embodiment of the present invention at KI of 1, 5, 15, 20, 30, and 40 respectively;

[0034] Figure 9 The Bode diagram of the system in the embodiment of the present invention at KP=0.01, 0.05, and 0.143;

[0035] Figure 10 This is a system control block diagram of a three-phase inverter resonance compensation control method with a multi-resonance harmonic compensator added in an embodiment of the present invention;

[0036] Figure 11 A Bode diagram of a system with a multi-resonant harmonic compensator added to an embodiment of the present invention;

[0037] Figure 12 The Bode diagram of the system in the embodiment of the present invention with the resonant harmonic compensator and the 50 Hz resonant compensation added;

[0038] Figure 13 This is a control block diagram of the entire system after adding quasi-proportional resonant control in an embodiment of the present invention;

[0039] Figure 14 1 is the open-loop Bode diagram of the system at loads Z=10, 50, and 10000 in an embodiment of the present invention;

[0040] Figure 15 : The closed-loop Bode diagram of the system at loads Z=10, 50, and 10000 in an embodiment of the present invention;

[0041] Figure 16 1 and 2 show the voltage and current waveforms and THD after the system control method is applied in an embodiment of the present invention. DETAILED DESCRIPTION

[0042] The following is further described in detail with reference to the accompanying drawings:

[0043] Example

[0044] A three-phase inverter resonance compensation control method, the principle diagram is as shown in the attached Figure 1 As shown, the following simulation experiments prove that the control parameters of the voltage and current dual closed-loop control system are adjusted while optimizing the refined control strategy. The specific derivation and result proof are explained as follows in conjunction with the accompanying drawings of the specification:

[0045] In this simulation verification experiment, the parameters are as follows:

[0046] f s =20KHz; L=500μH; r=0.2; ω=2π×50rad / s; C=22μF; V dc =300V.

[0047] Step 1: Detect the inductor current i on the inverter side through the sensor Labc , detect output voltage v oabc , detect the LC filter output current i oabc ;

[0048] Step 2: The frequency-locked loop (FLL) controls the system frequency to the reference frequency and converts the inverter frequency to an angle using the formula ph = 2π × ω.

[0049] Step 3: Adjust the current inner loop parameter K. The specific process is as follows:

[0050] like Figure 1 Current inner loop control block diagram, first adjust the current inner loop control parameter K, set the load impedance to Z, then Then the current loop open-loop transfer function is:

[0051]

[0052] The closed-loop transfer function can also be obtained:

[0053]

[0054] Assume the rated load Z = 10Ω, then:

[0055]

[0056] The current bandwidth is the switching frequency Pick The switching frequency is the bandwidth, that is: ω bi=2π×(0.2×20)KHz≈25Krad / s. The goal is to obtain a -3dB amplitude gain within the current bandwidth. According to the current loop gain formula:

[0057]

[0058] There are When G dB =20×log 10 G(jω bi )=20×log 10 (0.707)≈-3dB. The above formula shows that the system gain at the ω frequency is 0.707, which is -3dB. It ensures that the system response is attenuated to half of the original amplitude near the ω frequency, thereby reaching a balance point that can respond to frequency components without causing excessive oscillation of the system. Substituting all the calculated parameters into the current loop gain formula, we can obtain the gain K=15.4. Substituting the above derived parameters into the simulation model, we can obtain the Bode diagram of the system under different current inner loop gains as shown below: Figure 2 It can be guaranteed that under all loading conditions, the bandwidth will not be less than ω bi .

[0059] Step 4: Adjust the control parameters KP and KI of the voltage loop. The specific process is as follows:

[0060] like Figure 3 The control block diagram of the voltage outer loop is shown in Figure 2. According to the control block diagram, the control parameters KP and KI of the voltage loop are adjusted by the transfer function. Figure 3 It can be seen that the voltage outer loop includes the control of the current inner loop, so the control block diagram of the voltage control can be simplified to Figure 4 As shown, we have:

[0061]

[0062] The voltage control open-loop transfer function is:

[0063]

[0064] In the current loop. Let KI = 0;

[0065]

[0066] Given that K = 15.4, let KP = 0.15. Figure 5 The Bode plots for loads Z = 10, Z = 50, and Z = 100 are shown. It is intuitively clear from the plot that the inner current loop actively damps the LC resonance, and the system has sufficient phase margin. Even under light loads, the simplified transfer function, assuming Z → ∞, is as follows:

[0067]

[0068] but:

[0069]

[0070] Bandwidth selection is a compromise between transient response and anti-interference. The bandwidth of the voltage outer loop is generally designed to be greater than 10 times the fundamental frequency and less than 1 / 10 of the switching frequency. For the system involved in the simulation of this invention, it is greater than 500 Hz and less than 2 kHz. The simulation verification of this invention selects 1.3 kHz, then ω bv =2π×1.3kHZ≈8krad / s.

[0071] Let |G v (jω bv )| 2 =0.5, we can get KP=0.143.

[0072] Before simplification:

[0073]

[0074] Then there are:

[0075]

[0076] Where KP = 0.143, K = 15.4, KI = 0;

[0077] Draw as Figure 6 The Bode diagrams for loads Z=10, Z=50, and Z=100 shown in FIG. 1 show that the load has little effect on the system bandwidth and can be ignored, thereby ensuring the system bandwidth.

[0078] Applying the Routh-Hurwitz stability criterion to the voltage closed-loop transfer function G v (s) can obtain the stable boundary:

[0079]

[0080] Draw the open-loop Bode diagrams at KI of 10, 20, 30, and 40, as shown in the following example: Figure 7 As shown in the figure, a larger KI value eliminates steady-state errors; a smaller KI value ensures that the integral term does not affect other frequencies. Taking all factors into consideration, KI = 30 is selected. Therefore, the adjusted system control parameters are: K = 15.4, KP = 0.143, and KI = 30.

[0081] Step 5: Tune the PR controller parameter K h The specific process is as follows:

[0082] The same Figure 3 It can be seen that the control of the voltage outer loop can also write the transfer function:

[0083]

[0084] Draw out Z O (s) The Bode diagram at KP = 0.01, 0.05, 0.143 is as follows Figure 8 As shown in Figure 2, with the increase of KP, the harmonic voltage distortion is reduced, effectively suppressing the resonance peak. However, this control system is not sufficient to attenuate high-amplitude low-order harmonics, so a multi-harmonic compensator is required to achieve better control effects.

[0085] The principle of resonant harmonic compensation is: The control block diagram of adding multi-resonance harmonic compensator is as follows Figure 9 As shown, the transfer function based on this is:

[0086]

[0087] The transfer function and control parameters can be plotted as follows Figure 10 From the Bode diagram, we can see that the high-amplitude low-order harmonics are effectively suppressed, but the amplitude at the 50HZ fundamental wave is still high. On this basis, adding a resonance compensation at 50HZ can obtain Figure 11 Bode diagram of the voltage shown.

[0088] The added resonant compensator only affects the frequency near the resonance, so its impact on the dynamic performance of the inverter is negligible. In order to prevent the offset of the frequency w from causing the resonant compensator to be ineffective, the following methods can be used:

[0089]

[0090] By adding quasi-proportional resonant control, the control block diagram of the optimized entire system can be obtained as follows: Figure 12 As shown, the closed-loop transfer function of the entire system can be obtained:

[0091]

[0092] The open-loop transfer function can be obtained from the control block diagram:

[0093]

[0094] So the closed-loop transfer function is:

[0095]

[0096] The control parameters of the simulation verification experiment are K=15.4, KP=0.143, KI=30, Kh1 =2,K h5 =20,K h7 =10,K h11 =40,K h13 =50.

[0097] Step 6: v oabc The coordinate transformation is transformed from the abc coordinate system to the dq axis coordinate system, and then the control parameters of the voltage outer loop are optimized and the quasi-proportional resonant controller is used to eliminate the influence of low-order harmonics to obtain the inverter port reference current i ref abc ;

[0098] Step 7: Inverter port reference current i ref abc , inverter side inductor current i Labc , output voltage v oabc Transform from the abc coordinate system to the dq axis coordinate system, input it into the current loop, and then obtain the inverter port voltage u in the αβ coordinate system through coordinate transformation. αβ .

[0099] Step 8: Inverter port voltage u αβ The SVPWM modulation method is used to output the PWM drive signal of the inverter switch device, driving the inverter switch to generate stable three-phase AC power.

[0100] At load Z = 10, 50, 10000, the following is drawn: Figure 13 The system open loop Bode diagram shown in Figure 14 The closed-loop Bode diagram of the system is shown. From the Bode diagram, we can see that no matter how the load changes, it will not affect the performance and stability of the system. Figure 15 The voltage and current waveforms and system THD after applying the method of the present invention are shown. The dynamic response of the inverter output under different load conditions can be seen. The voltage waveform remains relatively stable under these different load conditions. The three-phase voltage waveform output by the inverter maintains a well-defined sinusoidal waveform throughout the test, demonstrating that the inverter has strong voltage stability. Frequency domain analysis shows that the inverter has low harmonic distortion, with a THD of 1.01%, far below the 5% specified by national and IEEE standards. This is a relatively ideal level, indicating the high output quality of the inverter.

[0101] The above are only embodiments of the present invention. The invention is not limited to the fields involved in this implementation case. Common knowledge such as the known specific structures and characteristics in the scheme is not described in detail here. Ordinary technicians in the relevant field are aware of all common technical knowledge in the technical field to which the invention belongs before the application date or priority date, can obtain all existing technologies in the field, and have the ability to apply conventional experimental means before that date. Ordinary technicians in the relevant field can improve and implement this scheme in combination with their own abilities under the inspiration given by this application. Some typical known structures or known methods should not become obstacles for ordinary technicians in the relevant field to implement this application. It should be pointed out that for those skilled in the art, without departing from the concept of the present invention, several variations and improvements can be made, which should also be regarded as the scope of protection of the present invention. These will not affect the effect of the implementation of the present invention and the practicality of the patent. The scope of protection required by this application shall be based on the content of its claims, and the specific implementation methods and other records in the specification can be used to interpret the content of the claims.

Claims

1. A three-phase inverter resonance compensation control method, characterized in that: The following steps are involved: Step 1: Detect the inductor current i Labc , detect output voltage v oabc , detect the LC filter output current i oabc ; Step 2: The frequency-locked loop (FLL) controls the system frequency to the reference frequency and converts the inverter frequency into an angle using the formula ph = 2π × ω. Step 3: Adjust the current loop control parameter K, where the transfer function of the current loop control is: Step 4: Adjust the voltage loop control parameters KP and KI, where the transfer function of the voltage loop control is: Adding a multi-resonant harmonic compensator, the transfer function is: Step 5: Tune the PR controller parameter K h , the transfer function of the proportional resonant controller is: Step 6: v oabc The coordinate transformation is transformed from the abc coordinate system to the dq axis coordinate system, and then the control parameters of the voltage outer loop are optimized and the quasi-proportional resonant controller is used to eliminate the influence of low-order harmonics to obtain the inverter port reference current i ref abc ; Step 7: Inverter port reference current i ref abc , inverter side inductor current i Labc , output voltage v oabc Transform from the abc coordinate system to the dq axis coordinate system, input it into the current loop, and then obtain the inverter port voltage u in the αβ coordinate system through coordinate transformation. αβ ; Step 8: Inverter port voltage u αβ The SVPWM modulation method is used to output the PWM drive signal of the inverter switching device, driving the inverter switch to generate stable three-phase AC power.

2. The three-phase inverter resonance compensation control method according to claim 1, characterized in that: Use the filter inductor current sensor to detect the inductor current i Labc , use the filter capacitor voltage sensor to detect the output voltage v oabc , use the filter output current sensor to detect the LC filter output current i oabc .

3. The three-phase inverter resonance compensation control method according to claim 1, characterized in that: In step 1, the inductor current i on the inverter side is detected. Labc .

4. The three-phase inverter resonance compensation control method according to claim 1, characterized in that: In step 2, an FLL frequency-locked loop is added to keep the inverter reference frequency ω at ω. ref .

5. The three-phase inverter resonance compensation control method according to claim 1, characterized in that: In step 5, a proportional resonant controller is added to the voltage outer loop.

6. The three-phase inverter resonance compensation control method according to claim 1, characterized in that: The voltage outer loop in step 6 adopts voltage feedforward.

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