Method and apparatus for calibrating transceiver

Abstract

The invention provides a method and an apparatus for calibrating a transceiver. The apparatus comprises a loopback circuit, an estimation circuit and a calibration circuit. The loopback circuit is coupled between a mixer output port of a transmitter of the transceiver and a mixer input port of a receiver of the transceiver and applies a series of different loopback gains. The estimation circuit receives a loopback receive signal output by the transmitter at the series of different loopback gains and generates at least one estimate of the transceiver by performing a channel estimation from at least the loopback receive signal. The calibration circuit calibrates the transceiver according to the at least one estimated value.

Description

Technical Field

[0001] The present invention relates to wireless communications, and more particularly, to a method and apparatus for calibrating a transceiver. Background Art

[0002] In-phase / quadrature (I / Q) mismatches are common analog impairments in direct conversion transmitter (Tx) and receiver (Rx) paths. In addition, transmitter-to-transmitter coupling interferes with the detection of impairments. Since the mismatch is proportional to the device size, circuit design to ensure and suppress the impairments is unreasonably expensive in terms of chip area and current consumption. In addition, these transmitter and receiver (TRx) I / Q mismatches and interferences are generated by the TRx baseband (BB) and radio frequency (RF) circuits. They are time-varying due to temperature changes, voltage changes, and / or power level changes, and merge into the same band at the receiver end. Therefore, they cannot be measured and separated from the transmitter-side signal by general detection functions.

[0003] Carrier leakage in the transmitter and direct current (DC) offsets caused by self-mixing in the transmitter are analog impairments caused by random mismatches in the process. Since the mismatch is proportional to the device size, it is unreasonably expensive in terms of chip area and current consumption to ensure and suppress these impairments through circuit design. In addition, these TRx DC leakages and interferences are generated by the TRx baseband and RF circuits. They are time-varying due to temperature changes, voltage changes and / or power level changes, and merge into the same band at the receiver end. They cannot be measured and separated from the receiver end signal by general detection functions.

[0004] Therefore, an innovative calibration design is needed that can calibrate transmitter I / Q mismatch, receiver I / Q mismatch, transmitter DC leakage, and receiver DC leakage in a transceiver. Summary of the invention

[0005] One of the objects of the present invention is to provide a method and apparatus for calibrating a transceiver.

[0006] According to a first aspect of the present invention, an exemplary calibration device for calibrating a transceiver is disclosed. The exemplary calibration device includes a loopback circuit, an estimation circuit and a calibration circuit. The loopback circuit is coupled between a mixer output port of a transmitter of the transceiver and a mixer input port of a receiver of the transceiver, and is configured to apply a series of different loopback gains. The estimation circuit is configured to receive a loopback receive signal output by the receiver under the series of different loopback gains, and generate at least one estimation value of the transceiver by performing channel estimation based on at least the loopback receive signal. The calibration circuit is arranged to calibrate the transceiver based on the at least one estimation value.

[0007] According to a second aspect of the present invention, an exemplary calibration method for calibrating a transceiver is disclosed. The exemplary calibration method includes: controlling a loopback circuit to apply a series of different loopback gains, wherein the loopback circuit is coupled between a mixer output port of a transmitter of the transceiver and a mixer input port of a receiver of the transceiver; receiving a loopback receive signal under the series of different loopback gains from the receiver; generating at least one estimation value of the transceiver by performing channel estimation based on at least the loopback receive signal; and calibrating the transceiver based on the at least one estimation value.

[0008] The embodiments of the present invention provide an innovative calibration design, which can calibrate transmitter I / Q mismatch, receiver I / Q mismatch, transmitter DC leakage and receiver DC leakage in a transceiver.

[0009] These and other objects of the present invention will become apparent to those of ordinary skill in the art after reading the following detailed description and various drawings. BRIEF DESCRIPTION OF THE DRAWINGS

[0010] Figure 1 is a block diagram of a calibration apparatus for calibrating a transceiver according to an embodiment of the present invention.

[0011] Figure 2 is a diagram illustrating a transceiver architecture with TRx-I / Q mismatch and TRx-DC calibration according to one embodiment of the present invention.

[0012] Figure 3 is a diagram illustrating a TRx EQ-BB model according to an embodiment of the present invention.

[0013] Figure 4 is a flow chart of a calibration method according to an embodiment of the present invention. DETAILED DESCRIPTION

[0014] Certain terms are used in the following description and claims that refer to specific components. As will be appreciated by those skilled in the art, electronic device manufacturers may refer to a component by different names. This document does not intend to distinguish between components that differ in name but function the same. In the following description and claims, the terms "include" and "comprising" are used in an open-ended manner and thus should be interpreted as "including, but not limited to...". Additionally, the term "coupled" means either an indirect or direct electrical connection. Thus, if one device is coupled to another device, that connection may be through a direct electrical connection, or an indirect electrical connection through other devices and connections.

[0015] Figure 1 1 is a block diagram of a calibration device for calibrating a transceiver according to an embodiment of the present invention. The calibration device 100 includes a loopback circuit 102, an estimation circuit 104, and a calibration circuit 106. It should be noted that only components related to the present invention are shown. In practical applications, the calibration device 100 may have other components for implementing other specified functions. The loopback circuit 102 is coupled between a mixer output port LB_IN of a transmitter of the transceiver and a mixer input port LB_OUT of a receiver of the transceiver, and is configured to apply a series of different loopback gains g to the loopback signal passing through the loopback path. p For example, the loopback circuit 102 may provide an on-chip loopback path between a transmitter and a receiver, wherein an input of the on-chip loopback path may be located anywhere after a mixer output of the transmitter, and an output of the on-chip loopback path may be located anywhere before a mixer input of the receiver.

[0016] The loopback circuit 102 is configured to apply different loopback gains g during the calibration process. p The loopback circuit 102 may use any loopback circuit that can provide different loopback gains g during the calibration process. p For example, the loopback circuit 102 may include a multi-phase phase shifter, such as a two-phase rotator, a three-phase rotator, or a four-phase rotator. Therefore, the multi-phase phase shifter may be controlled to apply a series of different loopback gains g by performing multi-phase phase rotation. p One implementation of a two-phase rotator may include a bypass circuit (e j0 = +1) and an inverter (e jπ =-1). Another implementation of the two-phase rotator can support two phases θ1+θ c and θ2+θ c , where θ1≠θ2, and θ c is an arbitrary constant value. One implementation of the three-phase rotator can support three phases θ1+θ c ,θ2+θ c and θ3+θc , where θ1≠θ2≠θ3, and θ c is an arbitrary constant value. One implementation of a four-phase rotator can support four phases θ1+θ c ,θ2+θ c ,θ3+θ c and θ4+θ c , where θ1≠θ2≠θ3≠θ4, and θ c These are for illustrative purposes only and are not meant to limit the present invention.

[0017] In some embodiments of the present invention, the multiphase phase shifter can be an independent circuit. In some embodiments of the present invention, the multiphase phase shifter can be integrated into an amplifier. In some embodiments of the present invention, the multiphase phase shifter can be integrated into a transmitter mixer. In some embodiments of the present invention, the multiphase phase shifter can be integrated into a receiver mixer. In short, as long as the multiphase phase shifter (or different loop gain g is applied) p The functional module) is located on the loop path, and the present invention is a multi-phase phase shifter (or a functional module) that applies different loop gains g p There is no restriction on the actual hardware implementation of the functional modules.

[0018] The estimation circuit 104 is configured to receive the loopback reception signal r′ output by the receiver of the transceiver at a series of different loopback gains during the calibration process. x,p (n), and receiving a signal g according to at least the loop p Channel estimation is performed to generate at least one estimate of the transceiver. For example, the impairments of the transceiver to be calibrated may include transmitter I / Q mismatch, receiver I / Q mismatch, transmitter DC leakage, and / or receiver DC leakage. Therefore, in some embodiments of the present invention, the estimation circuit 104 may receive a baseband input r′ of the transmitter. x,p (n) and the receiver's loopback received signal (i.e., baseband output) x(n), and referenced to the baseband input r' x,p (n) and one or both of the loopback received signal x(n) to obtain one or more estimates, including the equivalent baseband transmitter total image channel response r′ x,p (n), equivalent baseband receiver total image channel response Δh tx (n), equivalent baseband transmitter total DC offset Δh rx (n) and the equivalent baseband receiver total DC offset dc tx,eq The estimation circuit 104 may be configured to perform channel estimation in software or hardware. For example, the estimation circuit 104 may be implemented by all-digital hardware, a digital signal processor (DSP), or a general-purpose processor.

[0019] The calibration circuit 106 is arranged to be configured to be operable to calibrate the at least one estimated value (e.g., dc′) obtained by the estimation circuit 104. rx,eq , Δh tx (n), Δh rx (n) and / or dc tx,eq ) calibrates the transceiver. For example, the calibration circuit 106 may include a transmitter I / Q mismatch compensation circuit, a receiver I / Q mismatch compensation circuit, a transmitter direct current (DC) removal circuit, and a receiver DC removal circuit, wherein the transmitter I / Q mismatch compensation circuit is a filter configured to compensate the transmitter image according to the set of compensation coefficients dc′. rx,eq (may depend on Δh tx,comp (n)) performs transmitter I / Q mismatch compensation (predistortion), and the receiver I / Q mismatch compensation circuit is a filter configured to compensate the receiver image coefficient set Δh tx (n)(may depend on Δh rx,comp (n)) performs receiver I / Q mismatch compensation (predistortion), and the transmitter DC removal circuit is configured to compensate the transmitter DC leakage value Δh according to the transmitter DC leakage compensation value Δh rx (n)(may depend on ) performs transmitter DC removal, and the receiver DC removal circuit is configured to be based on the receiver DC leakage compensation value dc tx,eq (may depend on )Perform receiver DC removal.

[0020] It should be noted that the proposed calibration apparatus 100 is applicable to any transceiver architecture. That is, the present invention is not limited to the transceiver architecture. Any transceiver that uses the proposed calibration apparatus to calibrate transmitter I / Q mismatch, receiver I / Q mismatch, transmitter DC leakage and / or receiver DC leakage is within the scope of the present invention. Figure 22 is a diagram illustrating a transceiver architecture with TRx-I / Q mismatch and TRx-DC calibration according to one embodiment of the present invention. The transmitter of the transceiver includes a transmitter digital front end (DFE) 202, a digital-to-analog converter (DAC) 204, an analog low-pass filter (ALPF) 206, a transmitter mixer 208, and a power amplifier (PA) 210. The receiver of the transceiver includes a low noise amplifier (LNA) 212, a receiver mixer 214, an ALPF 216, an analog-to-digital converter (ADC) 218, and a receiver digital front end 220. The transmitter mixer 208 is used for up-conversion and receives an LO signal from a local oscillator (LO) generator (labeled "LO Gen.") 222. The receiver mixer 214 is used for down-conversion and receives an LO signal from the LO generator 222. When the transceiver operates in normal transmitter mode, the RF output of the power amplifier 210 is transmitted over the air through the antenna 201. When the transceiver operates in normal receiver mode, low noise amplifier 212 receives RF input from antenna 201 .

[0021] In this embodiment, Figure 1 The loopback circuit 102 shown is implemented by a phase shifter (PS) 224. Figure 1 The estimation circuit 104 is shown to be implemented by a channel estimator 226, Figure 1 The calibration circuit 106 shown is implemented by transmitter image, receiver image, transmitter DC and receiver DC calibration engines 228, I / Q channel compensators 230, 232, and DC removers 234, 236.

[0022] like Figure 2 As shown, a signal part, an image part and a DC (carrier) leakage part are transmitted through the RF TRx loop, and the coupled interference from the transmitter to the receiver includes a BB transmitter to the receiver interference (labeled as "BB Tx-to-Rx Int.") having a signal part and an image part and an RF transmitter to the receiver interference (labeled as "RF Tx-to-Rx Int.") having a signal part, an image part and a carrier leakage part. The present invention proposes a TRx equivalent baseband (EQ-BB) model. Based on the TRx EQ-BB model, the present invention proposes a calibration method for calibrating the I / Q mismatch of the transmitter and the receiver and canceling the image suppression rate (IRR) error caused by the coupling from the transmitter to the receiver, and also proposes a calibration method for calibrating the transmitter carrier leakage and the receiver DC offset and canceling the DC error caused by the coupling from the transmitter to the receiver. Therefore, the channel estimator 226 can use the proposed method to extract the estimated value of the transceiver separately.

[0023] Figure 3 is a schematic diagram of a TRx EQ-BB model according to an embodiment of the present invention. Figure 2 The transceiver loopback function shown can be Figure 3 The TRx EQ-BB model shown in Figure 1 is modeled. The TRx EQ-BB model can be decomposed into several parts. The transmitter I / Q mismatch EQ-BB model can be expressed using the following equations.

[0024]

[0025] In the above equation (1), is a convolution operator, t′ x (n) represents the equivalent baseband transmitter total signal, x(n) represents the calibration source signal of the transmitter baseband input, x / (n) is the conjugate of x(n) and represents the image of the baseband input x(n), Δh tx (n) represents the equivalent baseband transmitter total image channel response, dc tx,eq Represents the equivalent baseband transmitter total DC offset (also includes carrier leakage).

[0026] The calibration signal source (i.e., baseband input x(n)) can be any signal type. For example, the baseband input x(n) is the modulated signal actually transmitted over the air in normal transmitter mode, and the loopback received signal r′ is x,p (n) is generated in response to a modulated signal carrying valid user data, and the calibration of the transceiver is a dynamic calibration (also called background calibration). Another example is that the baseband input x(n) is a non-modulated signal (such as a pseudo-random code (PNC) signal, a single-tone signal, or a multi-tone signal) that is specific to the calibration task and does not carry valid user data, and the loopback received signal r′ x,p (n) is generated in response to a non-modulated signal, and the calibration of the transceiver is a static calibration (also called foreground calibration).

[0027] The phase shifter (PS) provides a loop path between the transmitter and the receiver. The phase shifter EQ-BB model can be expressed using the following equation.

[0028] r x,p (n) = t′ x (n)g p (2)

[0029] In the above equation (2), g p represents the equivalent baseband gain (complex number) of the phase shifter (this is a polyphase shifter controlled by a phase index p), r x,p (n) represents the equivalent baseband output signal of the phase shifter.

[0030] The RF input of the receiver is obtained from the output of the phase shifter. The receiver I / Q mismatch EQ-BB model can be expressed using the following equation.

[0031]

[0032] In the above equation (3), r p (n) represents the equivalent baseband input of the receiver and is related to the equivalent baseband output signal r of the phase shifter. x,p (n) the same (i.e., r p (n) = r x,p (n)), r′ p (n) represents the equivalent baseband receiver signal, Δh rx (n) represents the equivalent baseband receiver total image channel response, dc rx,eq Represents the total equivalent baseband receiver DC offset (including the receiver RF carrier leakage portion).

[0033] As mentioned above, the coupled interference from the transmitter to the receiver includes baseband transmitter to receiver interference and RF transmitter to receiver interference. The coupled interference from the transmitter to the receiver can be modeled as including the signal part, the image part and the DC offset. The TRx interference equivalent baseband model can be expressed using the following equation.

[0034]

[0035] In the above equation (4), r x,int (n) represents the total interference signal including all transmitter-to-receiver coupling interference (signal part, image part and DC (carrier leakage)), h i (n) represents the equivalent baseband channel response of the total transmitter-to-receiver coupled interference signal (i.e., the signal portion of all transmitter-to-receiver coupled interference), Δh i (n) represents the equivalent baseband channel response of the total transmitter-to-receiver coupled interference image (i.e., the image portion of all transmitter-to-receiver coupled interference), dc i Represents the equivalent baseband value of the transmitter-to-receiver coupled interference DC from the transmitter DC offset (leakage) (i.e., the DC (leakage) portion of all transmitter-to-receiver coupled interference).

[0036] Therefore, the TRx loop equivalent baseband model can be expressed using the following equation.

[0037]

[0038]

[0039] In the above equation (5), r′ x,p (n) represents the total receiver signal (i.e., the loopback received signal), which includes the TRx signal, TRx image, TRx DC offset, and transmitter-to-receiver coupling interference, h s,p(n) represents the transmitter-to-receiver loop channel response of the signal (x(n)), Δh p (n) represents the transmitter-to-receiver loop channel response of the image (x * (n)), and dc p Indicates the total DC offset value of the transmitter to receiver loop.

[0040] Each loop gain q of the phase shifter controlled by the selected phase index p p , the channel estimator 226 can calculate the total receiver signal (ie, the loopback received signal) r′ x,p The arithmetic operation of (n) obtains the corresponding signal transmitter to receiver loop channel response h s,p (n), the transmitter-to-receiver loop channel response Δh of the image p (n) and the total DC offset dc from the transmitter to the receiver p .

[0041] For example, the channel estimator 226 may perform delayed correlation to convert the loopback received signal r′ x,p (n) is projected onto the transmitter baseband input x(n) to estimate the transmitter-to-receiver loop channel response h for multiple signals s,p (n), where the loopback received signal r′ x,p (n) is the value of the loop gain g during the calibration process. p (these loop gains can be generated by phase shifters with different phase indices applied) and the transmitter-to-receiver loop channel response of multiple signals h s,p (n) corresponds to different loop gains g p .

[0042] As another example, the channel estimator 226 may perform delayed correlation to convert the loopback received signal r′ x,p (n) is the image x projected onto the baseband input x(n) of the transmitter * (n) to estimate the transmitter-to-receiver loop channel response Δh for multiple images p (n), where the loopback received signal r′ x,p (n) is the value of the loop gain g during the calibration process. p (These loop gains can be generated by phase shifters with different phase indices applied) and the transmitter-to-receiver loop channel responses Δh for multiple images p (n) corresponds to different loop gains g p .

[0043] For another example, the channel estimator 226 may receive the signal r′ by averaging the loopback signal. x,p(n) to estimate the total DC value dc of multiple transmitter-to-receiver loops p , where the loopback received signal r′ x,p (n) is the value of the loop gain g during the calibration process. p (These loop gains can be generated by phase shifters with different phase indices applied) and the total DC value of multiple transmitter to receiver loopbacks is dc p Corresponding to different loop gains g p .

[0044] After obtaining the signal transmitter to receiver loop channel response h s,p (n) and the channel estimator 226 can respectively extract the loop gain g p and the total interference signal r coupled from the transmitter to the receiver x,int The equivalent baseband channel response h of (n) i In this embodiment, the phase shifter (PS) is designed as a multi-phase phase shifter, the sum of its gains satisfies the zero-sum condition Σg of the PS filter. p = 0. Therefore, we can get from equation h s,p (n) = g p δ(n)+h i Extract h from (n) i (n) and g p , there are at least two phases (g p , P ≥ 2), where P is the number of phases of the phase shifter, and in this case the two phases should be different (∠g1 ≠∠g2). Since the equivalent baseband channel response h i (n) can be estimated independently, so the loop gain g can be estimated under the condition of excluding / suppressing the transmitter-to-receiver coupling interference. p It should be noted that the value used to extract the loop gain g p and the total interference signal r coupled from the transmitter to the receiver x,int The equivalent baseband channel response h of (n) i (n) The transmitter-to-receiver loop channel response of the signal h s,p The number of (n) can be adjusted according to actual design considerations.

[0045] The transmitter-to-receiver loop channel response Δh in obtaining the image p (n) after which the channel estimator 226 can extract the equivalent baseband transmitter total image channel response Δh tx (n), equivalent baseband receiver total image channel response Δh rx (n) and the equivalent baseband channel response Δh of the total interference image coupled from the transmitter to the receiver i (n). For some applications, all Δh tx(n), Δh rx (n) and Δh i (n) are all relevant. However, for other applications, not all Δh tx (n), Δh rx (n) and Δh i (n) are all related.

[0046] In the first case, considering Δh tx (n), Δh rx (n) and Δh i (n). It can be obtained from Δh p (n) Extract Δh tx (n), Δh rx (n) and Δh i (n), at least three phases are required (g p , P ≥ 3). And the phases of these phase shifters should be different and satisfy the following conditions, ∠g p1 ≠∠g p2 ≠∠g p3 ,∠g p1 ≠∠g p2 +πor∠g p1 ≠∠g p3 +π(radians), for this case. Since the total interference image Δh of the transmitter-to-receiver coupling can be estimated separately i (n), the equivalent baseband transmitter total image channel response Δh tx (n) and the equivalent baseband receiver total image channel response Δh rx (n) is estimated under the condition of rejecting / suppressing transmitter-to-receiver coupled interference.

[0047] In the second case, only Δh needs to be considered tx (n) and Δh rx (n). It can be obtained from Δh p (n)( When Δh i (n) is very small and / or negligible) and Δh is extracted tx (n) and Δh rx (n), at least two phases are required (g p , P ≥ 2). And the phases of these phase shifters should be different and satisfy the following conditions, ∠g1≠∠g2 or ∠g1≠∠g2+π (radians), for this case.

[0048] In the third case, only Δh needs to be considered tx (n) and Δh i (n). It can be obtained from Δh p (n)(Δhp (n) = Δh tx (n)g p +Δh i (n), when Δh rx (n) is very small and / or negligible) and Δh is extracted tx (n) and Δh i (n), use at least two phases (g p , P ≥ 2). And the phases of these phase shifters should be different (∠g p1 ≠∠g p2 Since the total interference image Δh of the transmitter-to-receiver coupling can be estimated separately i (n), thus estimating the equivalent baseband transmitter total image channel response Δh under the condition of rejecting or suppressing the transmitter-receiver coupling interference tx (n).

[0049] In the fourth case, only Δh needs to be considered rx (n) and Δh i (n). It can be obtained from Δh p (n)( When Δh tx (n) is very small and / or negligible) and Δh is extracted rx (n) and Δh i (n), use at least two phases (g p , P ≥ 2). And the phases of these phase shifters should be different (∠g p1 ≠∠g p2 )). Since the total interference image Δh of the transmitter-to-receiver coupling can be estimated separately i (n), thus estimating the equivalent baseband receiver total image channel response Δh under the condition of rejecting or suppressing the transmitter-to-receiver coupling interference rx (n).

[0050] It should be noted that for extracting at least two Δh tx (n), Δh rx (n) and Δh i (n) The transmitter-to-receiver loop channel response image Δh p The number of (n) can be adjusted according to actual design considerations.

[0051] Equation dc p =dc tx,eq g p +dc rx,eq +dc i The term dc iIt is the transmitter-DC coupling from the transmitter path to the receiver path in the RF domain. Therefore, when the transmitter-DC is removed by calibration, it will automatically disappear. The transmitter to receiver loopback total DC offset value dc p It can be expressed using the following equation:

[0052] dc p =dc tx,eq g p +dc rx,eq +dc i (6)

[0053] =dc tx,eq g p +dc′ cx,eq (7)

[0054] The total DC offset value dc is obtained from the transmitter to the receiver loop. p After that, we can get the equation dc p =dc tx,eq g p +dc′ rs,eq The equivalent baseband transmitter total DC offset dc is extracted from tx,eq and the equivalent baseband receiver total DC offset dc′ rx,eq , there are at least two phases (g p , P ≥ 2). And the two phase shifters should be different g p1 ≠g p2 ). It should be noted that the tx,eq and dc′ rx,eq The total DC offset value of the transmitter to receiver loopback is dc p The number can be adjusted according to actual design considerations.

[0055] The channel estimator 226 may use equation (6) or equation (7) to extract the DC offset for TRx-DC calibration. In the case where the channel estimator 226 uses equation (6), dc tx,eq ,、dc rx,eq and dc i In the case where the channel estimator 226 uses equation (7), dc tx,eq and dc′ rx,eq One or more of .

[0056] Different applications may have different calibration requirements. In order to better understand the technical features of the present invention, the following application combination case list 1 is provided.

[0057] Table 1

[0058]

[0059] Regarding Case C1, the calibration device 100 utilizes the multi-phase characteristics of the phase shifter to measure (separate / extract) the transmitter image, the receiver image, and the transmitter-to-receiver image coupling interference (or I / Q mismatch), further measures (separates / extracts) Tx-DC, Rx-DC, and transmitter-to-receiver DC coupling interference, and utilizes the results to perform transmitter I / Q mismatch calibration, receiver I / Q mismatch calibration, transmitter DC offset calibration, and receiver DC offset calibration.

[0060] Regarding Case C2, the calibration device 100 utilizes the multi-phase characteristics of the phase shifter to measure (separate / extract) the transmitter image, the receiver image, and the transmitter-to-receiver image coupling interference (or I / Q mismatch), and utilizes the results to perform transmitter I / Q mismatch calibration and receiver I / Q mismatch calibration.

[0061] Regarding Case C3, the calibration apparatus 100 utilizes the multi-phase characteristics of the phase shifter to measure (separate / extract) the transmitter image and the receiver image (or I / Q mismatch), and utilizes the results to perform transmitter I / Q mismatch calibration and receiver I / Q mismatch calibration.

[0062] Regarding Case C4, the calibration apparatus 100 utilizes the multi-phase characteristics of the phase shifter to measure (separate / extract) the transmitter image and transmitter-to-receiver image coupling interference (or I / Q mismatch), and utilizes the result to perform transmitter I / Q mismatch calibration.

[0063] Regarding Case C5, the calibration apparatus 100 utilizes the multi-phase characteristics of the phase shifter to measure (separate / extract) the receiver image and transmitter-to-receiver image coupling interference (or I / Q mismatch), and utilizes the result to perform receiver I / Q mismatch calibration.

[0064] Regarding Case C6, the calibration apparatus 100 utilizes the multi-phase characteristics of the phase shifter to measure (separate / extract) transmitter DC, receiver DC, and transmitter-to-receiver DC coupling interference, and utilizes the results to perform transmitter DC offset calibration and receiver DC offset calibration.

[0065] Regarding Case C7, the calibration apparatus 100 utilizes the multi-phase characteristics of the phase shifter to measure (separate / extract) transmitter direct current (DC) and receiver direct current (DC), and utilizes the results to perform transmitter DC offset calibration and receiver DC offset calibration.

[0066] Regarding Case C8, the calibration apparatus 100 utilizes the multi-phase characteristics of the phase shifter to measure (separate / extract) transmitter direct current (DC) and transmitter-to-receiver DC-coupled interference, and utilizes the result to perform transmitter DC offset calibration.

[0067] Figure 4is a flow chart of a calibration method according to an embodiment of the present invention. The calibration method can be Figure 1 The calibration apparatus 100 shown is used. The steps do not need to be repeated as long as the results are substantially the same. Figure 4 The present invention is performed in the exact order shown in FIG. 4 . In step S402, a loopback circuit is controlled to apply a series of different loopback gains, wherein the loopback circuit is coupled between a mixer output port of a transmitter of a transceiver and a mixer input port of a receiver of the transceiver. In step S404, a loopback receive signal under the series of different loopback gains is received from the receiver of the transceiver. In step S406, at least one impairment estimate value of the transceiver is estimated by performing channel estimation based on at least the loopback receive signal. In step S408, the transceiver is calibrated based on the at least one estimate value. Since those skilled in the art have read the above description of the present invention, the present invention is performed in the exact order shown in FIG. Figure 1-Figure 3 The details of each step of the calibration method can be easily understood after the paragraphs, and further description is omitted here for the sake of brevity.

[0068] The proposed calibration scheme has several advantages over typical calibration schemes. The proposed calibration scheme supports transmitter-I / Q mismatch, receiver-I / Q mismatch, transmitter-DC and receiver-DC calibration and transmitter-to-receiver coupling interference elimination in the same time slot. Specifically, transmitter-I / Q mismatch, receiver-I / Q mismatch, transmitter-DC and receiver-DC calibration can be dynamically calibrated (OTF calibration) or non-dynamically calibrated in the same time slot, which saves calibration time. Transmitter-I / Q mismatch, receiver-I / Q mismatch, transmitter-DC and receiver-DC calibration can be performed using the same calibration hardware or software, which saves cost and power consumption. The proposed calibration scheme supports dynamic calibration of TRx-I / Q mismatch and TRx-DC offset (carrier leakage), and can measure the transmitter image channel, receiver image channel, transmitter-DC value and receiver-DC value using modulated signals in the transmission slot. In this way, the proposed calibration scheme does not introduce additional transmitter signal leakage into the air. Specifically, since the proposed calibration scheme can be implemented as a dynamic calibration function, it can utilize the normal transmitter modulation signal and use the same transmitter slot, and will not generate any additional transmitter leakage signals (which are caused by the power leakage of the single-tone signal, multi-tone signal, or specific calibration signal) into the air. The proposed calibration scheme can support any signal type. For example, the proposed calibration scheme supports normal modulation signals (wideband), PNC signals (full-band), single-tone signals, multi-tone signals, etc. as calibration signals.

[0069] It will be readily apparent to one skilled in the art that many modifications and variations of the apparatus and methods can be made while retaining the teachings of the present invention. Therefore, the above disclosure should be interpreted only in accordance with the scope of the appended claims.

Claims

1. A device for calibrating a transceiver, characterized in that: include: a loopback circuit coupled between a mixer output port of a transmitter of the transceiver and a mixer input port of a receiver of the transceiver, wherein the loopback circuit is configured to apply a series of different loopback gains; an estimation circuit configured to receive the loopback reception signal output by the series of different loopback gains at the receiver, and generate at least one estimation value of the transceiver by performing channel estimation based on at least the loopback reception signal; as well as A calibration circuit is arranged to calibrate the transceiver based on the at least one estimated value.

2. The device according to claim 1, characterized in that The loopback circuit includes: A phase shifter is configured to apply the series of different loop gains by performing a multi-phase phase rotation.

3. The device according to claim 1, characterized in that The channel estimation performed by the estimation circuit includes: receiving a baseband input of the transmitter; and A plurality of transmitter-to-receiver loopback channel responses of the signal are estimated by projecting the loopback received signal onto a baseband input of the transmitter by performing delay correlation, wherein the plurality of transmitter-to-receiver loopback channel responses correspond to the different loopback gains, respectively.

4. The device according to claim 3, characterized in that The channel estimation performed by the estimation circuit further comprises: The different loop-back gains and equivalent baseband channel responses of the transmitter-to-receiver coupled total interference signals are extracted with reference to the multiple transmitter-to-receiver loop-back channel responses.

5. The device according to claim 1, characterized in that The channel estimation performed by the estimation circuit includes: receiving a baseband input of the transmitter; and A plurality of transmitter-to-receiver loopback channel responses of the image are estimated by performing delayed correlation to project the loopback received signal onto the image of the baseband input of the transmitter, wherein the plurality of transmitter-to-receiver loopback channel responses correspond to the different loopback gains, respectively.

6. The device according to claim 5, characterized in that The channel estimation performed by the estimation circuit further comprises: At least two equivalent baseband transmitter total image channel responses, equivalent baseband receiver total image channel responses, and equivalent baseband channel responses of transmitter-to-receiver coupled total interference images are extracted with reference to the images of the plurality of transmitter-to-receiver loopback channel responses.

7. The device according to claim 1, characterized in that The channel estimation performed by the estimation circuit includes: A plurality of transmitter-to-receiver loopback total DC values ​​are estimated by averaging the loopback received signal, wherein the plurality of transmitter-to-receiver loopback total DC values ​​correspond to the different loopback gains respectively.

8. The device according to claim 7, characterized in that The channel estimation performed by the estimation circuit further comprises: At least one equivalent baseband transmitter total DC offset and an equivalent baseband receiver total DC offset are extracted with reference to the plurality of transmitter-to-receiver loopback total DC values.

9. The device according to claim 1, characterized in that The calibration circuit includes at least one transmitter in-phase and quadrature mismatch compensation circuit, a receiver in-phase and quadrature mismatch compensation circuit, a transmitter direct current removal circuit, and a receiver direct current removal circuit.

10. The device according to claim 1, characterized in that The calibration is a dynamic calibration and the loopback receive signal is generated in response to a baseband input of the transmitter, the baseband input of the transceiver being a modulated signal.

11. The device according to claim 1, characterized in that The calibration is a static calibration and the loopback receive signal is generated in response to a baseband input of the transmitter, the baseband input of the transceiver being a non-modulated signal.

12. A method for calibrating a transceiver, characterized in that: include: controlling a loopback circuit to apply a series of different loopback gains, wherein the loopback circuit is coupled between a mixer output port of a transmitter of the transceiver and a mixer input port of a receiver of the transceiver; receiving, from the receiver, a loopback receive signal at the series of different loopback gains; generating at least one estimate value of the transceiver by performing channel estimation based on at least the loopback received signal; as well as The transceiver is calibrated based on the at least one estimated value.

13. The method according to claim 12, characterized in that The loopback circuit includes a phase shifter, and controlling the loopback circuit to apply the series of different loopback gains includes: The series of different loop gains are applied by controlling the phase shifter to perform multi-phase phase rotation.

14. The method according to claim 12, characterized in that The channel estimation performed by the estimation circuit includes: receiving a baseband input of the transmitter; and A plurality of transmitter-to-receiver loopback channel responses of the signal are estimated by projecting the loopback received signal onto a baseband input of the transmitter by performing delay correlation, wherein the plurality of transmitter-to-receiver loopback channel responses correspond to the different loopback gains, respectively.

15. The method according to claim 14, characterized in that Performing channel estimation based on at least the loopback received signal further includes referencing transmitter-to-receiver loopback channel responses of the plurality of signals to extract different loopback gains and equivalent baseband channel responses of the transmitter-to-receiver coupled total interference signal.

16. The method according to claim 12, characterized in that Performing channel estimation based on at least the loopback received signal includes: receiving a baseband input of a transmitter of the transceiver; and estimating transmitter-to-receiver loopback channel responses of multiple signals by performing delay correlation to project the loopback received signal onto an image of the baseband input of the transmitter of the transceiver, wherein the transmitter-to-receiver loopback channel responses of the multiple signals correspond to different loopback gains, respectively.

17. The method according to claim 16, characterized in that Performing channel estimation based on at least the loopback received signal further includes referencing transmitter-to-receiver loopback channel responses of the plurality of signals to extract at least two equivalent baseband transmitter total image channel responses, equivalent baseband receiver total image channel responses, and equivalent baseband channel responses of transmitter-to-receiver coupled total interference images.

18. The method according to claim 12, characterized in that Performing channel estimation based on at least the loopback received signal includes estimating transmitter-to-receiver loopback total DC values ​​of multiple signals by averaging the loopback received signal, wherein the transmitter-to-receiver loopback total DC values ​​of the multiple signals correspond to different loopback gains respectively.

19. The method according to claim 18, characterized in that Performing channel estimation based on at least the loopback received signal further includes referencing transmitter-to-receiver loopback total DC values ​​of the plurality of signals to extract at least one equivalent baseband transmitter total DC offset and an equivalent baseband receiver total DC offset.

20. The method of claim 12, wherein: Performing calibration on the transceiver based on the at least one estimated value includes performing at least one calibration operation including transmitter in-phase quadrature mismatch compensation, receiver in-phase quadrature mismatch compensation, transmitter DC removal, and receiver DC removal.

21. The method of claim 12, wherein: The calibration is a real-time calibration and the loopback receive signal is generated in response to a baseband input of a transmitter of the transceiver, the baseband input being a modulated signal.

22. The method of claim 12, wherein: The calibration is a static calibration and the loopback receive signal is generated in response to a baseband input of a transmitter of the transceiver, which is a non-modulated signal.

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