Inverter control device and electric vehicle

Abstract

An inverter control device that controls an inverter that converts a direct current into a three-phase alternating current by driving a three-phase inverter circuit and outputs the three-phase alternating current to a motor, the inverter control device being provided with: a PWM pulse generation unit that pulse-width modulates a three-phase voltage command using a carrier wave that changes periodically, and generates a PWM pulse; generating a PWM pulse signal for controlling the driving of the three-phase inverter circuit; and a target harmonic current calculation unit that calculates a target harmonic current on the basis of the driving conditions of the motor. The target harmonic current is used for reducing stator loss and rotor loss respectively generated in a stator and a rotor of the motor due to the harmonic current flowing through the motor according to the PWM pulse signal, and the PWM pulse generation part generates the target harmonic current according to the target harmonic current. The zero-phase voltage, which is the sum of the three-phase output voltages from the three-phase inverter circuit, is non-zero.

Description

Technical Field

[0001] The present invention relates to an inverter control device and an electric vehicle equipped with the inverter control device. Background Art

[0002] Inverter control devices that control the drive of an inverter by PWM (pulse width modulation) control to rotate and drive a motor are widely used. In inverter driving performed by such an inverter control device, in addition to switching losses caused by the switching action of the inverter and fundamental losses caused by fundamental current flowing through the motor, it is known that there is also high-frequency loss generated in the motor due to higher harmonic current flowing through the motor due to PWM control.

[0003] As a technology for reducing the high-frequency loss of a motor by adjusting the phase of a higher harmonic current by controlling a PWM pulse, for example, there is known a technology disclosed in Patent Document 1. Patent Document 1 discloses a motor control device that adjusts the phase difference between a three-phase voltage command and a triangular wave signal Tr as a carrier wave based on a d-axis current Id supplied to the motor and a motor rotation speed ωr indicating the rotation speed of the motor, thereby reducing the eddy current loss We generated in the magnet of the rotor of the motor.

[0004] Prior art literature

[0005] Patent Literature

[0006] Patent Document 1: Japanese Patent Application Publication No. 2022-18168 Summary of the invention

[0007] Problems to be solved by the invention

[0008] In the technology of Patent Document 1, it is necessary to control the phase of the carrier used in PWM control to be constant relative to the fundamental wave of the output current of the inverter corresponding to the three-phase voltage command. However, in synchronous PWM control such as Patent Document 1, the phase accuracy of the carrier relative to the fundamental wave is sometimes reduced due to the angle error caused by the detection error of the motor rotation position and the detection delay of the motor rotation position. In particular, in synchronous PWM control in a multi-pulse area where the number of PWM pulses per electrical angle cycle is more than 15 pulses, it is difficult to control the phase of the carrier relative to the fundamental wave with high accuracy because the width of the electrical angle of each pulse becomes relatively small. As a result, the effect of reducing the higher harmonic loss of the motor may be lost, and the superiority over the asynchronous PWM control may be lost.

[0009] A main object of the present invention is to solve the above-mentioned problems and to provide a technology capable of effectively reducing the high-frequency loss of a motor regardless of the number of PWM pulses per electrical angle cycle.

[0010] Means for solving problems

[0011] The inverter control device of the present invention controls an inverter, which converts a direct current into a three-phase alternating current by driving a three-phase inverter circuit and outputs it to a motor. The inverter control device includes: a PWM pulse generating unit, which uses a periodically changing carrier to perform pulse width modulation on a three-phase voltage instruction to generate a PWM pulse signal for controlling the drive of the three-phase inverter circuit; and a target higher harmonic current calculating unit, which calculates a target higher harmonic current according to the driving conditions of the motor, and the target higher harmonic current is used to reduce the stator loss and rotor loss respectively generated in the stator and rotor of the motor due to the higher harmonic current flowing through the motor according to the PWM pulse signal. The PWM pulse generating unit generates the PWM pulse signal in which the sum of the three-phase output voltages from the three-phase inverter circuit, that is, the zero-phase voltage is non-zero according to the target higher harmonic current.

[0012] The electric vehicle of the present invention comprises: an electric motor having a stator and a rotor; a battery for supplying a direct current; an inverter having a three-phase inverter circuit, which converts the direct current into a three-phase alternating current by driving the three-phase inverter circuit and outputs the three-phase alternating current to the electric motor; and an inverter control device for controlling the inverter, the inverter control device comprising: a PWM pulse generating unit, which performs pulse width modulation on a three-phase voltage instruction using a periodically changing carrier wave to generate a PWM pulse signal for controlling the drive of the three-phase inverter circuit; and a target harmonic current calculating unit, which calculates a target harmonic current according to the driving conditions of the electric motor, the target harmonic current being used to reduce the stator loss and the rotor loss respectively generated in the stator and the rotor due to the harmonic current flowing through the electric motor according to the PWM pulse signal, the PWM pulse generating unit generating the PWM pulse signal in which the zero-phase voltage, which is the sum of the three-phase output voltages from the three-phase inverter circuit, is non-zero according to the target harmonic current.

[0013] Effects of the Invention

[0014] According to the present invention, regardless of the number of PWM pulses per electrical angle cycle, the high-frequency loss of the motor can be effectively reduced. BRIEF DESCRIPTION OF THE DRAWINGS

[0015] Figure 1 It is a block diagram showing the configuration of a motor drive system including an inverter control device according to a first embodiment of the present invention.

[0016] Figure 2It is a diagram showing an example of a method for generating a PWM pulse signal in the inverter control device according to the first embodiment of the present invention.

[0017] Figure 3 This is a diagram showing an example of harmonic loss generated in a motor.

[0018] Figure 4 This is a diagram showing a method of calculating a PWM pulse signal.

[0019] Figure 5 This is a flowchart showing the flow of a calculation process of a target harmonic current and a zero-phase voltage command in the inverter control device according to the first embodiment of the present invention.

[0020] Figure 6 It is a block diagram showing the configuration of a motor drive system including an inverter control device according to a second embodiment of the present invention.

[0021] Figure 7 It is a diagram showing an example of a method for generating a PWM pulse signal in an inverter control device according to the second embodiment of the present invention.

[0022] Figure 8 This is a flowchart showing the flow of a calculation process of a target harmonic current and a target phase value in the inverter control device according to the second embodiment of the present invention.

[0023] Fig. 9 It is a diagram showing the structure of an electric vehicle according to a third embodiment of the present invention. DETAILED DESCRIPTION

[0024] The present invention provides an inverter control device, which controls an inverter through PWM control, and reduces the high-order harmonic loss of a motor, thereby reducing the generation of temperature degradation caused by heating of the motor, and making the inverter output higher by outputting a PWM pulse signal in which the sum of the three-phase AC voltage output from the inverter, i.e., the zero-phase voltage, is non-zero, thereby reducing the generation of temperature degradation caused by heating of the motor. The inverter output is increased. The embodiments of the present invention are described below using the accompanying drawings.

[0025] (First Embodiment)

[0026] Figure 1 1 is a block diagram showing a configuration of a motor drive system 1 including an inverter control device 200 according to a first embodiment of the present invention. The motor drive system 1 is connected to a battery 2 and includes an inverter 100 , an inverter control device 200 , and a motor 300 .

[0027] The battery 2 is a DC voltage source of the inverter 100. The inverter 100 has a three-phase inverter circuit, which is composed of a plurality of semiconductor switching elements corresponding to the three-phase AC voltages applied to the motor 300, and each semiconductor switching element of the three-phase inverter circuit is controlled to be turned on / off by the inverter control device 200. Thus, the DC voltage DCV of the battery 2 is converted into a three-phase AC voltage with a variable voltage and a variable frequency, and is output from the inverter 100 and applied to the motor 300.

[0028] The motor 300 is a synchronous motor that is driven to rotate by supplying a three-phase AC voltage thereto. In order to control the phase of the three-phase AC voltage applied from the inverter 100 to the motor 300 in accordance with the phase of the induced voltage of the motor 300, a rotation position sensor 320 is installed in the motor 300. Here, the rotation position sensor 320 can use, for example, a rotary transformer composed of an iron core and a winding. Alternatively, the rotation position sensor 320 can also be composed of a GMR sensor or a Hall element.

[0029] The inverter control device 200 has functional blocks of a current control unit 210, a current command unit 220, a PWM pulse generation unit 230, a target harmonic current calculation unit 240, a zero-phase voltage command calculation unit 250, a drive signal generation unit 260, a rotation position detection unit 270, and a current detection unit 280. The inverter control device 200 is composed of, for example, a microcomputer, and these functional blocks can be realized by executing a predetermined program in the microcomputer. Alternatively, a part or all of these functional blocks can be realized using hardware circuits such as logic ICs and FPGAs.

[0030] The rotation position detection unit 270 detects the rotation position (electrical angle) θ of the rotor in the motor 300 based on the output signal of the rotation position sensor 320 .

[0031] The current detection unit 280 obtains the three-phase current detection values ​​(Iu, Iv, Iw) flowing in the motor 300 from the current sensor Ict, and performs three-phase / two-phase conversion on these current detection values ​​based on the rotation position θ detected by the rotation position detection unit 270, thereby obtaining the current detection values ​​(Id, Iq) of the dq axes.

[0032] The inverter control device 200 has a current control function for controlling the output of the motor 300. The current command unit 220 calculates the current command value (Id*, Iq*) based on the torque command input from the upper controller (not shown) and the DC voltage DCV of the battery 2, and outputs it to the current control unit 210. The current control unit 210 outputs the voltage command (Vd*, Vq*) so that the current detection value (Id, Iq) detected by the current detection unit 280 is consistent with the current command value (Id*, Iq*) input from the current command unit 220.

[0033] The target harmonic current calculation unit 240 calculates target harmonic currents (Idn*, Iqn*) for reducing stator loss and rotor loss respectively generated in the stator and rotor of the motor 300 based on the torque command input from the upper controller (not shown), the DC voltage DCV of the battery 2, and the time change of the rotation position θ detected by the rotation position detection unit 270, that is, the motor speed. Here, due to the harmonic current flowing in the motor 300 according to the PWM pulse signal output from the PWM pulse generation unit 230 to the drive signal generation unit 260, high-frequency losses (stator loss, rotor loss) are generated in the stator and rotor of the motor 300. The target harmonic current calculation unit 240 determines the driving conditions of the motor 300 according to the torque command and the motor speed, and calculates the d-axis target harmonic current Idn* and the q-axis target harmonic current Iqn* for reducing these harmonic losses based on the determined driving conditions of the motor 300. Note that a method for calculating the target harmonic currents (Idn*, Iqn*) by the target harmonic current calculation unit 240 will be described in detail later.

[0034] The zero-phase voltage command calculation unit 250 calculates a zero-phase voltage command Vn* for adjusting the phase α of the PWM pulse signal generated by the PWM pulse generation unit 230 to a predetermined target phase value based on the target harmonic current (Idn*, Iqn*) calculated by the target harmonic current calculation unit 240 and the DC voltage DCV of the battery 2. The zero-phase voltage command Vn* is a voltage command for changing the zero-phase voltage Vn, which is the sum of the three-phase output voltages Vu, Vv, and Vw output by the inverter 100 to the motor 300 according to the PWM pulse signal, to a non-zero value. In addition, the method by which the zero-phase voltage command calculation unit 250 calculates the zero-phase voltage command Vn* will be described in detail later.

[0035] The PWM pulse generator 230 performs three-phase pulse width modulation (PWM) based on the voltage command (Vd*, Vq*) obtained by the current control unit 210, the zero-phase voltage command Vn* obtained by the zero-phase voltage command calculation unit 250, and the rotation position θ, and generates a PWM pulse signal for controlling the inverter 100. Specifically, for example, the PWM pulse generator 230 performs a 2-phase / 3-phase conversion on the voltage command (Vd*, Vq*) using the rotation position θ, thereby calculating the three-phase voltage commands Vu*, Vv*, and Vw* (U-phase voltage command value Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw*). Then, by subtracting the zero-phase voltage command Vn* from the obtained three-phase voltage commands Vu*, Vv*, and Vw*, the three-phase voltage commands Vcu*, Vcv*, and Vcw* after zero-phase voltage adjustment are obtained, and these voltage command values ​​are compared with the carrier wave that changes periodically at the carrier frequency fc. Thus, the position (phase) of the pulse edge of each phase can be determined, and the PWM pulse signal of each phase can be generated. In this case, the carrier frequency fc can be set constant or changed according to the rotation speed (rotation speed) of the motor 3.

[0036] In addition, as described above, the zero-phase voltage command Vn* output from the zero-phase voltage command calculation unit 250 is a voltage command for making the zero-phase voltage Vn, which is the sum of the phase voltages output by the inverter 100 to the motor 300 according to the PWM pulse signal, non-zero. This is calculated in the zero-phase voltage command calculation unit 250 based on the target harmonic current (Idn*, Iqn*) calculated by the target harmonic current calculation unit 240. Therefore, in the PWM pulse generation unit 230, a PWM pulse signal can be generated in which the zero-phase voltage Vn, which is the sum of the phase voltages output from the three-phase inverter circuit of the inverter 100, becomes non-zero based on the target harmonic current (Idn*, Iqn*).

[0037] The drive signal generator 260 converts the PWM pulse signal generated by the PWM pulse generator 230 into a drive signal DR, and outputs the drive signal DR to the inverter 100. In the inverter 100, each semiconductor switch element of the three-phase inverter circuit is turned on / off controlled according to the drive signal DR. Thus, the three-phase inverter circuit is driven according to the control of the inverter control device 200, and the output voltage from the inverter 100 to the motor 300 is adjusted to adjust the three-phase AC current flowing through the motor 300, thereby controlling the torque of the motor 300.

[0038] In addition, in the above, by Figure 1 The configuration example of the motor drive system 1 in which the torque of the motor 300 is controlled according to the torque command from the host controller has been described. However, the configuration example of the motor drive system 1 in which the torque of the motor 300 is controlled according to the torque command from the host controller can also be applied when other control methods are adopted. Figure 1For example, when the rotation speed of the motor 300 is controlled, the motor rotation speed ωr is calculated based on the time change of the rotation position θ, and a voltage command or a current command is generated in a manner consistent with the speed command from the upper controller. In addition, when the current of the motor 300 is controlled, the current command (Id*, Iq*) from the upper controller is input to the current control unit 210 and the target harmonic current calculation unit 240, respectively.

[0039] Next, use Figure 2 , Figure 3 , Figure 4 , Figure 5 , the details of the PWM pulse control in the inverter control device 200 of this embodiment are described.

[0040] Figure 2 An example of a method for generating a PWM pulse signal in the inverter control device 200 according to the first embodiment of the present invention is shown. Figure 2 In the figure, the arc-shaped modulation wave signal MW shown by the dotted line is equivalent to the fundamental wave component of the three-phase voltage instructions Vu*, Vv*, and Vw* before the zero-phase voltage is adjusted. In addition, the triangular wave carrier signal CW shown by the solid line represents an example of an up-down count timer used in pulse width modulation, which is equivalent to a carrier that changes periodically with a carrier period Tc (carrier frequency fc).

[0041] In the inverter control device 200, the control process of the inverter 100 is performed in each predetermined control cycle in synchronization with the timing at which the carrier signal CW is reset to 0. The inverter control device 200 has an A / D converter (not shown), and in each control cycle, various detection signals used in the control of the inverter 100, such as the detection signal of the current sensor Ict, are sampled and held by the A / D converter. Then, after the A / D conversion time of the A / D converter, the detection signal that has changed into a digital value is obtained, and control processes such as current control are performed. In the sampling and holding operation of the A / D converter, when the capacitor built into the A / D converter is charged with the detection signal, a voltage ripple is generated in the input voltage of the A / D converter. The period of this voltage ripple (A / D conversion noise) is equal to the control period (carrier period Tc). Therefore, the period of the voltage pulsation superimposed on the input voltage in the A / D converter can be regarded as the control period of the inverter control device 200 , and the control period of the inverter control device 200 can be known by measuring the voltage pulsation.

[0042] In addition, in the inverter control device 200, it is not necessarily necessary to make the control period consistent with the current detection period of the current sensor Ict, and for example, the current detection period can be an integer multiple of the control period, such as 2 times. However, even in such a case, according to the relationship of the microcomputer processing in the inverter control device 200, it is necessary to synchronize the current detection timing of the current sensor Ict with the control period of the inverter control device 200, so that the control period of the inverter control device 200 becomes an integer multiple of the current detection period. In short, in the inverter control device 200, the value of an integer multiple of the period of the voltage ripple (A / D conversion noise) generated in the input voltage of the A / D converter can be regarded as the control period (carrier period Tc of the PWM pulse signal).

[0043] The inverter control device 200 of this embodiment has the following features: Figure 1 The control of each power block diagram described in the above can implement phase control of the PWM pulse signal synchronized with the fundamental wave period of the output voltage of the inverter 100 even in the multi-pulse region where the number of PWM pulses per electrical angle period is more than 15 pulses in the asynchronous PWM control state. Figure 2 Provide explanation.

[0044] In the inverter control device 200 of the present embodiment, in the PWM pulse generating unit 230, for example, an adjustment width ΔMW is set for the modulation wave signal MW before the zero-phase voltage is adjusted, and the level difference of the modulation wave signal MW is adjusted to the adjustment width ΔMW in the upward interval where the carrier signal CW increases and the downward interval where the carrier signal CW decreases. Specifically, the modulation wave signal MW is adjusted so that the modulation wave level is (MW-ΔMW / 2) in the upward interval of the carrier signal CW, and the modulation wave level is (MW+ΔMW / 2) in the downward interval of the carrier signal CW. By adjusting the level of the modulation wave signal MW in this way, the edge timing of the PWM pulse signal output from the PWM pulse generating unit 230 can be advanced from θa to θb. That is, the phase of the PWM pulse signal can be advanced in synchronization with the fundamental wave period of the output voltage (output current) of the inverter 100.

[0045] In addition, contrary to the above example, the level of the modulation wave signal MW may be adjusted so that the modulation wave level becomes (MW+ΔMW / 2) in the upward interval of the carrier signal CW and the modulation wave level becomes (MW-ΔMW / 2) in the downward interval of the carrier signal CW. In this way, the timing of each edge of the PWM pulse signal output from the PWM pulse generator 230 can be delayed. That is, the phase of the PWM pulse signal can be delayed in synchronization with the fundamental wave period of the output voltage (output current) of the inverter 100.

[0046] Here, as described above, changing the level of the modulation wave signal MW up and down according to each 1 / 2 cycle of the carrier signal CW, that is, according to each 1 / 2 cycle of the PWM control, is equivalent to changing the zero-phase voltage Vn of the three-phase AC voltage output from the inverter 100 to the motor 300. Therefore, in the PWM pulse generating unit 230, as described above, the zero-phase voltage command Vn* obtained by the zero-phase voltage command calculating unit 250 is used to obtain the three-phase voltage commands Vcu*, Vcv*, and Vcw* after the zero-phase voltage adjustment, and the PWM pulse signals of each phase are generated according to these voltage command values, thereby enabling the above-mentioned phase control of the PWM pulse signal. In addition, in Figure 2 In FIG. 1 , the period of the triangular wave carrier signal CW, that is, the carrier period Tc, represents the period of PWM control.

[0047] In the inverter control device 200 of the present embodiment, as described above, the zero-phase voltage command Vn* is calculated in the zero-phase voltage command calculation unit 250, and the zero-phase voltage command Vn* is used to change the level of the modulation wave signal MW up and down for each 1 / 2 cycle of the PWM control in the PWM pulse generation unit 230, thereby performing phase control of the PWM pulse signal. As a result, for example, in a state of constant torque control at a relatively low rotation speed of the motor 300, even in a multi-pulse region where the number of PWM pulses per electrical angle cycle is more than 15 pulses, it is possible to implement phase control of the PWM pulse signal synchronized with the fundamental wave cycle of the output voltage (output current) of the inverter 100.

[0048] It is known that in the output current of the inverter 100, especially in the frequency twice the carrier frequency fc, the harmonic current component corresponding to the PWM pulse signal becomes larger. Such harmonic current components in the output current of the inverter 100 cause losses (harmonic losses) in the stator and rotor of the motor 300. In the inverter control device 200 of the present embodiment, by performing the phase control of the PWM pulse signal as described above, the phase of the PWM pulse signal relative to the fundamental wave of the output voltage of the inverter 100 is controlled to a predetermined value that can reduce the harmonic losses of the motor 300. At this time, the distribution of the stator loss and the rotor loss can also be adjusted so that the harmonic losses added together are reduced as a whole. As a result, the temperature rise of the stator winding and the temperature rise of the rotor magnet can be reduced according to the operating state of the motor 300, so that the temperature of the motor 300 can be suppressed from reaching the upper limit temperature of the operation, and the continuous output of the motor 300 can be improved.

[0049] Figure 3 2 shows an example of harmonic loss generated in the motor 300. Figure 3In the figure, an example of the magnitude of various higher harmonic losses (AC copper loss, magnet loss, rotor iron loss, core back iron loss, tooth iron loss) generated in the motor 300 is shown for each case where the number of PWM pulses per electrical angle cycle is 9, 15, 21, 27, 33, 39, and 45 and the fundamental wave (sine wave) of the output voltage (output current) of the inverter 100.

[0050] exist Figure 3 Among them, AC copper loss is the resistance loss caused by the AC current flowing through the coil provided in the stator of the motor 300. Magnet loss refers to the loss caused by the eddy current flowing through the permanent magnet provided in the rotor of the motor 300. Rotor iron loss refers to the iron loss generated in the rotor of the motor 300. Core back iron loss refers to the iron loss generated in the core back part of the stator of the motor 300. Tooth iron loss refers to the iron loss generated in the tooth part of the stator of the motor 300. Among these higher harmonic losses, AC copper loss, core back iron loss and tooth iron loss are equivalent to stator loss, and magnet loss and rotor iron loss are equivalent to rotor loss.

[0051] The above-mentioned high-order harmonic losses can be Figure 3 The calculation is performed by using the mathematical formulas (1) to (4) shown in the figure. In the mathematical formulas (1) to (4), n represents the order of the higher harmonics, and N represents the upper limit of the order included in the calculation object of the higher harmonic loss. In addition, f1 represents the frequency of the fundamental wave of the output voltage (output current) of the inverter 100, Ian represents the nth harmonic component of the output current, and Idn represents the nth harmonic component of the d-axis current.

[0052] In mathematical formula (1), Wih_S represents the stator iron loss obtained by adding the core back iron loss and the tooth iron loss together, Kieh_s represents the proportional constant of the stator iron loss, αs represents the fitting coefficient s, es1 represents the fitting coefficient s1, and es2 represents the fitting coefficient s2. Similarly, in mathematical formula (2), PAC_h represents the AC copper loss, KAC_h represents the proportional constant of the AC copper loss, γ represents the fitting coefficient, and ec represents the fitting coefficient c. In mathematical formula (3), Wieh_R represents the rotor iron loss, Kieh_R represents the proportional constant of the rotor iron loss, αr represents the fitting coefficient r, er1 represents the fitting coefficient r1, and er2 represents the fitting coefficient r2. In mathematical formula (4), Pmag represents the magnet loss, Kmag represents the proportional constant of the magnet loss, χ represents the fitting coefficient, and em represents the fitting coefficient m. The various proportional constants and fitting coefficients in these mathematical formulas can be pre-set in the inverter control device 200 based on the results of the magnetic field analysis and actual measurement performed in advance.

[0053] exist Figure 3In the example of higher harmonic loss shown in FIG. 1 , it can be seen that the smaller the number of PWM pulses per electrical angle cycle is compared to the rightmost sine wave component, the higher the sensitivity of AC copper loss and magnet loss is as the higher harmonic current increases, and these higher harmonic losses increase. In addition, as shown in mathematical formulas (2) and (4), the magnet loss can be derived from an approximate formula represented by the exponential power of the nth harmonic component of the d-axis current based on the PWM pulse signal, and the AC copper loss can be derived from an approximate formula represented by the exponential power of the nth harmonic component of the output current based on the PWM pulse signal.

[0054] Figure 4 In the inverter control device 200 of the present embodiment, as described above, the PWM pulse generating unit 230 generates a PWM pulse signal based on the zero-phase voltage command Vn* calculated by the zero-phase voltage command calculating unit 250. Figure 4 As shown, the phase α of the PWM pulse signal can be controlled to a predetermined target phase value while keeping the pulse width tp of the PWM pulse signal constant (the fundamental voltage constant). As a result, the d-axis harmonic voltage Vdn and the q-axis harmonic voltage Vqn in the output voltage of the inverter 100 corresponding to the PWM pulse signal can be controlled respectively, and the d-axis harmonic current Idn and the q-axis harmonic current Iqn can be adjusted to the desired values. In addition, the phase α of the harmonic voltages Vdn, Vqn and the harmonic currents Idn, Iqn and the PWM pulse signal is respectively determined by Figure 4 The mathematical formula (5) and (6) show that the zero-phase voltage Vn offset Vz corresponding to the zero-phase voltage command Vn* is given by Figure 4 The mathematical formula (7) shows that.

[0055] As described above, in the PWM pulse generating unit 230, by adjusting the phase α of the PWM pulse signal, the average output voltage of the inverter 100 is maintained at the desired voltage, while the magnitude and phase of the harmonic component are changed respectively, and the magnitude of the d-axis harmonic current Idn and the q-axis harmonic current Iqn is adjusted, so that the harmonic loss in the motor 300 can be reduced. Specifically, for example, in a high rotation area where the requirement for magnet temperature is strict, by controlling the phase α to near 0 or π, the magnet loss can be minimized, thereby suppressing the increase in magnet temperature. In addition, the d-axis harmonic current Idn can also be reduced, and the high-frequency loss (AC copper loss and magnet loss) corresponding to the q-axis harmonic current Iqn can be adjusted. For example, in an area where it is desired to reduce the torque pulsation caused by the harmonic current, such as a low speed, low torque area, etc., by controlling the phase α to near ±π / 2, although the magnet loss increases, the torque pulsation can be minimized.

[0056] In addition, since the selection of the harmonic loss and the torque pulsation in the motor 300 is a trade-off relationship, it is preferable to tabulate the relationship between the torque and the speed in advance for each value of the phase α of the PWM pulse signal in the inverter control device 200. In addition, as the motor 300, a motor whose q-axis component inductance Lq is about twice or more of the d-axis component inductance Ld is sometimes used. In such a case, when determining the phase α, a voltage value of about twice the d-axis harmonic voltage Vdn can be allocated to the q-axis harmonic voltage Vqn.

[0057] In the present embodiment, as described above, the target harmonic current calculation unit 240 calculates the target harmonic current (Idn*, Iqn*) corresponding to the driving condition of the motor 300, and the PWM pulse generation unit 230 adjusts the phase α of the PWM pulse signal by adjusting the zero-phase voltage Vn according to the zero-phase voltage command Vn* calculated by the zero-phase voltage command calculation unit 250 based on the target harmonic current (Idn*, Iqn*). Thus, the d-axis harmonic voltage Vdn and the q-axis harmonic voltage Vqn in the output voltage of the inverter 100 are controlled respectively, and the d-axis harmonic current Idn and the q-axis harmonic current Iqn are feed-forward controlled to desired values. However, feedback control may be performed instead of such feed-forward control. Specifically, for example, the d-axis harmonic current Idn and the q-axis harmonic current Iqn can be detected from the current detection values ​​(Id, Iq) obtained by the current detection unit 280, and the differences between these values ​​and the target harmonic current (Idn*, Iqn*) can be calculated to determine the zero-phase voltage command Vn*. In addition, in addition to this, it is also possible to determine the zero-phase voltage command Vn* corresponding to the target harmonic current (Idn*, Iqn*) by any method to adjust the phase α of the PWM pulse signal.

[0058] Figure 5 This is a flowchart showing the flow of calculation processing of the target harmonic current (Idn*, Iqn*) and the zero-phase voltage command Vn* in the inverter control device 200 according to the first embodiment of the present invention. Figure 5 The process shown in the flowchart is executed by the target harmonic current calculation unit 240 and the zero-phase voltage command calculation unit 250 of the inverter control device 200, for example, at each predetermined control cycle.

[0059] In step S1, the target harmonic current calculation unit 240 obtains the target harmonic current (Idn*, Iqn*) corresponding to the driving condition of the motor 300. For example, for each combination of the DC voltage DCV, the torque command, and the motor speed, the value of the target harmonic current (Idn*, Iqn*) that can most effectively reduce the harmonic loss relative to these driving conditions is pre-calculated by simulation or experiment, and based on the calculation result, the value of the optimal target harmonic current (Idn*, Iqn*) for each driving condition is pre-set by a mathematical formula or a table. By using this mathematical formula or table, the target harmonic current (Idn*, Iqn*) corresponding to the driving condition of the motor 300 can be obtained.

[0060] In step S2, the target harmonic current calculation unit 240 estimates the harmonic loss and torque ripple caused by the target harmonic current (Idn*, Iqn*) calculated in step S1. Specifically, for example, Figure 4 The nth harmonic component Ian of the output current of the inverter 100 and the nth harmonic component Idn of the d-axis current are calculated by using the mathematical formula (5) shown in FIG. Figure 3 The mathematical formulas (1) to (4) shown in the figure calculate various harmonic losses (stator iron loss, AC copper loss, rotor iron loss, and magnet loss) generated in the motor 300. In addition, as described above, the sensitivity of AC copper loss and magnet loss increases as the number of PWM pulses per electrical angle cycle decreases, so only AC copper loss and magnet loss can be used as the calculation object of higher harmonic loss.

[0061] In step S3, the target harmonic current calculation unit 240 determines whether the magnitude of the torque pulsation estimated in step S2 is greater than a predetermined value. Here, the magnitude of the torque pulsation obtained in step S2 is compared with a predetermined threshold value set in advance. As a result, if the magnitude of the torque pulsation is greater than the threshold, the process proceeds to step S4. If it is less than the threshold, the value of the target harmonic current (Idn*, Iqn*) calculated in step S1 is output to the zero-phase voltage command calculation unit 250, and the process proceeds to step S5.

[0062] In step S4, the target harmonic current calculation unit 240 corrects the target harmonic current (Idn*, Iqn*) calculated in step S1. Here, based on the DC voltage DCV, torque command, motor speed and other motor driving conditions used when calculating the target harmonic current (Idn*, Iqn*) in step S1, the magnitude of the harmonic loss and torque pulsation estimated in step S2, etc., a correction coefficient equivalent to the weighted coefficient of the d-axis target harmonic current Idn* and the q-axis target harmonic current Iqn* is determined, and the target harmonic current (Idn*, Iqn*) is weightedly corrected using the correction coefficient. For example, the target harmonic current (Idn*, Iqn*) can be corrected using a correction coefficient pre-set for each motor driving condition. After the processing of step S4 is completed, the values ​​of the target harmonic currents (Idn*, Iqn*) corrected in step S4 are output to the zero-phase voltage command calculation unit 250, and the process proceeds to step S5.

[0063] In step S5, the zero-phase voltage command calculation unit 250 obtains the target harmonic current (Idn*, Iqn*) output from the target harmonic current calculation unit 240, and based on the target value relative to the phase α of the PWM pulse signal is obtained. Figure 4 The target value of the phase α corresponding to the target harmonic current (Idn*, Iqn*) is calculated using equation (6).

[0064] In step S6, the zero-phase voltage command calculation unit 250 obtains the offset Vz of the zero-phase voltage Vn based on the target value of the phase α obtained in step S5. Figure 4 The mathematical formula (7) is used to calculate the offset Vz of the zero-phase voltage Vn corresponding to the target value of the phase α.

[0065] In step S7, the zero-phase voltage command calculation unit 250 outputs a zero-phase voltage command Vn* corresponding to the offset Vz of the zero-phase voltage Vn obtained in step S6. The PWM pulse generation unit 230 adjusts the zero-phase voltage Vn according to the zero-phase voltage command Vn*. Figure 2 As described in , the modulated wave signal MW is fluctuated up and down every 1 / 2 cycle of the PWM control, and a level difference is generated between the upward interval where the carrier signal CW increases and the downward interval where the carrier signal CW decreases. As a result, the phase α of the PWM pulse signal is adjusted to match the target value obtained in step S5.

[0066] In the inverter control device 200 of the present embodiment described above, the target harmonic current calculation unit 240 calculates the target harmonic current (Idn*, Iqn*) corresponding to the driving condition of the motor 300, and the zero-phase voltage command calculation unit 250 outputs the zero-phase voltage command Vn* based on the target harmonic current (Idn*, Iqn*). In the PWM pulse generation unit 230, a PWM pulse signal is generated according to the zero-phase voltage command Vn* output from the zero-phase voltage command calculation unit 250, thereby adjusting the phase α of the PWM pulse signal. Therefore, not limited to the case of the synchronous PWM control state, in the case of the asynchronous PWM control state in which the ratio of the period of the three-phase voltage commands Vu*, Vv*, and Vw* for generating the PWM pulse signal to the period of the carrier is a non-integer, the phase α of the PWM pulse signal can be adjusted to an arbitrary value. Therefore, even in a multi-pulse region where the number of PWM pulses per electrical angle period is large, it is possible to achieve a reduction in harmonic loss by controlling the phase of the PWM pulse signal. For example, in a constant torque control region where the rotation speed of the motor 300 is relatively low, harmonic losses (stator losses, rotor losses) can be reduced even for frequencies where harmonic current components in the output current of the inverter 100 are large (frequencies twice the carrier frequency fc).

[0067] In addition, in the inverter control device 200 of the present embodiment described above, by adjusting the phase α of the PWM pulse signal based on the zero-phase voltage command Vn*, it is also possible to adjust the distribution of the stator loss and the rotor loss in the higher harmonic loss, and the distribution of the higher harmonic loss and the torque pulsation. As a result, for example, according to the driving conditions of the motor 300, the torque pulsation can be adjusted while reducing the temperature rise of the stator winding and the rotor magnet, so that the continuous output of the motor and low noise and low vibration can be achieved at the same time.

[0068] In addition, in the inverter control device 200 of the present embodiment, when the target harmonic current (Idn*, Iqn*) is weightedly corrected in such a manner as to reduce the d-axis harmonic current Idn and increase the q-axis harmonic current Iqn in order to minimize the harmonic loss of the motor 300, the torque pulsation caused by the harmonic current in the motor 300 may become larger. In such a case, in order to reduce the harmonic current flowing in the motor 300 and suppress the harmonic loss, the frequency of the carrier used in the generation of the PWM pulse signal may be switched to a higher frequency than usual in the PWM pulse generator 230. Specifically, for example, in the PWM pulse generator 230, the frequency of the carrier is set to be switchable between the high frequency side and the low frequency side. Furthermore, the frequency of the carrier wave is normally set to the low frequency side, and when the output torque of the motor 300 corresponding to the torque command input from the upper controller is within a predetermined first range, and the magnitude of the higher harmonic loss (stator loss and rotor loss) estimated by the zero-phase voltage command calculation unit 250 is within a predetermined second range, the frequency of the carrier wave is switched from the low frequency side to the high frequency side. In this way, the torque pulsation generated in the motor 300 can be suppressed, and the higher harmonic loss of the motor 300 can be reduced. In particular, in the inverter 100, when a semiconductor switching element of the three-phase inverter circuit uses a high-response semiconductor element such as SiC and GaN, a good effect can be obtained.

[0069] According to the first embodiment of the present invention described above, the following effects are achieved.

[0070] (1) The inverter control device 200 controls the inverter 100, which converts a direct current into a three-phase alternating current by driving a three-phase inverter circuit and outputs the converted current to the motor 300. The inverter control device 200 includes: a PWM pulse generating unit 230, which performs pulse width modulation on three-phase voltage commands Vu*, Vv*, and Vw* using a periodically changing carrier wave to generate a PWM pulse signal for controlling the driving of the three-phase inverter circuit; and a target harmonic current calculating unit 240, which calculates a target harmonic current (Idn*, Iqn*) based on the driving conditions of the motor, and the target harmonic current (Idn*, Iqn*) is used to reduce the stator loss and rotor loss respectively generated in the stator and rotor of the motor 300 due to the harmonic current flowing in the motor 300 according to the PWM pulse signal. The PWM pulse generator 230 generates a PWM pulse signal in which the zero-phase voltage Vn, which is the sum of the three-phase output voltages Vu, Vv, and Vw from the three-phase inverter circuit, becomes non-zero based on the target harmonic current (Idn*, Iqn*). As a result, the phase α of the PWM pulse signal can be controlled to a predetermined target phase value, and the magnitudes of the d-axis harmonic current Idn and the q-axis harmonic current Iqn in the motor 300 can be adjusted. As a result, regardless of the number of PWM pulses per electrical angle cycle, the high-frequency loss of the motor 300 can be effectively reduced.

[0071] (2) The inverter control device 200 includes a zero-phase voltage command calculation unit 250, which calculates a zero-phase voltage command Vn* for changing the zero-phase voltage Vn for each 1 / 2 cycle of the carrier based on the target higher harmonic current (Idn*, Iqn*). The PWM pulse generation unit 230 generates a PWM pulse signal based on the zero-phase voltage command Vn*. In this way, the zero-phase voltage Vn can be changed to non-zero, and a PWM pulse signal can be generated that can control the phase α of the PWM pulse signal to a predetermined target phase value.

[0072] (3) The PWM pulse generating unit 230 generates a PWM pulse signal in a manner that causes the zero-phase voltage Vn to change synchronously with the three-phase voltage commands Vu*, Vv*, Vw*, in an asynchronous PWM control state in which the ratio of the period of the three-phase voltage commands Vu*, Vv*, Vw* to the period of the carrier is a non-integer. Thus, even in the case of an asynchronous PWM control state, the phase α of the PWM pulse signal can be adjusted to a predetermined target phase value to reduce the high-frequency loss of the motor 300.

[0073] (4) The target harmonic current calculation unit 240 corrects the calculated target harmonic current (Idn*, Iqn*) (step S4). In this case, the PWM pulse generation unit 230 generates a PWM pulse signal based on the corrected target harmonic current (Idn*, Iqn*). Thus, for example, when the magnitude of the torque pulsation of the motor 300 is greater than a predetermined value, the high-frequency loss of the motor 300 can be adjusted to achieve the suppression of the torque pulsation.

[0074] (5) The PWM pulse generator 230 may switch the carrier frequency to a higher frequency than usual when the output torque of the motor 300 is within a predetermined first range and the magnitude of the harmonic loss obtained by adding the stator loss and the rotor loss is within a predetermined second range. In this way, the torque pulsation generated in the motor 300 can be suppressed and the harmonic loss of the motor 300 can be reduced.

[0075] (Second Embodiment)

[0076] Next, a second embodiment of the present invention will be described. In this embodiment, an example is described in which a target phase value relative to the phase α of the PWM pulse signal is set without using the above-mentioned zero-phase voltage command Vn*, and a PWM pulse signal is generated based on the target phase value.

[0077] Figure 6 2 is a block diagram showing a configuration of a motor drive system 1A including an inverter control device 200A according to a second embodiment of the present invention. The motor drive system 1A of this embodiment is different from the motor drive system 1A described in the first embodiment. Figure 1 Compared with the motor drive system 1 of FIG. 1 , the inverter control device 200A has a part of a structure different from that of FIG. 1 . Other points are the same as those of the motor drive system 1 , and therefore, description thereof will be omitted unless otherwise required.

[0078] The inverter control device 200A includes functional blocks of a current control unit 210, a current command unit 220, a PWM pulse generation unit 230A, a target harmonic current calculation unit 240, a target phase value calculation unit 250A, a drive signal generation unit 260, a rotation position detection unit 270, and a current detection unit 280. In addition, the functional blocks other than the PWM pulse generation unit 230A and the target phase value calculation unit 250A are common to the functional blocks of the inverter control device 200 in the first embodiment, and therefore the description thereof is omitted below.

[0079] The target phase value calculation unit 250A calculates the target phase value relative to the phase α of the PWM pulse signal generated by the PWM pulse generation unit 230 based on the target harmonic current (Idn*, Iqn*) calculated by the target harmonic current calculation unit 240 and the DC voltage DCV of the battery 2, and outputs a phase command value α* corresponding to the target phase value. The phase command value α* is equivalent to the target value of the phase α used in the calculation of the zero-phase voltage command Vn* in the first embodiment. Figure 4 It is calculated using the mathematical formula (6) described in .

[0080] The PWM pulse generator 230A performs three-phase pulse width modulation (PWM) based on the voltage command (Vd*, Vq*) obtained by the current control unit 210, the phase command value α* output from the target phase value calculation unit 250A, and the rotation position θ, and generates a PWM pulse signal for controlling the inverter 100. Specifically, for example, the PWM pulse generator 230A performs a 2-phase / 3-phase conversion on the voltage command (Vd*, Vq*) using the rotation position θ, thereby calculating the three-phase voltage commands Vu*, Vv*, and Vw* (U-phase voltage command value Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw*). Then, based on the obtained three-phase voltage commands Vu*, Vv*, and Vw* and the phase command value α*, a comparison matching timer for the carrier is set in each phase, and the comparison is made with the carrier that changes periodically with the carrier frequency fc. In this way, the position (phase) of the pulse edge of each phase can be determined, and the PWM pulse signal of each phase can be generated.

[0081] Figure 7 FIG. 2 shows an example of a method for generating a PWM pulse signal in an inverter control device 200A according to a second embodiment of the present invention. Figure 7 As shown, the carrier signal CW is not a triangular wave, but is implemented by a sawtooth wave up-down counting timer.

[0082] In the inverter control device 200A of the present embodiment, in the PWM pulse generating unit 230A, the value of the comparison matching register set according to the three-phase voltage instructions Vu*, Vv*, and Vw* is offset based on the phase instruction value α*. Specifically, for example, Figure 7As shown in FIG. 1 , the comparison matching register values ​​CMP1A and 2A set corresponding to the three-phase voltage commands Vu*, Vv*, and Vw* are offset to values ​​reduced by an amount corresponding to the phase command value α*, thereby setting the comparison matching register values ​​CMP1B and 2B. By offsetting the comparison matching register values ​​in this way, the edge timing of the PWM pulse signal output from the PWM pulse generating unit 230A can be advanced from θa to θb, similarly to the first embodiment. That is, the phase of the PWM pulse signal can be advanced in synchronization with the fundamental wave period of the output voltage (output current) of the inverter 100.

[0083] In addition, contrary to the above example, the value of the comparison matching register may be offset to increase the value corresponding to the phase command value α*. In this way, the timing of each edge of the PWM pulse signal output from the PWM pulse generating unit 230A can be delayed. That is, the phase of the PWM pulse signal can be delayed in synchronization with the fundamental wave period of the output voltage (output current) of the inverter 100.

[0084] In the inverter control device 200A of the present embodiment, as described above, the target phase value corresponding to the phase α of the PWM pulse signal is calculated in the target phase value calculation unit 250A, and the value of the comparison matching register is shifted in the PWM pulse generation unit 230A using the phase command value α* corresponding to the target phase value, thereby performing phase control of the PWM pulse signal. Thus, as in the first embodiment, for example, in a state of constant torque control where the rotation speed of the motor 300 is relatively low, even in a multi-pulse region where the number of PWM pulses per electrical angle cycle is more than 15 pulses, it is possible to perform phase control of the PWM pulse signal synchronized with the fundamental wave period of the output voltage (output current) of the inverter 100. In addition, in the present embodiment, the phase of the PWM pulse signal can be controlled only by shifting the value of the comparison matching register, so there is also an advantage that the processing load of the microcomputer used as the inverter control device 200A can be reduced.

[0085] Figure 8 This is a flowchart showing the flow of a calculation process of a target harmonic current (Idn*, Iqn*) and a target phase value in the inverter control device 200A according to the second embodiment of the present invention. Figure 8 The process shown in the flowchart is executed by the target harmonic current calculation unit 240 and the target phase value calculation unit 250A of the inverter control device 200A, for example, at each predetermined control cycle.

[0086] In steps S1 to S5, the same steps as those described in the first embodiment are respectively performed. Figure 5The same processing as steps S1 to S5 is performed. In the present embodiment, the processing of step S5 is performed by the target phase value calculation unit 250A.

[0087] In step S7A, the target phase value calculation unit 250A outputs a phase command value α* corresponding to the target value of the phase α obtained in step S5. The PWM pulse generation unit 230A shifts the value of the comparison matching register according to the phase command value α*, thereby Figure 7 As described in , the phase α of the PWM pulse signal is adjusted to match the target value obtained in step S5.

[0088] According to the second embodiment of the present invention described above, the inverter control device 200A includes a target phase value calculation unit 250A, which calculates a target phase value relative to the phase α of the PWM pulse signal based on the target higher harmonic current (Idn*, Iqn*). The PWM pulse generation unit 230A generates a PWM pulse signal based on the target phase value. Thus, as in the first embodiment, a PWM pulse signal that can control the phase α of the PWM pulse signal to a predetermined target phase value can be generated.

[0089] (Third Embodiment)

[0090] Next, a third embodiment of the present invention will be described. In this embodiment, an example in which the motor drive system described in each of the first and second embodiments is mounted on an electric vehicle will be described.

[0091] Fig. 9 The structure of an electric vehicle 600 according to a third embodiment of the present invention is shown. The electric vehicle 600 has a power transmission system that uses a motor drive system 1B1 (motor 300B1, inverter 100B1, inverter control device 200B1) for traveling and a motor drive system 1B2 (motor 300B2, inverter 100B2, inverter control device 200B2) for power generation. The inverter control devices 200B1 and 200B2 have the same structure as the inverter control device 200 described in the first embodiment or the inverter control device 200A described in the second embodiment. In addition, the motors 300B1 and 300B2 have the same structure as the motor 300, and the inverters 100B1 and 100B2 have the same structure as the inverter 100.

[0092] A front wheel axle 601 is rotatably supported at the front of the electric vehicle 600, and front wheels 602 and 603 are provided at both ends of the front wheel axle 601. A rear wheel axle 604 is rotatably supported at the rear of the electric vehicle 600, and rear wheels 605 and 606 are provided at both ends of the rear wheel axle 604.

[0093] A differential gear 611 as a power distribution mechanism is provided at the center of the front wheel axle 601, and the rotational driving force transmitted by the engine 610 or the motor 300B1 via the transmission 612 is distributed to the left and right front wheel axles 601. The engine 610 and the motors 300B1 and 300B2 have a mechanism that can be mechanically connected / disconnected using a mechanism installed in the transmission 612. Therefore, it is possible to selectively switch between a series hybrid mode and a parallel hybrid mode. In the series hybrid mode, the engine 610 and the motor 300B2 are mechanically combined, and only the mechanical output of the motor 300B1 is transmitted to the differential gear 611 to make the vehicle run. In the parallel hybrid mode, the mechanical output of the engine 610 and the motor 300B1 are transmitted to the differential gear 611 together to make the vehicle run. As a result, the electric vehicle 600 can arbitrarily switch between the series hybrid mode and the parallel hybrid mode to match the driving scenes such as urban driving and highway driving, and achieve a balance between the vehicle's sports performance and the vehicle's cruising range.

[0094] The inverter 100B1 converts the DC power supplied from the high-voltage battery 622 into three-phase AC power according to the control of the inverter control device 200B1. The three-phase AC power output from the inverter 100B1 is supplied to the stator coil of the stator of the motor 300B1. As a result, the rotor of the motor 300B1 rotates, generating a rotational driving force corresponding to the three-phase AC power (current). The rotor of the motor 300B2 is driven to rotate by the mechanical output from the engine 610. As a result, the three-phase AC power is generated in the stator coil of the motor 300B2 and input to the inverter 100B2. According to the control of the inverter control device 200B2, the inverter 100B2 converts the three-phase AC power input from the motor 300B2 into DC power and charges the high-voltage battery 622.

[0095] When the electric vehicle 600 is traveling, in order to extend the traveling distance, it is desirable to reduce the loss of the motors 300B1 and B2. In this embodiment, as described in the first and second embodiments, the inverter control devices 200B1 and B2 calculate the target harmonic currents (Idn*, Iqn*), respectively, and adjust the phase α of the PWM pulse signal based on the target harmonic currents, thereby reducing the harmonic loss of the motors 300B1 and B2.

[0096] For example, the inverter control device 200B1 of the motor drive system 1B1 inputs a torque command corresponding to the accelerator pedal operation amount of the electric vehicle 600, performs current feedback control so that it becomes a three-phase AC power (current) corresponding to the torque command, and calculates the voltage command value for the inverter 100B1. In addition, the voltage command value and the carrier are compared to perform pulse width modulation to generate a PWM pulse signal for driving the inverter 100B1. At this time, in the target harmonic current calculation unit 240 of the inverter control device 200B1, the target harmonic current (Idn*, Iqn*) corresponding to the driving condition of the motor 300B1 is calculated using the torque command, the time change of the rotation position θ, that is, the motor speed, and the DC voltage DCV of the high-voltage battery 622. In the zero-phase voltage command calculation unit 250 or the target phase value calculation unit 250A, the target phase value corresponding to the phase α of the PWM pulse signal is obtained based on the target higher harmonic current (Idn*, Iqn*) calculated by the target higher harmonic current calculation unit 240, and the zero-phase voltage command Vn* or the phase command value α* is calculated. In the PWM pulse generation unit 230, based on the zero-phase voltage command Vn* or the phase command value α* from the zero-phase voltage command calculation unit 250 or the target phase value calculation unit 250A, a PWM pulse signal is generated in such a way that the phase α becomes the target phase value. Thus, a PWM pulse signal is generated in which the zero-phase voltage Vn, which is the sum of the three-phase output voltages Vu, Vv, and Vw output by the inverter 100B1 to the motor 300B1, becomes non-zero. Then, the three-phase inverter circuit included in the inverter 100B1 is driven by the drive signal DR generated by the drive signal generating unit 260 based on the PWM pulse signal, thereby reducing the high-frequency loss (stator loss and rotor loss) of the motor 300B1.

[0097] In addition, in the electric motor drive system 1B2 for power generation, similarly to the electric motor drive system 1B1 for traveling, the high-frequency loss of the electric motor 300B2 can be reduced.

[0098] As described above, in the electric vehicle 600, by adjusting the phase of the PWM pulse signal in the motor drive systems 1B1 and 1B2, it is possible to eliminate specific order components in the harmonic components in the three-phase AC power output from the inverters 100B1 and 100B2 to the motors 300B1 and 300B2. As a result, it is possible to reduce the harmonic losses in the motors 300B1 and 300B2, and as a result, for example, it is possible to alleviate the output limitation caused by the temperature rise of the motors 300B1 and 300B2 on an uphill road.

[0099] In the present embodiment, the case where the electric vehicle 600 is a hybrid vehicle is described, but the present invention is also applicable to a plug-in hybrid vehicle, an electric vehicle, or the like, and the same effects can be obtained in such a case.

[0100] In each of the above-mentioned embodiments, examples are described in which the inverter control device is different from the inverter and the motor in different structures. However, as long as the above-mentioned functions are present, the present invention can also be applied to an inverter device in which the inverter control device is integrated with the inverter, or a motor drive system in which the inverter device is integrated with the motor. In addition, in each of the above-mentioned embodiments, an application example in a multi-pulse region in an asynchronous PWM control state is described. However, the present invention can also be applied in a synchronous PWM control state in which the ratio of the period of the three-phase voltage instructions Vu*, Vv*, and Vw* used in the generation of the PWM pulse signal to the period of the carrier is an integer. By applying the present invention in both the asynchronous PWM control state and the synchronous PWM control state, the entire rotation speed range of the motor can be covered.

[0101] In addition, the present invention is not limited to the above-mentioned embodiment, and various changes can be made without departing from the scope of the present invention.

[0102] Description of Reference Numerals

[0103] 1, 1A, 1B1, 1B2…Motor drive system

[0104] 2…Battery

[0105] 100, 100B1, 100B2…Inverter

[0106] 200, 200A, 200B1, 200B2…Inverter control device

[0107] 210…Current control unit

[0108] 220…Current command unit

[0109] 230, 230A…PWM pulse generating unit

[0110] 240…Target harmonic current calculation unit

[0111] 250…Zero-phase voltage command calculation unit

[0112] 250A…Target phase value calculation unit

[0113] 260…Drive signal generating unit

[0114] 270…Rotation position detection unit

[0115] 280…Current detection unit

[0116] 300, 300B1, 300B2…Motor

[0117] 320...Rotary Position Sensor

[0118] 600…Electric vehicles.

Claims

1. An inverter control device for controlling an inverter, wherein the inverter converts a direct current into a three-phase alternating current by driving a three-phase inverter circuit and outputs the three-phase alternating current to a motor, characterized in that: The inverter control device comprises: a PWM pulse generating unit that performs pulse width modulation on a three-phase voltage command using a periodically changing carrier wave to generate a PWM pulse signal for controlling the driving of the three-phase inverter circuit; as well as a target harmonic current calculation unit, which calculates a target harmonic current according to a driving condition of the motor, the target harmonic current being used to reduce a stator loss and a rotor loss respectively generated in a stator and a rotor of the motor due to a harmonic current flowing through the motor according to the PWM pulse signal, The PWM pulse generating unit generates the PWM pulse signal such that a zero-phase voltage, which is a sum of three-phase output voltages from the three-phase inverter circuit, is non-zero, based on the target harmonic current.

2. The inverter control device according to claim 1, characterized in that: The inverter control device includes a zero-phase voltage command calculation unit that calculates a zero-phase voltage command for changing the zero-phase voltage for each 1 / 2 cycle of the carrier based on the target harmonic current. The PWM pulse generating unit generates the PWM pulse signal according to the zero-phase voltage command.

3. The inverter control device according to claim 1, characterized in that: The inverter control device includes a target phase value calculation unit that calculates a target phase value with respect to the phase of the PWM pulse signal based on the target harmonic current. The PWM pulse generating unit generates the PWM pulse signal based on the target phase value.

4. The inverter control device according to claim 1, characterized in that: The PWM pulse generating unit generates the PWM pulse signal such that the zero-phase voltage changes in synchronization with the three-phase voltage command in an asynchronous PWM control state in which a ratio between a period of the three-phase voltage command and a period of the carrier wave is a non-integer.

5. The inverter control device according to claim 1, characterized in that: The target harmonic current calculation unit corrects the calculated target harmonic current. The PWM pulse generating unit generates the PWM pulse signal based on the corrected target harmonic current.

6. The inverter control device according to claim 1, characterized in that: When the output torque of the motor is within a predetermined first range and the magnitude of the harmonic loss obtained by adding the stator loss to the rotor loss is within a predetermined second range, the PWM pulse generating unit switches the frequency of the carrier to a frequency higher than normal.

7. An electric vehicle, characterized in that: have: an electric motor having a stator and a rotor; a battery, which supplies a direct current; an inverter having a three-phase inverter circuit, which converts the direct current into a three-phase alternating current by driving the three-phase inverter circuit and outputs the three-phase alternating current to the motor; and an inverter control device, which controls the inverter, The inverter control device comprises: a PWM pulse generating unit that performs pulse width modulation on a three-phase voltage command using a periodically changing carrier wave to generate a PWM pulse signal for controlling the driving of the three-phase inverter circuit; as well as a target harmonic current calculation unit, which calculates a target harmonic current according to a driving condition of the motor, the target harmonic current being used to reduce a stator loss and a rotor loss respectively generated in the stator and the rotor due to a harmonic current flowing through the motor according to the PWM pulse signal, The PWM pulse generating unit generates the PWM pulse signal such that a zero-phase voltage, which is a sum of three-phase output voltages from the three-phase inverter circuit, is non-zero, based on the target harmonic current.

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