Motor rotating speed control system and method

Abstract

The invention discloses a motor rotating speed control system which comprises a current acquisition module, a rotational inertia identification module, a load torque observation module, a rotating speed prediction module, a value function evaluation module and a PWM signal generation module. The output end of the current acquisition module is connected with the input end of the rotational inertia identification module and the input end of the load torque observation module. The output end of the rotational inertia identification module is connected with the input end of the load torque observation and rotation speed prediction module. The output end of the load torque observation module is respectively connected with the input ends of the rotating speed prediction module and the value function evaluation module; the output end of the rotating speed prediction module is connected with the input end of the value function evaluation module; the output end of the value function evaluation module is connected with the input end of the PWM signal generation module; and the PWM signal generation module is connected with the inverter. According to the invention, the dynamic response performance of the motor control system is effectively improved, the adjustment complexity caused by traditional PI controller cascading is avoided, and the system is more stable and efficient.

Description

Technical Field

[0001] The invention relates to the technical field of motor control, and in particular to a motor speed control system and method based on Taylor series and moment of inertia identification. Background Art

[0002] With the widespread application of electric drive systems, traditional motor control methods, such as PI controllers, have certain deficiencies in dynamic performance and parameter adjustment. Especially for multi-phase motor systems, due to their complex structure, traditional PI control methods perform poorly in response to load changes and complex environments, with slow system response and prone to overshoot and steady-state errors.

[0003] As an advanced control method, model predictive control has been widely used in the field of motor control. Model predictive control predicts the future behavior of the system by optimizing the control input under the current state, thereby achieving more precise control. However, traditional model predictive control methods are often limited by the control set and the amount of calculation, especially in multi-phase motor control. Model prediction needs to process a large number of voltage vectors, which increases the amount of calculation and system delay, reducing the real-time performance of the control system.

[0004] At present, although some improvements have been proposed for model predictive control methods for multi-phase motors, such as simplifying the control set and accelerating the calculation strategy, these methods often sacrifice control accuracy and dynamic performance and cannot fully meet the requirements for motor control accuracy and response speed under complex working conditions. Therefore, a new method is urgently needed to improve the dynamic response capability of the system while ensuring control accuracy. Summary of the invention

[0005] In view of the deficiencies in the prior art, the present invention provides a motor speed control system and method to solve the technical problems of low motor control accuracy and slow response speed in the prior art.

[0006] The present invention provides a motor speed control system, comprising: a current acquisition module, a rotational inertia identification module, a load torque observation module, a speed prediction module, a value function evaluation module and a PWM signal generation module;

[0007] The output end of the current acquisition module is respectively connected to the input end of the moment of inertia identification module and the input end of the load torque observation module; the output end of the moment of inertia identification module is respectively connected to the input end of the load torque observation and the input end of the speed prediction module; the output end of the load torque observation module is respectively connected to the input end of the speed prediction module and the input end of the value function evaluation module; the output end of the speed prediction module is connected to the input end of the value function evaluation module; the output end of the value function evaluation module is connected to the input end of the PWM signal generation module; the output end of the PWM signal generation module is connected to the inverter;

[0008] The current acquisition module is used to collect the d and q axis currents of the dual three-phase permanent magnet synchronous motor; the moment of inertia identification module is used to identify the moment of inertia; the load torque observation module is used to identify the load torque; the speed prediction module predicts the speed through the second-order Taylor series; the value function evaluation module selects the optimal voltage vector based on the predicted speed and actual load; the PWM signal generation module generates the inverter drive signal according to the selected optimal voltage vector.

[0009] Furthermore, the motor speed control system further comprises: a load torque feedforward compensation module; the output end of the load torque observation module is also connected to the input end of the load torque feedforward compensation module; the output end of the load torque feedforward compensation module is connected to the input end of the value function evaluation module;

[0010] The load torque feedforward compensation module performs current compensation according to the identified load torque changes.

[0011] Furthermore, the moment of inertia identification module is a model reference adaptive identifier.

[0012] Furthermore, the load torque observation module is a Romberg observer.

[0013] The present invention also provides a control method for a motor speed control system, which is applicable to the above motor speed control system, and the method comprises the following steps:

[0014] Step 1: Obtain the d-axis and q-axis currents of the dual three-phase permanent magnet synchronous motor;

[0015] Step 2: Obtain the moment of inertia based on the d-axis and q-axis currents;

[0016] Step 3: Obtain the load torque based on the moment of inertia;

[0017] Step 4: Predict the speed using the second-order Taylor series according to the moment of inertia and load torque, and obtain the corresponding regulating current according to the predicted speed;

[0018] Step 5: Obtain the minimum voltage vector as the optimal voltage vector through the value function according to the regulated current and the actual load;

[0019] Step 6: Generate a PWM signal based on the optimal voltage vector to control and adjust the motor speed.

[0020] Furthermore, the step 4 also includes compensating the q-axis current in the regulated current.

[0021] Furthermore, in step 2, the moment of inertia is obtained according to the d-axis and q-axis currents through a model reference adaptive identification model, wherein the reference model of the model reference adaptive identification model is:

[0022]

[0023] The adjustable models are:

[0024]

[0025] The deviation output of the reference model from the adjustable model is:

[0026]

[0027] The adaptive algorithm is:

[0028]

[0029] In the formula, are the sampling values ​​of the mechanical angular velocity at time k, k-1, and k-2 respectively; b is in, are the sampling values ​​of the electromagnetic torque at time k-1 and k-2 respectively; “^” is the observed value of the corresponding variable; is the difference between the actual value and the observed value of the mechanical angular velocity at time k; β is the adaptive gain.

[0030] Furthermore, the speed predicted in step 4 obtains the corresponding regulating current as:

[0031]

[0032] In the formula, u d 、u q are dq axis voltage respectively; i d 、i q are dq axis current respectively; L is dq axis inductance; Rs is stator resistance; ω e is the electrical angular velocity; ψ f is the permanent magnet flux; T e is the electromagnetic torque; T L is the load torque; ω m is the mechanical angular velocity; J is the moment of inertia; n p is the pole pair number; Ts For the electrical control cycle.

[0033] Furthermore, the specific formula for compensating the q-axis current in the regulated current in step 4 is:

[0034] In the formula, The q-axis regulating current corresponding to the predicted speed; is the compensation current, T L is the load torque, T e is the actual electromagnetic torque; n p is the pole pair number; ψ f is the permanent magnet flux.

[0035] Furthermore, in step 5, the value function is:

[0036]

[0037] In the formula, and Represent the reference values ​​of d-axis and q-axis currents respectively; and Represent the predicted values ​​of d-axis and q-axis current respectively.

[0038] Beneficial effects of the present invention:

[0039] The present invention can not only effectively improve the dynamic response performance of the motor control system, but also avoid the adjustment complexity brought by the traditional PI controller cascade, making the system more stable and efficient. In addition, through real-time moment of inertia identification, the present invention can accurately adapt to different load changes and ensure the reliability and performance of the motor under complex working conditions. BRIEF DESCRIPTION OF THE DRAWINGS

[0040] The features and advantages of the present invention will be more clearly understood by referring to the accompanying drawings, which are schematic and should not be construed as limiting the present invention in any way. In the accompanying drawings:

[0041] Figure 1 is a system structure block diagram of a specific embodiment of the present invention;

[0042] Figure 2 A topological diagram of a dual three-phase motor drive system according to a specific embodiment of the present invention;

[0043] Figure 3 A distribution diagram of the voltage vector in the αβ plane of a specific embodiment of the present invention;

[0044] Figure 4 A distribution diagram of a voltage vector in an xy plane according to a specific embodiment of the present invention;

[0045] Figure 5FIG. 4 is a distribution diagram of a virtual voltage vector in an αβ plane according to a specific embodiment of the present invention. DETAILED DESCRIPTION

[0046] In order to make the purpose, technical solution and advantages of the embodiments of the present invention clearer, the technical solution in the embodiments of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. Obviously, the described embodiments are part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative work are within the scope of protection of the present invention.

[0047] The present invention is further illustrated below in conjunction with specific embodiments. Those skilled in the art should understand that these embodiments are only used to illustrate the present invention and are not used to limit the scope of the present invention, and modifications to various equivalent forms of the present invention fall within the scope defined by the appended claims of this application.

[0048] like Figure 1 As shown, the present invention provides a motor speed control system, including: a current acquisition module 1, a rotational inertia identification module 2, a load torque observation module 3, a speed prediction module 4, a load torque feedforward compensation module 5, a value function evaluation module 6 and a PWM signal generation module 7;

[0049] The moment of inertia identification module 2 is preferably a model reference adaptive identifier, which can perform online identification of the moment of inertia according to the real-time state information of the motor;

[0050] The load torque observation module 3 is preferably a Romberg observer, which can estimate the torque of the motor by sampling the motor phase current and rotor angle information in real time;

[0051] The output end of the current acquisition module 1 is respectively connected to the input end of the moment of inertia identification module 2 and the input end of the load torque observation module 3; the output end of the moment of inertia identification module 2 is respectively connected to the input end of the load torque observation and the input end of the speed prediction module 4; the output end of the load torque observation module 3 is respectively connected to the input end of the speed prediction module 4, the input end of the load torque feedforward compensation module 5 and the input end of the value function evaluation module 6; the output end of the speed prediction module 4 is connected to the input end of the value function evaluation module 6; the output end of the value function evaluation module 6 is connected to the input end of the PWM signal generation module 7; the output end of the PWM signal generation module 7 is connected to the inverter;

[0052] The current acquisition module 1 is used to collect the d and q axis currents of the dual three-phase permanent magnet synchronous motor; the moment of inertia identification module 2 is used to identify the moment of inertia; the load torque observation module 3 is used to identify the load torque; the speed prediction module 4 predicts the speed through the second-order Taylor series; the output end of the load torque feedforward compensation module 5 is connected to the input end of the value function evaluation module 6; the value function evaluation module 6 selects the optimal voltage vector based on the predicted speed and the actual load; the PWM signal generation module 7 generates a voltage source inverter drive signal according to the selected optimal voltage vector. The dual three-phase motor is driven by a six-phase voltage source inverter. The topology diagram is as shown in FIG. Figure 2 shown.

[0053] The present invention also provides a control method for a motor speed control system, which is applicable to the above motor speed control system, and the method comprises the following steps:

[0054] Step 1: Obtain the d-axis and q-axis currents of the dual three-phase permanent magnet synchronous motor;

[0055] Since the upper and lower switching devices of each bridge arm work in complementary conduction states, each bridge arm has two switching states, and the entire inverter has a total of 2 6 =64 switch states, and the 64 voltage vectors corresponding to the conversion switches are:

[0056]

[0057] In the formula, a=e j30° ;s A ~s F is the switch state of each bridge arm. Set the upper bridge arm to "1" when it is turned on, and "0" when it is turned off. The basic voltage vectors are numbered in the order of ABC and DEF, and the switch state combination is represented in octal, such as Figure 3 and Figure 4 As shown, the distribution of the voltage vector in the αβ plane and the xy plane can be determined.

[0058] like Figure 3 and Figure 4 As shown, each includes 60 effective voltage vectors and 4 zero vectors. These voltage vectors divide the αβ plane and the xy plane into four regular dodecagons with different amplitudes, which are as follows from the inside to the outside: L1 (0.133U dc ), L2(0.333U dc ), L3(0.471U dc ), L4(0.644U dc ).

[0059] Because only the fundamental wave component of the αβ plane participates in the electromechanical energy conversion, the xy plane causes loss, so the virtual voltage vector method is introduced for optimization, then v44 , and v 65 , the directions in the αβ plane are the same, but in the opposite directions in the xy plane. By acting on two vectors in one control cycle and distributing the action time of the two vectors, the effects produced in the xy plane are offset. The synthesis principle is as follows:

[0060]

[0061] Where V L3 and V L4 Represent the voltage vectors of the third and fourth layers respectively; V i is the synthesized virtual voltage vector.

[0062] According to the above principles, 12 virtual voltage vectors with uniform amplitude and phase angle distribution can be constructed, such as Figure 5 As shown, the αβ plane is divided into 12 sectors, and the angle of each sector is 30°.

[0063] The synthesized virtual voltage vector amplitude can be expressed as:

[0064]

[0065] Since the virtual voltage vector is used as the control set of the prediction algorithm, the voltage equation of the harmonic plane can be ignored. The fundamental plane voltage equation of the dual three-phase permanent magnet motor in the rotating coordinate system is:

[0066]

[0067] In the formula, u d ,u q is the dq axis voltage; i d ,i q is the dq axis current; L is the dq axis inductance; R s is the stator resistance; ω e is the electrical angular velocity; ψ f is the permanent magnet flux.

[0068] The torque equation of the dual three-phase motor in the rotating coordinate system:

[0069] T e =3n p i q [i d (L d -L q )+ψ f ]

[0070] Mechanical equations of dual three-phase motor in rotating coordinate system:

[0071]

[0072] According to the d-axis voltage equation, we can get: d The second-order differential equation is:

[0073]

[0074] According to the q-axis voltage equation, we can get: q The second-order differential equation is:

[0075]

[0076] Step 2: Obtain the moment of inertia through the model reference adaptive identification model according to the d-axis and q-axis currents;

[0077] The construction process of the model reference adaptive identification model is as follows:

[0078] Step 21: Ignoring the damping coefficient of the motor, the equation of motion of the motor is:

[0079]

[0080] Step 22: Discretize the equation of motion into a difference equation:

[0081]

[0082] Step 23: Order And subtract the two discretized equations of the difference equation in step 22 to obtain:

[0083]

[0084] Step 24: Use the difference result as the reference model to derive the adjustable model:

[0085]

[0086] The deviation output between the reference model and the adjustable model is recorded as:

[0087]

[0088] Using Landau adaptive law, the adaptive algorithm is designed as follows:

[0089]

[0090] In the formula, is the estimated speed signal; ω m is the speed signal; are the sampling values ​​of the mechanical angular velocity at time k, k-1, and k-2 respectively; b is in, are the sampling values ​​of the electromagnetic torque at time k-1 and k-2 respectively; “^” is the observed value of the corresponding variable; is the difference between the actual value and the observed value of the mechanical angular velocity at time k; β is the adaptive gain.

[0091] The real-time observation value of the moment of inertia can be obtained based on the reference model, the adjustable model and the adaptive algorithm.

[0092] Step 3: Obtain the load torque based on the moment of inertia. The process of constructing the load torque observer is as follows:

[0093] Step 31: According to the mechanical equation of the dual three-phase motor rotating coordinate system, select As the state variable, select u = T e As input, select y = ω m As output, construct the state space equation:

[0094]

[0095] Step 32: The form of the Romberg observer is:

[0096]

[0097] Select according to the form of standard Romberg observer C = [1 0], D is a zero matrix;

[0098] Step 33: Setting up the Matrix According to the form of the Romberg observer, rewriting the state space equation can obtain the load torque observer based on the Romberg form, and the observed value of the load torque can be obtained:

[0099]

[0100] Step 4: Predict the speed through the second-order Taylor series according to the moment of inertia and the load torque, obtain the corresponding regulating current according to the predicted speed, and compensate the q-axis current in the regulating current;

[0101] According to the second-order Taylor series, we can obtain i d The prediction model is:

[0102]

[0103] After sorting, we can get:

[0104]

[0105] According to the second-order Taylor series, we can obtain i q The prediction model is:

[0106]

[0107] The specific formula for compensating the q-axis current in the regulated current is:

[0108]

[0109] In the formula, The q-axis regulating current corresponding to the predicted speed; is the compensation current, T L is the load torque, T e is the actual electromagnetic torque; n p is the pole pair number; ψ f is the permanent magnet flux.

[0110] Step 5: Obtain the minimum voltage vector as the optimal voltage vector through the value function according to the regulated current and the actual load;

[0111] The value function is designed as:

[0112]

[0113] In the formula, and Represent the reference values ​​of d and q axis currents respectively, and Represent the predicted values ​​of d-axis and q-axis current respectively.

[0114] The voltage vector that minimizes g is evaluated as the optimal vector.

[0115] Step 6: The PWM signal generation module generates an inverter drive signal according to the selected optimal voltage vector based on the optimal voltage vector. The voltage source inverter converts the drive signal into the voltage vector required by the motor and drives the dual three-phase permanent magnet synchronous motor to operate.

[0116] Although the embodiments of the present invention have been described in conjunction with the accompanying drawings, those skilled in the art may make various modifications and variations without departing from the spirit and scope of the present invention, and such modifications and variations are all within the scope defined by the appended claims.

Claims

1. A motor speed control system, characterized in that: include: Current acquisition module, moment of inertia identification module, load torque observation module, speed prediction module, value function evaluation module and PWM signal generation module; The output end of the current acquisition module is respectively connected to the input end of the moment of inertia identification module and the input end of the load torque observation module; the output end of the moment of inertia identification module is respectively connected to the input end of the load torque observation and the input end of the speed prediction module; the output end of the load torque observation module is respectively connected to the input end of the speed prediction module and the input end of the value function evaluation module; the output end of the speed prediction module is connected to the input end of the value function evaluation module; the output end of the value function evaluation module is connected to the input end of the PWM signal generation module; the output end of the PWM signal generation module is connected to the inverter; The current acquisition module is used to collect the d and q axis currents of the dual three-phase permanent magnet synchronous motor; the moment of inertia identification module is used to identify the moment of inertia; The load torque observation module is used to identify the load torque; the speed prediction module predicts the speed through the second-order Taylor series; The value function evaluation module selects the optimal voltage vector based on the predicted rotation speed and the actual load; the PWM signal generation module generates the inverter drive signal according to the selected optimal voltage vector.

2. The motor speed control system according to claim 1, characterized in that: The motor speed control system further comprises: a load torque feedforward compensation module; the output end of the load torque observation module is also connected to the input end of the load torque feedforward compensation module; the output end of the load torque feedforward compensation module is connected to the input end of the value function evaluation module; The load torque feedforward compensation module performs current compensation according to the identified load torque changes.

3. The motor speed control system according to claim 1 or 2, characterized in that: The moment of inertia identification module is a model reference adaptive identifier.

4. The motor speed control system according to claim 1 or 2, characterized in that: The load torque observation module is a Romberg observer.

5. A control method for a motor speed control system, applicable to the motor speed control system as claimed in any one of claims 1 to 4, the method comprising the following steps: Step 1: Obtain the d-axis and q-axis currents of the dual three-phase permanent magnet synchronous motor; Step 2: Obtain the moment of inertia based on the d-axis and q-axis currents; Step 3: Obtain the load torque based on the moment of inertia; Step 4: Predict the speed using the second-order Taylor series according to the moment of inertia and load torque, and obtain the corresponding regulating current according to the predicted speed; Step 5: Obtain the minimum voltage vector as the optimal voltage vector through the value function according to the regulated current and the actual load; Step 6: Generate a PWM signal based on the optimal voltage vector to control and adjust the motor speed.

6. The control method of the motor speed control system according to claim 5, characterized in that: In step 2, the moment of inertia is obtained according to the d-axis and q-axis currents through the model reference adaptive identification model, wherein the reference model of the model reference adaptive identification model is: The adjustable models are: The deviation output of the reference model from the adjustable model is: The adaptive algorithm is: In the formula, are the sampling values ​​of the mechanical angular velocity at time k, k-1, and k-2 respectively; b is in, are the sampling values ​​of electromagnetic torque at time k-1 and k-2 respectively; "^" is the observed value of the corresponding variable; is the difference between the actual value and the observed value of the mechanical angular velocity at time k; β is the adaptive gain.

7. The control method of the motor speed control system according to claim 5, characterized in that: The speed predicted in step 4 is used to obtain the corresponding regulating current: In the formula, u d 、u q are dq axis voltage respectively; i d 、i q are dq axis current respectively; L is dq axis inductance; Rs is stator resistance; ω e is the electrical angular velocity; ψ f is the permanent magnet flux; T e is the electromagnetic torque; T L is the load torque; ω m is the mechanical angular velocity; J is the moment of inertia; n p is the pole pair number; T s For the electrical control cycle.

8. The control method of the motor speed control system according to claim 5, characterized in that: The step 4 also includes compensating the q-axis current in the regulated current.

9. The control method of the motor speed control system according to claim 8, characterized in that: The specific formula for compensating the q-axis current in the regulated current in step 4 is: In the formula, The q-axis regulating current corresponding to the predicted speed; is the compensation current, T L is the load torque, T e is the actual electromagnetic torque; n p is the pole pair number; ψ f is the permanent magnet flux.

10. The control method of the motor speed control system according to claim 5, characterized in that: In step 5, the value function is: In the formula, and Represent the reference values ​​of d-axis and q-axis currents respectively; and Represent the predicted values ​​of d-axis and q-axis current respectively.

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