Improved frequency offset compensation method for ultra-wideband system

Abstract

The invention belongs to the field of integrated circuits, and discloses an improved frequency offset compensation method for an ultra wide band (UWB) system, which comprises the following steps of: performing primary compensation on UWB signals through coarse frequency offset estimation; after SYNC detection, loop bandwidth is set according to the SYNC length, and filtering parameters are pre-stored to control the convergence speed of the PLL; using a Cordic algorithm to calculate the signal after carrier stripping to obtain a phase difference; and taking the phase difference as the input of loop filtering, calculating an output correction value, and feeding back the output correction value to the NCO, thereby adjusting the phase and frequency of the local signal, and completing the accurate compensation of the frequency offset of the UWB signal. The problems that in the prior art, when a UWB communication system uses a long SYNC, locking is too slow, and when a short SYNC is used, locking cannot be achieved or tracking precision is too low are solved, and the system is made to adapt to a complex communication environment, especially a high-dynamic channel scene. The performance and the efficiency are considered, and the resource utilization rate and the real-time performance of frequency offset correction are improved.

Description

Technical Field

[0001] The invention belongs to the field of integrated circuits, and in particular relates to an improved frequency offset compensation method for an ultra-wideband system. Background Art

[0002] After the application of Ultra-Wide Band (UWB) technology in handheld devices, its application in smart cars, indoor positioning, product interconnection and other fields is also gradually spreading. Impulse-Radio UWB (IR-UWB) technology uses nanosecond narrowband pulses for data transmission instead of traditional continuous sine waves. It has the advantages of low power consumption, strong anti-interference ability, accurate positioning and high security. These characteristics make impulse UWB technology have considerable development potential in the field of indoor positioning.

[0003] When electromagnetic waves propagate in an indoor environment with complex dimensions and many obstacles, the effects of emission and refraction cause the signal to propagate to the receiving end along multiple paths, and the multipaths arrive at the receiving end at different times, resulting in signal attenuation and inter-code crosstalk. IR-UWB signals propagate indoors in clustered, dense multipath form. Since the duty cycle of the pulse signal is extremely low and the time domain does not overlap, it is easier to separate the multipath components in time. The dense multipath also makes the IR-UWB signal more sensitive in propagation time, thus enabling it to have centimeter-level precise positioning capabilities.

[0004] In practical applications, IR-UWB systems perform positioning through ranging and angle measurement, which are key technologies for achieving high-precision positioning. However, carrier frequency offset (CFO) is the most common problem in IR-UWB systems, which has a great impact on the ranging technology in IR-UWB systems, resulting in ranging errors. In IR-UWB ranging, it is necessary to measure the round-trip propagation time of the signal. The time accuracy in the IR-UWB system is very high. The existence of frequency offset causes errors in the phase and frequency of ranging, which also causes errors in the round-trip propagation time, ultimately leading to ranging errors. In IR-UWB systems, the received signal may drift in phase due to the influence of CFO. This phase error will cause errors in the calculation of the arrival time of the received signal, which will affect the ranging results.

[0005] For a signal with frequency deviation, the phase change after frequency deviation can be expressed as:

[0006] θ(t)=2πΔft+φ0,

[0007] Among them, Δf is the frequency deviation, t is the time, and φ0 is the initial phase; even in a short period of time, the frequency deviation will gradually affect the accumulation of the phase, resulting in a time error in the measurement, which will cause a deviation in the ranging result. The phase error caused by the frequency deviation not only affects the synchronization of the signal, but also directly affects the estimation of the arrival time. Because the estimation of ToA (arrival time) usually depends on the time domain characteristics of the pulse signal, such as the timestamp of the pulse arrival. When the frequency of the signal is offset, the receiver will misjudge the start time or end time of the signal, resulting in a deviation in the distance estimation. The impact of frequency deviation on UWB ranging accuracy can be quantified through a mathematical model, which is usually expressed as the relationship between the frequency deviation Δf and the ranging error d. Assuming that the time resolution of the system is Δt, the ranging error Δd caused by the frequency deviation can be expressed as:

[0008]

[0009] Among them, c is the speed of light and Δt is the signal propagation time difference. This shows that the increase of frequency offset will directly lead to the increase of ranging error, especially in high-speed dynamic scenes, the cumulative effect of error is particularly obvious.

[0010] In the prior art, the wireless receiver usually uses a phase-locked loop (PLL) to track and compensate for frequency deviation. The advantage is that once the PLL is locked, the frequency deviation value can be accurately determined. The existing PLL frequency deviation tracking method mainly multiplies the input signal by the local signal conjugate, and then enters the phase detector to calculate the residual phase after removing the carrier. The loop filter is then used to convert the phase difference into a frequency adjustment value and send it to the numerically controlled oscillator (NCO) to change the output local carrier. Figure 2 The continuous iteration of the process shown makes the frequency and phase of the NCO output signal gradually consistent with the input signal, completing the locking, and the frequency deviation can be corrected.

[0011] However, this method has a slower synchronization speed. Figure 3 As shown in the figure, it is a UWB frame structure diagram, which is mainly used to illustrate the position of the signal synchronization field (SYNC) in the entire communication frame. The horizontal axis represents the symbol sequence in the time axis or signal frame, and shows the time distribution from the frame header to the SYNC field and the subsequent data area. It shows that the frequency deviation tracking and SFD detection will start after the signal synchronization field SYNC detection is completed. Among them, SHR (synchronization header), SFD (frame delimiter), T SYNC Indicates the duration of the SYNC sequence, T SFD Indicates the duration of the start frame delimiter SFD, T PREIndicates the total time of the preamble field. The highest mean pulse frequency (MPRF) of the UWB system specified in the latest IEEE 802.15.4z protocol is 124.8MHz, and the highest data rate is 27.2Mbps; and the preamble sequence used for synchronization is only 16 symbols at the shortest, as shown in Table 1.

[0012] Table 1 Sync sequence specified by IEEE 802.15.4z protocol

[0013]

[0014] Table 1 is used to indicate the number of SYNC sequence symbols corresponding to different SYNC sequence types, that is, the length of the preamble sequence. The longer the SYNC sequence, the higher the reliability of signal detection, and the higher the PLL locking accuracy. The short sequence Short in Table 1 has only 16 symbols, which further increases the difficulty of frequency deviation tracking and correction. When the PLL loop bandwidth is fixed, it is difficult to balance the convergence speed and synchronization accuracy. The traditional method fails to dynamically adjust the bandwidth according to the preamble sequence length (SYNC length). The SYNC sequence is short, and the difficulty of frequency deviation tracking and correction increases. The locking accuracy of PLL is affected by the length of the SYNC sequence, and the accuracy is insufficient under short SYNC sequences. Therefore, it is necessary to improve the traditional PLL frequency deviation tracking method to make it suitable for IR-UWB communication systems. Summary of the invention

[0015] To solve the above technical problems, the present invention provides an improved frequency offset compensation method for an ultra-wideband system, which divides frequency offset tracking into a coarse estimation stage and a fine tracking stage, with the successful detection of SYNC as the boundary. In the coarse estimation stage, the frequency offset value is roughly estimated, and in the fine tracking stage, the frequency offset value is accurately estimated, thereby accelerating the overall processing speed and achieving precision tracking.

[0016] An improved frequency offset compensation method for an ultra-wideband system according to the present invention comprises the following steps:

[0017] Step 1: For the UWB signal received at the current moment, the real part and the imaginary part are used to calculate the current carrier frequency deviation value, and the UWB signal is initially compensated based on the value of the frequency deviation;

[0018] Step 2: Perform SYNC detection on the UWB signal after the initial compensation to determine the synchronization state; after the detection is completed, set the loop bandwidth according to the SYNC length and pre-store the filter parameters to control the convergence speed of the PLL;

[0019] Step 3: Process the sinusoidal signal sampling values ​​pre-stored in the NCO to obtain the required local signal, conjugate and multiply the local signal with the input signal to complete carrier stripping;

[0020] Step 4: Use the Cordic algorithm to calculate the signal after stripping the carrier to obtain the phase difference;

[0021] Step 5: Use the phase difference obtained in step 4 as the input of the loop filter, calculate the output correction value, and feed it back to the NCO to adjust the phase and frequency of the local signal and complete the precise frequency deviation compensation of the UWB signal.

[0022] Furthermore, step 1 is specifically as follows:

[0023] For the UWB signal S(t) received at the current moment:

[0024] S(t)=I(t)+jQ(t)

[0025] Where t is the current phase, I and Q are the real and imaginary parts of the signal, and j is the imaginary unit; expanded using Euler's formula, it is:

[0026]

[0027] Where R(t) is the modulus of S(t), is the phase angle of 0 or π, and θ0 is the initial phase at the receiving end. Since the UWB signal is a spread spectrum signal, the despread signal is expressed as:

[0028]

[0029] Among them, Δω is the phase difference caused by the frequency deviation of the transmitting and receiving carriers, Or I, θ0 is the initial phase at the receiving end, and n(t) is the zero-mean Gaussian noise generated by the Gaussian channel;

[0030] In order to obtain Δω, the two segments of data obtained by intercepting the received UWB signal are conjugate multiplied for the two adjacent segments of data r(k) and r(k-1) at time k. d(k) represents the signal at the current time k, and r*(k-1) is the complex conjugate of r(k-1), which represents the conjugate form of the signal at the previous time:

[0031] d(k)=r(k)×r * (k-1),

[0032] The phase of d(k) represents the value of the current carrier frequency deviation. To increase accuracy, multiple d(k) are averaged to calculate the frequency deviation Δf:

[0033]

[0034] Among them, Im(d i ) and Re(d i ) is the imaginary part and the real part of the sum of multiple d(k), T is the symbol period after despreading; the signal is initially compensated using the coarse estimated frequency offset value obtained by calculation.

[0035] Furthermore, step 2 is specifically as follows:

[0036] In a second-order PLL system, the loop bandwidth BW is:

[0037]

[0038] Where ωn is the natural frequency of the system, ζ is the damping coefficient, and the loop locking time t L have:

[0039]

[0040] Substituting into BW we have:

[0041]

[0042] After the loop bandwidth is determined, the parameters of the second-order loop filter are calculated. The filter coefficients C0 and C1 are controlled so that the PLL algorithm can adjust the phase error.

[0043]

[0044] Among them, T s is the sampling period of a single SYNC symbol, which is determined by the length of the Preamble sequence, and K is the loop gain. To save computing time and system resources, the C0 and C1 to be used are calculated and stored in advance, and can be looked up in the table according to the current loop bandwidth during communication.

[0045] Furthermore, step 3 is specifically as follows:

[0046] A sampled value of a cosine function in [0, I / 2] is pre-stored in the NCO. The sampling phase index is calculated according to the input phase value. The corresponding phase is searched in the sampling result to obtain the sine and cosine values. Finally, the sign and position of the orthogonal component are adjusted according to the quadrant where the input phase is located to obtain the required local signal.

[0047] The I and Q signals within a certain phase interval are accumulated to obtain the input signal Z = S I +jS Q , where S I , S Q are the accumulated results of I and Q signals respectively, j is the imaginary unit; the rotated signal Z′=Z·e jθ , for e jθExpanding using Euler's formula, we get:

[0048] Z′=(S I +jS Q )·(cosθ+jsinθ)

[0049] Expand to:

[0050] Z′=(S I cosθ-S Q sinθ)+j(S I sinθ+S Q cosθ)

[0051] Z′ is the signal after stripping the carrier, and cosθ and sinθ are the outputs of NCO, S I ′=S I cosθ-S Q sinθ and S Q ′=S I sinθ+Qcosθ are the new real and imaginary parts.

[0052] In the loop iteration, the input phase of the NCO is the output of the loop filter, and the NCO applies this correction value to the previous changing frequency; therefore, if the phase in the signal is ahead of the local, the frequency of the NCO will increase; if the phase in the signal lags behind the local, the frequency of the NCO will decrease. The essence of the NCO output is the residual carrier phase in the target signal.

[0053] Furthermore, step 4 is specifically as follows:

[0054] Suppose a vector (x i ,y i ), for the vector (x i ,y i ) applies a rotation of angle θ0, by y i Determine the direction of rotation a i :

[0055]

[0056] The rotated vector:

[0057]

[0058] tanθ0 is the sine value after the first rotation, tanθ n is the sine value after rotation n times, tanθ0 and tanθ n The value of is pre-calculated and stored in the lookup table. The rotation angle is accumulated after each rotation. This method is used to calculate the vector (S I , S Q) is iterated for a preset number of times to obtain a phase difference Δθ.

[0059] Furthermore, step 5 is specifically as follows:

[0060] The function of the loop filter is to receive the phase difference Δθ, and after processing, output the correction value to interfere with the NCO, thereby changing the local signal; by controlling the Δθ ratio (i.e., C0 and C1) and its accumulated value, the loop filter makes the output phase smooth.

[0061] The loop filter output at time t is:

[0062]

[0063] Among them, S p [t] is the accumulated phase compensation value up to time t, and is also the input phase of the NCO. From this point on, steps 1 to 4 form a closed loop, which can be repeated to complete the frequency deviation correction of the UWB signal.

[0064] The beneficial effects described in the present invention are as follows: the present invention improves the existing frequency offset compensation method for the UWB communication system, and solves the problem that the existing technology is too slow to lock when using long SYNC, and cannot lock or the tracking accuracy is too low when using short SYNC in the UWB communication system through a coarse-first-fine tracking method. The method described in the present invention uses a coarse frequency offset estimation method of symbol accumulation to perform coarse frequency offset estimation in advance in the SYNC detection stage, which greatly shortens the convergence time of the PLL; the dynamically adjusted PLL parameter optimization mechanism enables the PLL to quickly adapt to high-frequency signals under short SYNC sequences. The present invention can improve the frequency offset tracking accuracy, the symbol accumulation method enhances the noise suppression capability, and can still achieve high-precision frequency offset correction under short SYNC conditions; for long SYNC sequences, the locking accuracy is improved by narrow bandwidth, which is suitable for high-precision synchronization scenarios. The present invention can still achieve efficient frequency offset tracking and correction under the conditions of a pulse frequency of 124.8MHz and a data rate of 27.2Mbps. The dynamic adjustment mechanism enables the system to adapt to complex communication environments, especially high-dynamic channel scenarios. Taking into account both performance and efficiency, the PLL parameter settings are optimized through a table lookup mechanism, which reduces the real-time calculation burden and improves the resource utilization and real-time performance of frequency offset correction. BRIEF DESCRIPTION OF THE DRAWINGS

[0065] Figure 1 is a flow chart of the method of the present invention;

[0066] Figure 2 It is the way the protocol specifies how the PLL algorithm handles the residual carrier;

[0067] Figure 3 This is the frame structure diagram of UWB;

[0068] Figure 4 It is local signal generation and carrier stripping;

[0069] Figure 5 It is the original frequency offset correction result with a preamble length of 31;

[0070] Figure 6 This is the frequency offset correction result after the preamble length is improved to 31;

[0071] Figure 7 It is the original frequency offset correction result with a preamble length of 127;

[0072] Figure 8 It is the result of frequency offset correction after the preamble length is improved to 127. DETAILED DESCRIPTION

[0073] In order to make the contents of the present invention more clearly understood, the present invention is further described in detail below based on specific embodiments in conjunction with the accompanying drawings.

[0074] The present invention provides an improved frequency offset compensation method for ultra-wideband system, based on the UWB signal format specified in the IEEE 802.15.4z protocol, whose signal synchronization field (SYNC) is a fixed length sequence composed of ternary codes {-1, 0, 1} that is repeated continuously, as shown in Table 2, the data defines the format and characteristics of the synchronization preamble sequence for ultra-wideband system (UWB), which is shown as a short sequence and a long sequence, and these data provide an important basis for subsequent frequency offset tracking and correction, and the specific sequence and its repetition times before synchronization are known at the transceiver. Using this characteristic, the frequency offset tracking compensation process can be divided into two parts.

[0075] Table 2 Preamble sequence specified by IEEE 802.15.4z protocol

[0076]

[0077] The frequency deviation tracking is divided into a coarse estimation stage and a fine tracking stage, with the successful detection of SYNC as the boundary. In the coarse estimation stage, the frequency deviation value is roughly estimated, and in the fine tracking stage, the frequency deviation value is accurately estimated, which speeds up the overall processing speed and achieves accurate tracking.

[0078] like Figure 1 As shown, the improved frequency offset compensation method for an ultra-wideband system according to the present invention comprises the following steps:

[0079] Step 1: For the UWB signal received at the current moment, the real part and the imaginary part are used to calculate the current carrier frequency deviation value, and the UWB signal is initially compensated based on the frequency deviation value to reduce the influence of the signal frequency deviation on subsequent processing and ensure the preliminary conditions for synchronous detection and PLL operation;

[0080] Step 2: Perform SYNC detection on the UWB signal after the initial compensation to determine the synchronization state; after the detection is completed, set the loop bandwidth according to the SYNC length and pre-store the filter parameters to control the convergence speed of the PLL; pre-store the PLL parameters corresponding to the loop bandwidth to support the subsequent steps to quickly obtain through table lookup;

[0081] Step 3: After scaling and flipping the sine signal sampling values ​​pre-stored in the NCO, the required local signal is obtained, and the local signal is conjugate-multiplied with the input signal to complete carrier stripping;

[0082] Step 4: Use the Cordic algorithm to calculate the signal after stripping the carrier to obtain the phase difference;

[0083] Step 5: Use the phase difference obtained in step 4 as the input of the loop filter, calculate the output correction value, and feed it back to the NCO to adjust the phase and frequency of the local signal; by controlling the ratio and its accumulated value, the loop filter makes the output phase smooth and gradually completes the precise frequency deviation compensation of the UWB signal.

[0084] Among them, in step 1, the signals received at the current moment are:

[0085] S(t)=I(t)+jQ(t),

[0086] Where t is the current phase, I and Q are the real and imaginary parts of the signal, and j is the imaginary unit. Expand using Euler's formula:

[0087]

[0088] in Or π, θ0 is the initial phase. Since the UWB signal is a spread spectrum signal, the despread signal can be expressed as:

[0089]

[0090] Where Δω is the phase difference caused by the frequency deviation of the transmitting and receiving carriers, Or I, θ0 is the initial phase at the receiving end, and n(t) is the zero-mean Gaussian noise generated by the Gaussian channel.

[0091] In order to obtain Δω, conjugate multiplication is performed on the two adjacent data segments r(k) and r(k-1) at time k:

[0092] d(k)=r(k)×r * (k-1),

[0093] The phase of d(k) represents the value of the current carrier frequency deviation. To increase accuracy, an appropriate number of d(k) can be averaged to calculate the frequency deviation:

[0094]

[0095] Where Im(d i ) and Re(d i ) is the sum of the imaginary and real parts, and T is the symbol period after despreading. The signal is initially compensated using the coarse estimated frequency offset value obtained by calculation.

[0096] In step 2, SYNC detection is completed and the fine tracking phase begins. First, parameter setting is performed. Offset correction is performed after SYNC detection is completed, so the convergence speed of the PLL should be determined by the length of SYNC. As shown in Table 2, corresponding to different preamble sequence lengths (N sync ) Select different loop bandwidths to control the convergence speed. In a second-order PLL system, the loop bandwidth:

[0097]

[0098] Where BW is the loop bandwidth, ωn is the system natural frequency, ζ is the damping coefficient, and the loop locking time t L have:

[0099]

[0100] Substituting into BW we have:

[0101]

[0102] It is easy to obtain that when the damping coefficient is fixed, the locking time t L Inversely proportional to the loop bandwidth BW, increasing the loop bandwidth will reduce the locking time, but will increase the impact of noise. The loop bandwidth corresponding to different Sync lengths is determined based on calculation and simulation results and pre-stored at the receiving end.

[0103] After the loop bandwidth is determined, the parameters of the second-order loop filter need to be calculated. The filter coefficients C0 and C1 are controlled so that the PLL can properly adjust the phase error.

[0104]

[0105] Where Ts is the sampling period of a single SYNC symbol, which is determined by the length of the Preamble sequence, and K is the loop gain. To save computing time and system resources, the required C0 and C1 are calculated and stored in advance, and can be looked up in the table according to the current loop bandwidth during communication.

[0106] In step 3, the local signal generation is separated from the carrier. A sampled value of a cosine function in [0, I / 2] is pre-stored in the NCO. By scaling and flipping it, the cosine and sine values ​​of any phase can be obtained, such as Figure 4As shown, the horizontal axis is the sampling point used to locate the corresponding amplitude of the input phase in the interval [0, I / 2], and the vertical axis is the amplitude representing the amplitude value of the cosine function at the corresponding phase. Through the combination of the horizontal and vertical axes, the corresponding amplitude can be quickly found when the phase value is input.

[0107] The sampling phase index is calculated based on the input phase value (initial value is 0). Figure 4 The corresponding phase is found in the sampling results shown to obtain the sine and cosine values, and finally the sign and position of the orthogonal component are adjusted according to the quadrant where the input phase is located to obtain the desired local signal. In the loop iteration, the input phase of the NCO is the output of the loop filter, and the NCO uses this correction value for the previous changing frequency. Therefore, if the phase in the signal is ahead of the local, the frequency of the NCO will increase; if the phase in the signal lags behind the local, the frequency of the NCO will decrease. The essence of the NCO output is the residual carrier phase in the target signal. It can be seen that carrier stripping is essentially adding a factor of e to the input signal. jθ The I and Q signals within a certain phase interval are accumulated to obtain the input signal Z = S I +jS Q , where S1, S Q are the accumulated results of I and Q signals respectively, and j is an imaginary unit. The rotated signal Z′=Z·e jθ , for e jθ Expanding using Euler's formula, we get:

[0108] Z′=(S I +jS Q )·(cosθ+jsinθ)

[0109] Expand to:

[0110] Z′=(S I cosθ-S Q sinθ)+j(S I sinθ+S Q cosθ)

[0111] Z′ is the signal after stripping the carrier, and cosθ and sinθ are the outputs of NCO, S I ′=S I cosθ-S Q sinθ and S Q ′=S I sinθ+Qcosθ are the new real and imaginary parts.

[0112] Step 4, calculate the phase difference. It is difficult to directly calculate θ = arg(Z′), so the Cordic algorithm is used to calculate the real and imaginary parts of Z′ to obtain the phase. The Cordic algorithm rotates a two-dimensional vector to the real axis step by step through iterative rotation, while accumulating these rotation angles; the final accumulated angle is the phase of the complex number.

[0113] Suppose a vector (x i ,y i ), for the vector (x i ,y i ) applies a rotation of angle θ0, by y i Determine the direction of rotation a i :

[0114]

[0115] The rotated vector:

[0116]

[0117] The value of tanθ0 and tanθ after rotation n times n The value of is pre-calculated and stored in the lookup table. The rotation angle is accumulated after each rotation. This method is used to calculate the vector (S I , S Q ) is iterated a certain number of times to obtain the phase difference Δθ.

[0118] Step 5: Loop filtering. The function of the loop filter is to receive the phase difference Δθ, and after processing, output the correction value to interfere with the NCO, thereby changing the local signal. By controlling the Δθ ratio and its accumulated value, the loop filter makes the output phase smooth. The loop filter output at time t is:

[0119]

[0120] Among them, S p [t] is the accumulated phase compensation value up to time t, and is also the input phase of the NCO. From this point on, steps 1 to 4 form a closed loop, which can be repeated to complete the frequency deviation correction of the UWB signal.

[0121] This improved frequency offset compensation method is added to the receiving end of the UWB communication system for testing.

[0122] like Figure 5 and Figure 6 As shown, compared with the frequency offset compensation method before improvement, the method of the present invention has the same transmission and reception configuration of data packets with the preamble length of 31. Figure 5In the case where the frequency offset compensation is not improved, the sampling iteration number does not converge until 200 times, while the method of the present invention roughly converges when the iteration number is about 150, and has obvious convergence behavior compared with the sampling number without improvement. Under the same sampling number, the frequency offset correction behavior after improvement is significantly improved.

[0123] like Figure 7 and Figure 8 As shown, when the data packet receiving and sending configurations are consistent and the preamble length is 127, the method of the present invention is compared with the frequency offset compensation method before improvement. Before improvement, the PLL algorithm is locked after about 250 iterations and the frequency offset correction is performed. After improvement, the algorithm performs coarse convergence frequency offset correction after 100 iterations after signal reception. It is shown that SYNC synchronization is successful after 200 iterations, and fine convergence is performed, and the frequency offset correction is successful. In summary, the improved algorithm has a significant performance improvement in frequency offset correction.

[0124] The above description is only a preferred embodiment of the present invention and is not intended to be a further limitation of the present invention. All equivalent changes made using the contents of the present specification and drawings are within the protection scope of the present invention.

Claims

1. An improved frequency offset compensation method for ultra-wideband system, characterized in that: The following steps are involved: Step 1: For the UWB signal received at the current moment, the real part and the imaginary part are used to calculate the current carrier frequency deviation value, and the UWB signal is initially compensated based on the value of the frequency deviation; Step 2: Perform SYNC detection on the UWB signal after the initial compensation to determine the synchronization state; After the detection is completed, the loop bandwidth is set according to the SYNC length, and the filter parameters are pre-stored to control the convergence speed of the PLL; Step 3: Process the sinusoidal signal sampling values ​​pre-stored in the NCO to obtain the required local signal, conjugate and multiply the local signal with the input signal to complete carrier stripping; Step 4: Use the Cordic algorithm to calculate the signal after stripping the carrier to obtain the phase difference; Step 5: Use the phase difference obtained in step 4 as the input of the loop filter, calculate the output correction value, and feed it back to the NCO to adjust the phase and frequency of the local signal and complete the precise frequency deviation compensation of the UWB signal.

2. The improved frequency offset compensation method for ultra-wideband system according to claim 1, characterized in that: Step 1 is as follows: For the UWB signal S(t) received at the current moment: S(t)=I(t)+jQ(t) Where t is the current phase, I and Q are the real and imaginary parts of the signal, and j is the imaginary unit; expanded using Euler's formula, it is: Where R(t) is the modulus of S(t), is the phase angle of 0 or I, θ0 is the initial phase of the receiving end; since the UWB signal is a spread spectrum signal, the despread signal is expressed as: Where Δω is the phase difference caused by the frequency deviation of the transmitting and receiving carriers, and n(t) is the zero-mean Gaussian noise generated by the Gaussian channel; The two segments of data received by intercepting the received UWB signal are conjugate multiplied for the two adjacent segments of data r(k) and r(k-1) at time k; d(k) is the signal value at the current time k, r * (k-1) is the complex conjugate of r(k-1), which represents the conjugate form of the signal at the previous moment: d(k)=r(k)×r * (k-1), The phase of d(k) represents the value of the current carrier frequency deviation. To increase accuracy, multiple d(k) are averaged to calculate the frequency deviation Δf: Among them, Im(d i ) and Re(d i ) is the imaginary part and the real part of the sum of multiple d(k), T is the symbol period after despreading; the signal is initially compensated using the coarse estimated frequency offset value obtained by calculation.

3. The improved frequency offset compensation method for ultra-wideband system according to claim 1, characterized in that: Step 2 is as follows: In a second-order PLL system, the loop bandwidth BW is: Among them, ω n is the natural frequency of the system, ζ is the damping coefficient; for the loop locking time tL: Substituting into BW we have: After the loop bandwidth is determined, the parameters of the second-order loop filter are calculated. The filter coefficients C0 and C1 are controlled so that the PLL algorithm can adjust the phase error. Among them, T s is the sampling period of a single SYNC symbol, which is determined by the length of the Preamble sequence, and K is the loop gain.

4. The improved frequency offset compensation method for ultra-wideband system according to claim 1, characterized in that: Step 3 is as follows: A sampled value of a cosine function in [0, I / 2] is pre-stored in the NCO. The sampling phase index is calculated according to the input phase value. The corresponding phase is searched in the sampling result to obtain the sine and cosine values. Finally, the sign and position of the orthogonal component are adjusted according to the quadrant where the input phase is located to obtain the required local signal. The I and Q signals within a certain phase interval are accumulated to obtain the input signal Z = S I +jS Q , where S I , S Q are the accumulated results of I and Q signals respectively, j is the imaginary unit; the rotated signal right Expanding using Euler's formula, we get: Z′=(S I +jS Q )·(cosθ+jsinθ) Expand to: Z′=(S I cosθ-S Q sinθ)+j(S I sinθ+S Q cosθ) Among them, Z′ is the signal after stripping the carrier, cosθ and sinθ are the outputs of NCO, S I ′=S I cosθ-S Q sinθ and S Q ′=S I sinθ+Qcosθ are the new real and imaginary parts.

5. The improved frequency offset compensation method for ultra-wideband system according to claim 1, characterized in that: Step 4 is as follows: Suppose a vector (x i ,y i ), for the vector (x i ,y i ) applies a rotation of angle θ0, by y i Determine the direction of rotation a i : The rotated vector: tanθ0 is the sine value after the first rotation, tanθ n is the sine value after rotation n times, tanθ0 and tanθ n The value of is pre-calculated and stored in the lookup table. The rotation angle is accumulated after each rotation. This method is used to calculate the vector (S I , S Q ) is iterated for a preset number of times to obtain a phase difference Δθ.

6. The improved frequency offset compensation method for ultra-wideband system according to claim 5, characterized in that: Step 5 is as follows: The loop filter output at time t is: Among them, S p [t] is the accumulated phase compensation value up to time t, and is also the input phase of the NCO.

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