Direct-current converter with low input current ripple and high voltage gain

Abstract

In order to solve the problems of large input current ripple and low output voltage of a battery in new energy grid connection, the invention discloses a low-input current ripple and high-voltage gain Boost converter adopting a three-winding coupling inductor. The converter is based on the topology of a secondary cascade Boost circuit, is combined with a three-winding coupling inductor, a tap inductor, a switched capacitor and a passive clamping structure, and specifically comprises eight diodes, eight capacitors, a metal oxide semiconductor (MOS) tube, a tap inductor, a three-winding coupling inductor and an inductor. A current ripple offset structure is arranged at the input end of the converter, a three-winding coupling inductor and two voltage multiplication structures are arranged in the middle of the converter, and a cascade structure enables voltage gain to be multiplied. In addition, the series connection of output capacitors and a passive clamping structure greatly reduce the voltage stress of a switching tube and improve the efficiency of the converter. The whole converter has the advantages of being low in input current ripple, low in switching voltage stress, small in duty ratio and small in coupling inductor turn ratio while achieving extremely high voltage gain.

Description

Technical Field

[0001] The low input current ripple and high voltage gain involved in the present invention belong to the field of power electronics, and specifically relate to a cascade boost converter structure using a three-winding coupled inductor. Background Art

[0002] Energy, as the engine of economic development, drives various industries and transportation systems, provides people with basic living needs, and supports the rapid development of information technology and the digital age. Transportation, industrial manufacturing, agricultural production, and the increasingly digital social structure all rely on reliable and sufficient energy supply. With the continuous development of human society and the accelerated advancement of industrialization, people's demand for energy has exploded. At present, most of the energy still comes from non-renewable fossil fuels such as coal, oil and natural gas, making the world face a serious energy shortage problem. In addition, the excessive exploitation and use of traditional fossil fuels not only accelerates the greenhouse effect and causes climate change, but also causes serious environmental damage. This makes us more urgent to develop and exploit renewable energy. Among them, photovoltaic power generation, as a clean and renewable energy technology, has multiple advantages such as no need for fuel, no emission of pollutants, and low cost. It is one of the most popular and promising industries today.

[0003] The output voltage of renewable energy power generation is usually low, which is difficult to meet the input requirements of the grid-connected system. For example, the output voltage range of a single photovoltaic module is 20-45V, which needs to be increased to 380-400V by a boost converter before it can be transmitted to the subsequent grid-connected inverter. In the renewable energy power generation system, the input current ripple not only affects the normal operation and service life of the power generation module, but also causes EMI and causes additional power loss, resulting in a decrease in the efficiency of the entire system. Therefore, a DC-DC converter with low input current ripple, high voltage gain, and high efficiency is an indispensable part of the renewable energy power generation system. Traditional Boost converters have the disadvantages of low voltage gain, large input current ripple, large switching voltage stress, and low efficiency, and can no longer meet today's application requirements. Summary of the invention

[0004] In view of the deficiencies in the prior art, the present invention provides a DC-DC converter structure with low input current ripple and high voltage gain, which has the advantages of low input current ripple, high voltage gain and low switch voltage stress, and meets the requirements of boost application in grid-connected systems.

[0005] In order to achieve the above object, the present invention adopts the following technical solution:

[0006] (1) Use ripple cancellation structure: Add a ripple cancellation structure on the input side. The ripple cancellation structure consists of a tapped inductor, an auxiliary inductor and an auxiliary capacitor. The two windings of the tapped inductor are coupled to each other. When the current in one inductor changes, its magnetic field will be transferred to the inductor coupled with it, causing the current in the other inductor to change. The tap end of the tapped inductor is connected in series with the auxiliary inductor and capacitor. When the circuit is working, the energy between the auxiliary inductor and the auxiliary capacitor is transferred to each other, and a current waveform passing through the origin can be formed in this branch. When the converter circuit is working, the ripple of the input current, the mutual inductance current of the tapped inductor and the auxiliary current of the branch cancel each other out, which can reduce the overall input current ripple.

[0007] (2) Use a two-stage boost stack topology: Cascade two boost converters and integrate two active switch tubes through circuit transformation to form a two-stage boost converter. After cascading, the voltage gain of the converter doubles, and the complexity of the control circuit and the stability of the system remain unchanged, but the voltage stress of the switch tube is increased. To this end, the output of the two-stage boost converter and the flyback converter are connected in series, and the main switch and coupling inductor are shared, which further improves the voltage gain. The output is connected in series with multiple capacitors, which reduces the voltage stress of the switch tube.

[0008] (3) Use voltage multiplication structure: The voltage multiplication circuit is an efficient, low-cost and simple topology structure that can be constructed using inductors, capacitors and diodes. The voltage multiplication module is placed in the circuit after the main switch. It stores and releases energy as the main switch is turned on and off, which can improve the voltage gain of the converter. There are two voltage multiplication modules, one added to the middle side of the converter and the other added to the output side of the converter.

[0009] (4) Use a passive clamping structure: A passive clamping structure can be created using only a diode and a capacitor. This can greatly reduce the voltage stress of the switch tube, recover the energy of the leakage inductance and release it to the output capacitor, further improving the voltage gain and efficiency of the converter.

[0010] (5) Use a three-winding coupled inductor: The inductor in the converter is coupled with the inductors in the two voltage multiplier structures to form a three-winding coupled inductor. By adjusting the turns ratio, the voltage gain of the converter can be greatly increased.

[0011] Compared with the prior art, the present invention has the following advantages and beneficial effects:

[0012] The ripple cancellation structure on the input side makes the converter have very low input current ripple, which can easily realize more accurate maximum power point tracking control of the module integrated circuit that collects photovoltaic energy. And it is relatively simple to achieve zero ripple, just increase the auxiliary inductor L rThe value of can be set to be equal to the mutual inductance M of the tap inductor. The voltage multiplier and the three-winding coupled inductor are used at the same time, so that the circuit structure can achieve high voltage gain at a lower duty cycle. In addition, the voltage stress of the switch tube is low, and a switch tube with lower on-resistance can be selected, which can effectively reduce the conduction loss of the switch. In addition, the voltage stress of each device can be flexibly adjusted by adjusting the three-winding coupled inductor. BRIEF DESCRIPTION OF THE DRAWINGS

[0013] In order to more clearly illustrate the technical solutions of the exemplary embodiments of the present invention, the following briefly introduces the drawings required for use in the embodiments. It should be understood that the following drawings only illustrate certain embodiments of the present invention and should not be regarded as limiting the scope. For those of ordinary skill in the art, other relevant drawings can be obtained based on these drawings without creative work. In the drawings:

[0014] Figure 1 is a schematic diagram of a low input current ripple high voltage gain converter in an embodiment of the present invention;

[0015] Figure 2 is an equivalent structural diagram of a low input current ripple high voltage gain converter in an embodiment of the present invention;

[0016] Figure 3 is a voltage and current waveform diagram of a converter in an embodiment of the present invention;

[0017] Figure 4 The circuit working mode I (t0-t1) in the embodiment of the present invention;

[0018] Figure 5 The circuit working mode II (t1-t2) in the embodiment of the present invention;

[0019] Figure 6 The circuit working mode III (t2-t3) in the embodiment of the present invention;

[0020] Figure 7 The circuit working mode IV (t3-t4) in the embodiment of the present invention;

[0021] Figure 8 is the circuit working mode V (t4-t5) in the embodiment of the present invention;

[0022] Fig. 9 The circuit working mode VI (t5-t6) in the embodiment of the present invention;

[0023] Fig.10 is a curve showing the relationship between the voltage gain and the duty cycle of the converter in the embodiment of the present invention;

[0024] Fig.11 is a curve showing the relationship between the voltage stress of the switch tube and the duty cycle in the embodiment of the present invention;

[0025] Fig.12 is a curve showing the relationship between the switch tube current stress and the duty cycle in the embodiment of the present invention;

[0026] Fig.13 The converter input current waveform in the embodiment of the present invention;

[0027] Fig.14 is the converter output voltage waveform in the embodiment of the present invention;

[0028] Fig.15 This is a physical verification diagram in an embodiment of the present invention. DETAILED DESCRIPTION

[0029] In order to make the objectives, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below in conjunction with embodiments and drawings. The exemplary embodiments of the present invention and their description are only used to explain the present invention and are not intended to limit the present invention.

[0030] (I) Converter topology analysis

[0031] The converter schematic diagram of the present invention is as follows: Figure 1 As shown, the circuit modularization includes an input side, an intermediate side and an output side; the input side includes a DC voltage source V in , tap inductance L A , L B , auxiliary inductor L r and auxiliary capacitor C r The middle side includes diodes D1 to D5, coupled inductor M1, capacitors C1, C2, C3, C S , switch tube S; the output side includes diodes D6, D7, D8, output capacitor C o1 , C o2 , C o3 , resistor R o Among them: N1, N2, N3 are three windings of coupled inductor; L A , L B They are a pair of tapped inductors, their mutual inductance is M, the tapped part and the auxiliary inductor L r 、Auxiliary capacitor C r connected in series to form a ripple cancellation structure; the voltage source V in , tapped inductor, diodes D1, D2, capacitor C1 and switch tube S together form a primary boost circuit, voltage source V in The tapped inductor first charges capacitor C1, and then capacitor C1 serves as the input source of the secondary boost circuit to transfer energy to the output capacitor C o1;Diode D3 and capacitor C S The passive clamping structure can not only suppress the voltage spike when the switch tube S is turned off, reduce the voltage stress of the switch tube S, but also recover the energy of the leakage inductance and improve the efficiency of the converter; diodes D3, D4, D5, capacitor C S , C2, C3 and coupled inductor secondary winding N2 form voltage multiplier module 1; diodes D7, D8, output capacitor C o2 , C o3 The voltage multiplier module 2 is formed with the coupled inductor secondary winding N3; the capacitor C1 and the coupled inductor primary winding n1 are used as input terminals, and together with all subsequent devices, a secondary boost circuit is formed.

[0032] The equivalent circuit diagram of the converter designed by the present invention is as follows: Figure 2 As shown, L m is the equivalent excitation inductance of the three-winding coupled inductor, L k1 , L k2 , L k3 The tapped inductance can be decoupled and equivalent to three single inductors. The equivalent inductance of the tapped end and the auxiliary inductance L r Combined, we finally get inductors L1, L2, and L3. The equivalent values ​​of the three inductors are:

[0033]

[0034] (II) Converter operating mode analysis

[0035] To further illustrate the basic characteristics and working principle of a low input current ripple and high voltage gain DC-DC converter, this embodiment performs modal analysis on a Boost-based low input current ripple and high voltage gain converter. To simplify the circuit analysis, the following assumptions are made:

[0036] (1) The converter operates in continuous current mode (CCM);

[0037] (2) All capacitors are large enough that the output voltage of these capacitors can be considered constant in steady state;

[0038] (3) All diodes in the converter are ideal, and the switching tube only considers the effect of its parasitic capacitance on transient state;

[0039] (4) Since the leakage inductance is much smaller than the excitation inductance, k is assumed to be 1 during calculation, and the turns ratio of the coupled inductor is N1:N2:N3=1:n2:n3.

[0040] The converter has 6 modes in one cycle, namely Mode I to VI, among which Mode II, III, V, and IV are the main working modes. The working waveforms of the key components of the converter in one cycle are as follows: Figure 3As shown, the equivalent circuits of its six switching modes are as follows Figures 4 to 9 This paper analyzes the working principle of the converter in CCM as follows:

[0041] 1) Mode I [t0~t1]: At t0, the switch S starts to close, and the equivalent circuit of the converter is as follows: Figure 4 As shown. At this time, diodes D1, D6, and D8 are turned on, and diodes D2, D3, D4, D5, and D7 are turned off. Input voltage source V in The inductor L2 is charged through D1 and S, and the current i L2 The inductor L3 starts to increase linearly to the auxiliary capacitor C r Discharge, current i L3 The linear decrease offsets the current i L2 The current ripple generated by the rise makes the current i L1 Basically remain unchanged. Capacitor C1 starts to supply excitation inductance L m Charging, current i Lm 、i Lk1 Linear increase. Due to the existence of leakage inductance, the current flowing through the three windings N1, N2, and N3 of the coupled inductor cannot change suddenly, and the leakage inductance discharge causes the diodes D6 and D8 to continue to conduct. Excitation inductance L m The voltage across N1 changes from discharge state to charge state, resulting in a voltage v Lm The polarity of is reversed, which in turn changes the voltage polarity across N2 and N3, accelerating the leakage inductance i Lk2 、i Lk3 The discharge process makes the current i D6 、i D8 Rapidly decreases. When the current i Lm with i Lk1 When they are equal, the current flowing through the windings N1, N2, and N3 is 0, and the circuit enters the next mode, which is a short transition state.

[0042] 2) Mode II [t1~t2]: At t1, the switch S remains closed, and the equivalent circuit of the converter is as follows: Figure 5 At this time, the leakage inductance L k2 , L k3 The energy is released, diodes D6 and D8 are turned off, and diodes D4 and D7 are turned on. Capacitor C3 stops transferring energy to the output side, and capacitor C1 continues to transfer energy to the excitation inductor L. m Charge, and transfer part of the energy to N2 and N3 through the coupling inductor N1. N2 and capacitor C S Capacitor C2 is charged through D4 and S, and N3 charges the output capacitor C through D7. o3 Charging, current i D4 and i D7It rises linearly, and the states of the remaining components are the same as in the previous mode.

[0043] 3) Mode III [t2~t3]: At t2, the switch S remains closed, and the equivalent circuit of the converter is as follows: Figure 6 At this time, diode D5 starts to conduct, and N2 starts to charge capacitor C3 through D5. The current i D5 As diode D5 conducts and shunts the current, the current i D4 It will drop briefly, and then continue to rise as the current in winding N2 increases. L2 The increase in the amplitude is getting larger and larger, and at this time it exceeds the current i of the inductor L1 L1 , capacitor C r The inductor L3 starts to be charged, and the current i flowing through the inductor L3 is L3 Start to reverse and continue to offset i L2 The ripple generated, the state of other components is the same as the previous mode.

[0044] 4) Mode IV [t3~t4]: At t3, the switch S starts to turn off, and the equivalent circuit of the converter is as follows: Figure 7 As shown. At this time, due to the existence of leakage inductance and MOSFET parasitic capacitance, the switch tube S cannot be turned off immediately. Excitation inductance L m The change from charging to discharging causes the voltage across windings N1, N2, and N3 to reverse, accelerating the leakage inductance L k2 , L k3 The discharge process, current i D1 、i D4 、i D5 、i D7 Rapidly decreases. Current i L2 、i Lm Starts to decrease, current i L3 The ripple cancellation makes the current i L1 When the switch tube S is completely turned off, the circuit enters the next mode, which is a short transition state.

[0045] 5) Mode V [t4~t5]: At t4, the switch S remains off, and the equivalent circuit of the converter is as follows: Figure 8 As shown. At this time, the switch tube S is completely turned off, the diodes D1, D4, D5, and D7 are turned off, and the diodes D2, D3, D6, and D8 are turned on. The voltage source V in Together with inductor L2, capacitor C1 is charged. S D3 is used to recover the leakage inductance energy and clamp the voltage of the switch tube S to avoid a spike voltage when the switch tube S is turned off. m The energy is transferred to the output capacitor C at the N3 end through the winding N1 end.o2 , and together with winding N2, capacitors C2 and C3, through D6 to the output capacitor C o1 Charging, inductors L1, L3 and capacitors C r The state of is the same as the previous mode. The expression of this mode is:

[0046] 6) Mode VI [t5-t6]: At t5, the switch S remains off, and the equivalent circuit of the converter is as follows: Fig. 9 As shown. At this time, the diode D3 is cut off and the capacitor C S No more charging, the states of other devices remain unchanged. At t6, the switch tube S is closed, this mode ends, and the next switching cycle begins.

[0047] (III) Converter voltage gain analysis

[0048] In order to simplify the steady-state analysis process of the converter, the influence of leakage inductance is ignored in the calculation, that is, the coupling coefficient k = 1. Mode I and IV are short-term transition modes, and here we mainly analyze Mode II, III, V, and VI. According to Kirchhoff's voltage law, the working state of the converter in Mode II and III can be analyzed, and the voltage relationship of the converter when the switch tube S is turned on can be obtained:

[0049] v L1 (t)+v L2 (t) = V in (2)

[0050] v L1 (t)+v L3 (t) = V in -V Cr (3)

[0051] v Lm (t) = V C1 (4)

[0052] n2v Lm (t) = V C2 -V CS =V C3 (5)

[0053] n3v Lm (t) = V Co3 (6)

[0054] V Co1 +V Co2 +V Co3 =V o (7)

[0055] From the working modes V and VI, we can get the voltage relationship of the converter when the switch tube S is turned off:

[0056] v L1 (t)+v L2 (t) = V in -V C1 (8)

[0057] v L1 (t)+v L3 (t) = V in -V Cr (9)

[0058] v Lm (t) = V C1 -V CS (10)

[0059] n2v Lm (t) = V C2 +V C3 +V CS -V Co1 (11)

[0060] n3v Lm (t)=-V Co2 (12)

[0061] According to the volt-second balance principle, the inductances L1, L2, L3, L m The average voltage in a switching cycle is 0. Combining equations (2), (3), (4), (8), (9), and (10), we can obtain the following relationship:

[0062]

[0063] According to formula (13), we can get:

[0064]

[0065] V Cr =V in (15)

[0066]

[0067] Combining equations (4), (5), (6) and (10), (11), (12) we can obtain the following relationship:

[0068]

[0069] According to formula (17), we can get:

[0070]

[0071] V C3 =n2V C1 (19)

[0072]

[0073]

[0074] V Co3 =n3V C1 (twenty two)

[0075] Substituting equation (14) into equation (16) and equations (18) to (22) respectively, the voltage expressions of each capacitor can be obtained as follows:

[0076]

[0077] Substituting equations (26), (27), and (28) into equation (7), we can obtain the output voltage of the converter and then calculate the voltage gain G:

[0078]

[0079] According to formula (28), it can be seen that the voltage gain of the converter proposed in this paper is not only related to the switch duty cycle D, but also to the turns ratio of the three-winding coupled inductor. Different voltage gains can be obtained by reasonably designing the duty cycle D and the turns ratio of the coupled inductor. Fig.10 The relationship curve between the voltage gain G and the duty cycle D under different turns ratios is shown. It can be seen that a higher voltage gain can be achieved under a lower duty cycle D.

[0080] (IV) Converter voltage stress analysis

[0081] By analyzing and determining the maximum voltage that each device can withstand, the appropriate component specifications can be selected to avoid damage caused by exceeding the maximum withstand limit of the component. This is especially important for the switches, diodes, and capacitors used in the design, and the analysis of voltage stress helps to identify high-loss areas in the system. By optimizing these areas, energy losses can be reduced and the overall efficiency of the converter can be improved. Therefore, analyzing device voltage stress is crucial to ensuring the reliability and long-term stable operation of the converter.

[0082] The analysis of each mode of the converter shows that Figure 5 and 8 The corresponding Mode II and Mode V can obtain the maximum voltage that the switch tube S and diodes D1 to D8 can withstand, and the relationship expression is as follows:

[0083]

[0084] It can be seen from equations (31) to (36) that the voltage stress on the device is related to the duty cycle D and the turns ratio. By selecting appropriate duty cycle D and turns ratio, the voltage stress of the device can be effectively reduced. Fig.11 The relationship curve between the voltage stress of the switch tube S and the duty cycle D under different turns ratios is shown. It can be seen that the voltage stress of the switch tube S decreases with the increase of the turns ratio. When the turns ratio n2≥1 and n3≥1, no matter what the value of the duty cycle D is, the voltage stress of the switch tube S is not greater than 0.25 times the output voltage.

[0085] (V) Converter input current ripple and device current stress analysis

[0086] During the conduction phase of the switch tube S, the current ripples of the three inductors L1, L2, and L3 at the input end are according to Figure 2 The converter equivalent circuit shown in the figure can obtain the current i in (t) = i L1 (t), that is, the current ripple Δi in =Δi L1 .Depend on Figure 5 , 6 The relationship between the three current ripples can be obtained from Mode II and III as shown:

[0087]

[0088] Rewrite equations (2) and (3) into the following form:

[0089]

[0090] Combining equations (15) and (37) to (39), we can get the current ripple expressions of the three inductors:

[0091]

[0092]

[0093] It can be seen from equation (40) that the smaller the inductance L3, the smaller the input current ripple Δi of the converter. in The lower the inductance, when L3 = 0, zero input current ripple can be achieved in theory. From formula (1), we can get l3 = L r -M, so by reasonably setting the mutual inductance M and auxiliary inductance L of the tapped inductor r A value of 0 can achieve approximately zero ripple input current.

[0094] In the following current stress analysis, because the inductors L1, L2 and L m The inductance value is large, and its current ripple is small, so a small ripple approximation can be made, that is, the current flowing through the three inductors is a DC value during the entire switching cycle. According to Kirchhoff's current law and the principle of capacitor charge balance, it can be obtained that each device has a switching cycle TS The average current in is:

[0095] I D3 =I D4 =I D5 =I D6 =I D7 =I D8 =I N2 =I o (43)

[0096] I D2 =I Lk1 =I Lm +I N1 =I Lm +n2I N2 =I Lm +n2I o (44)

[0097] I in =I L1 =I L2 =I D1 +I D2 =I D3 +I DS (45)

[0098] I L3 =I N3 =0 (46)

[0099] Assume that the average current of diode D1 during the conduction period of switch S is I' D1 , assuming that the average currents of diodes D3, D6, and D8 during the cut-off period of switch tube S are I” D3 , I” D6 , I” D8 ,Depend on Figure 3 It can be seen from the waveform that diode D1 only has current when the switch tube S is turned on, and diodes D3, D6, and D8 only have current when the switch tube S is turned off. According to the principle of conservation of current and charge, we can get:

[0100]

[0101] By analyzing Mode II and III, we can get the current relationship of the converter during the conduction period of the switch tube S:

[0102] I L2 =I' D1 (51)

[0103] Combining equations (45), (47) and (51) we can get the current relationship between diodes D1 and D2:

[0104]

[0105] Assume that the average current of windings N1, N2, and N3 during the cut-off period of switch tube S is I” N1 , I” N2 , I” N3 By analyzing Mode V and Mode VI, we can get the current relationship of the converter during the cut-off period of the switch tube S:

[0106] I Lk1 =I Lm +I” N1 =I” D3 +I” D6 (53)

[0107] I” N2 =I” D6 (54)

[0108] I” N3 =I” D8 (55)

[0109] i N1 (t)+n2i N2 (t)+n3i N3 (t)=0 (56)

[0110] Combining equations (48) to (50) and (53) to (56) we can get the magnetizing inductance L m The average current is:

[0111]

[0112] From formula (48), we can get:

[0113]

[0114] According to equations (57) and (58), the excitation inductance of the converter operating in critical conduction mode (BCM) can be obtained as:

[0115]

[0116] Substituting formula (57) into formula (44), (45), and (52) yields:

[0117]

[0118] Fig.12The relationship curve between the current stress of the switch tube S and the duty cycle D under different turns ratios is shown. It can be seen that the current stress of the switch tube S increases with the increase of the turns ratio. When the duty cycle D<0.6, the current stress of the switch tube is low. Once the duty cycle D>0.6, the current stress of the switch tube increases significantly, so the converter is not suitable for working under the condition of high duty cycle D.

[0119] (VI) Hardware platform verification

[0120] In order to verify the steady-state performance of the converter designed in the present invention, an experimental prototype with a rated power of 160W was built. Fig.13 Its specific parameters are shown in Table 1.

[0121] Table 1 Experimental prototype parameters

[0122] parameter Numeric <![CDATA[Switching frequency f s > 50kHz Capacitor Cr 3.3μF <![CDATA[Capacitor C1]]> 470μF <![CDATA[Capacitors C2, C3, C S > 47μF <![CDATA[Capacitor C o1 , C o2 , C o3 > 220μF <![CDATA[Tapped inductor L A 、L B > 33μH Tap inductor mutual inductance M 22μH <![CDATA[Inductor L r > 22μH <![CDATA[Excitation inductance L m > 230μH <![CDATA[Leakage inductance L k1 , L k2 , L k3 > 3μH <![CDATA[Turn ratio N1:N2:N3]]> 1：1：1 Switching tube S IRFP250NPBF(200V / 30A) <![CDATA[Diodes D1, D2]]> SK86C(60V / 8A) <![CDATA[Diodes D3 to D8]]> SS520(200V / 5A) Load resistance R 1KΩ

[0123] When the input voltage is 23V, the output of the converter is as follows: Fig.14 As shown, it can be seen that the output voltage is stable at 400V. At the same time, the input current ripple is measured as Fig.15 As shown, it can be seen that the input current ripple rate Δi in / I in Only 0.25%, which fully meets the requirements of photovoltaic grid-connected system.

[0124] The specific implementation methods described above further illustrate the objectives, technical solutions and beneficial effects of the present invention in detail. It should be understood that the above description is only a specific implementation method of the present invention and is not intended to limit the scope of protection of the present invention. Any modifications, equivalent substitutions, improvements, etc. made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A low input current ripple high voltage gain converter based on cascaded Boost and three-winding coupled inductor, comprising: Including DC source V in , tap inductance L A , L B , three-winding coupled inductor M1, auxiliary inductor L r , capacitor C r , C1, C2, C3, C S , C o1 , C o2 , C o3 , diodes D1, D2, D3, D4, D5, D6, D7, D8, MOS tube S, load resistor R o . The L A , L B They are a pair of tapped inductors, their mutual inductance is M, the tapped part and the auxiliary inductor L r 、Auxiliary capacitor C r connected in series to form a ripple cancellation structure; the voltage source V in , tapped inductor, diodes D1, D2, capacitor C1 and switch tube S together form a primary boost circuit, voltage source V in The tapped inductor first charges capacitor C1, and then capacitor C1 serves as the input source of the secondary boost circuit to transfer energy to the output capacitor C o1 ;Diode D3 and capacitor C S The passive clamping structure can not only suppress the voltage spike when the switch tube S is turned off, reduce the voltage stress of the switch tube S, but also recover the energy of the leakage inductance and improve the efficiency of the converter; diodes D3, D4, D5, capacitor C S , C2, C3 and coupled inductor secondary winding N2 form voltage multiplier module 1; diodes D7, D8, output capacitor C o2 , C o3 The voltage multiplier module 2 is formed with the coupled inductor secondary winding N3; the capacitor C1 and the coupled inductor primary winding N1 are used as input terminals, and together with all subsequent devices, a secondary boost circuit is formed.

2. The low input current ripple and high voltage gain converter based on cascaded Boost and three-winding coupled inductor according to claim 1, characterized in that: Flowing through the auxiliary inductor L r and auxiliary capacitor C r The average current is 0, and the peak value of the auxiliary current is small, which is mainly related to the tap inductance L B The ripples generated by the charging and discharging of the terminals are offset, making the tapped inductor L A The current ripple at the end is very small.

3. The low input current ripple and high voltage gain converter based on cascaded Boost and three-winding coupled inductor according to claim 1, characterized in that: The converter has two steps of boosting. The first step is to boost the voltage with V in For input V C1 The second time is V C1 For input V o For the output, the double boost doubles the converter voltage gain.

4. The low input current ripple and high voltage gain converter based on cascaded Boost and three-winding coupled inductor according to claim 1, characterized in that: There are two voltage multiplier modules. The first one is located behind the switch tube S and integrated with the N2 end of the three-winding coupled inductor. It can not only greatly improve the voltage gain of the converter, but also clamp the voltage of the switch tube S to avoid voltage spikes, greatly reduce the voltage stress of the switch tube S, recover leakage inductance energy, and improve converter efficiency. The second one is located at the output capacitor C o1 On the upper side, with the output capacitor C o2 , C o3 Combined with the N3 terminal of the three-winding coupled inductor, the voltage gain of the converter is further improved.

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