Resonant LLC switching power supply converter and control method thereof

Abstract

The invention discloses a resonant LLC switching power supply converter and a control method thereof, and the resonant LLC switching power supply converter comprises a first switch module, a second switch module, a third switch module, a fourth switch module, a first capacitor module, a second capacitor module, a third capacitor module, an energy storage and isolation module, and a control module. The control module respectively controls the first switch module and the second switch module through the first control signal and the second control signal to obtain an output current, compares the output current with a preset current, and controls the on-off state of the third switch module according to a comparison result and the state of the first control signal. And the on-off state of the fourth switch module is controlled according to the comparison result and the state of the second control signal. By monitoring the output current in real time, the dynamic regulation of the resonant LLC switching power supply converter is realized, the zero-voltage turn-off or turn-on is ensured to be realized in a large-current load, the capacitor module is disconnected in a small-current load, and the power loss is reduced.

Description

Technical Field

[0001] The present invention relates to the technical field of switching power supplies, and in particular to a resonant LLC switching power converter and a control method thereof. Background Art

[0002] In the prior art, PC power supplies, server power supplies, LED power supplies, adapters, rail power supplies, high-end industrial power supplies, etc. are all high-frequency switching power supplies with high conversion efficiency requirements. The main power topology circuit of the DC / DC isolation converter in these industry power supplies widely uses high-efficiency, low-cost resonant half-bridge LLC switching power converter. It is mainly composed of input filter capacitors, high-voltage side switching transistors on the switching bridge arm, low-voltage side switching transistors on the switching bridge arm, resonant inductors, resonant capacitors, isolation transformers, secondary output rectifier diodes (or synchronous rectifier MOS tubes), output filter capacitors, control circuits, etc.

[0003] like Figure 1 As shown, in the prior art resonant half-bridge LLC switching power converter, the first switch module and the second switch module generally select N-channel field effect transistors, and in the process of realizing complementary opening or closing of the first switch module and the second switch module, the high current hard-off working state of one transistor is a necessary condition for realizing zero voltage opening of the other transistor. Therefore, this serious problem is an inherent defect of the prior art resonant half-bridge LLC switching power converter topology technology, and cannot be solved by debugging the parameters of components.

[0004] In order to solve the problem of high current hard turn-off loss of the first switch module and the second switch module of the bridge arm of the resonant LLC switching power converter, the following method is generally adopted:

[0005] 1. Adjust the frequency of the converter when it is working at full load to near the second resonant frequency of the resonant cavity to minimize the current value when the first switch module or the second switch module is turned off, and minimize its hard turn-off loss.

[0006] The above technical solution has the following technical defects: Due to the rapid development of the industry economy, the demand for instantaneous output peak power of the switching power supply in the personal computer, automation industry, robotic arm and other industries has increased, and the peak output power of the power supply is generally required to reach about 2-3 times the average output power. Under this requirement, as mentioned above, in order to ensure that the LLC switching power converter can work normally under any load, it is necessary to design its operating frequency f at peak load. w is reliably higher than the first resonant frequency f1 of its resonant cavity, resulting in its operating frequency f at average load wIt will also be much higher than the second resonant frequency f2 of its resonant cavity. At this time, the first switch module or the second switch module is hard-turned off at a larger current, causing the switching power supply to have serious problems such as increased average turn-off loss, enhanced electromagnetic interference EMI radiation, and reduced efficiency.

[0007] 2. Connecting a capacitor in parallel between the drain D and the source S of the first switch module and the second switch module of the resonant half-bridge LLC switching power converter can reduce the loss of the switch tube when it is hard-turned off at a high current.

[0008] The above technical solution has the following technical defects: a capacitor is directly connected in parallel between the drain D and the source S of the first switch module and the second switch module. When the driving pulse is a discontinuous pulse group output, the first switch module and the second switch module are in a hard-on state at the first pulse after the driving pulse is interrupted. At this time, the auxiliary capacitor added by the first switch module and the second switch module will increase the switching loss. Because under normal circumstances, when the resonant LLC switching power converter is in no-load standby and extremely small load, the control module adjusts the operating frequency to increase to the set maximum safe frequency value in order to stabilize the output voltage, but it still cannot achieve the voltage stabilization requirement. At this time, in order to stabilize the output voltage, the control module will enter a state of continuously interrupted driving pulse group output. Therefore, the above-mentioned prior art method will increase the loss of the LLC switching power converter in standby and small load, and reduce the efficiency. Summary of the invention

[0009] The embodiment of the present invention provides a resonant LLC switching power converter and a control method thereof to solve the above technical problems.

[0010] A first aspect of an embodiment of the present invention provides a resonant LLC switching power converter, comprising: the resonant LLC switching power converter comprises a first switch module, a second switch module, a third switch module, a fourth switch module, a first capacitor module, a second capacitor module, a third capacitor module, an energy storage and isolation module, and a control module;

[0011] The first end of the first switch module is connected to the first end of the second capacitor module, the second end of the first switch module is respectively connected to the first end of the second switch module, the second end of the third switch module, the first end of the third capacitor module and the first end of the energy storage and isolation module, the second end of the second capacitor module is connected to the first end of the third switch module, the second end of the second switch module is respectively connected to the second end of the fourth switch module and the first end of the first capacitor module, the second end of the first capacitor module is connected to the second end of the energy storage and isolation module, the second end of the third capacitor module is connected to the first end of the fourth switch module, and the control module is respectively connected to the control end and the second end of the first switch module, the control end and the second end of the second switch module, the control end and the second end of the third switch module and the control end and the second end of the fourth switch module;

[0012] The control module is used to output a first control signal to the first switch module to control the on / off state of the first switch module, and output a second control signal to the second switch module to control the on / off state of the second switch module;

[0013] The control module is also used to obtain the output current of the resonant LLC switching power converter, compare the output current with a preset current, and control the on-off state of the third switch module according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module according to the comparison result and the state of the second control signal.

[0014] Optionally, the control module includes:

[0015] a first control module, connected to a control end of the first switch module and a control end of the second switch module, respectively, and configured to output a first control signal to the first switch module to control the on / off state of the first switch module, and output a second control signal to the second switch module to control the on / off state of the second switch module;

[0016] a second control module, which is respectively connected to the control end of the first switch module, the control end of the second switch module, the control end of the third switch module and the control end of the fourth switch module, and is used to obtain the output current of the resonant LLC switching power converter, compare the output current with a preset current, and control the on-off state of the third switch module according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module according to the comparison result and the state of the second control signal.

[0017] Optionally, the second control module includes:

[0018] A reference voltage module, used for outputting a reference voltage;

[0019] a current detection module, wherein a first input end thereof is connected to the second end of the first capacitor module, and a second input end thereof is connected to the output end of the reference voltage module, and is used to collect the output current of the resonant LLC switching power converter, convert the output current into a collection voltage, and compare the collection voltage with the reference voltage to output a first comparison signal;

[0020] a first control submodule, wherein a first input end thereof receives the first control signal, a second input end thereof is connected to an output end of the reference voltage module, a third input end thereof is connected to an output end of the current detection module, an output end thereof is connected to a control end of the third switch module, and a first on-off control signal is output to the third switch module according to the first control signal, the reference voltage and the first comparison signal;

[0021] A second control submodule, whose first input end receives the second control signal, whose second input end is connected to the output end of the reference voltage module, whose third input end is connected to the output end of the current detection module, and whose output end is connected to the control end of the fourth switch module, outputs a second on-off control signal to the fourth switch module according to the second control signal, the reference voltage and the first comparison signal.

[0022] Optionally, the first control submodule includes:

[0023] a first delay trigger module, whose first input terminal receives the first control signal, whose second input terminal receives the reference voltage, and whose third input terminal receives the first comparison signal, and is used to output a first weak current on-off control signal according to the first control signal, the reference voltage and the first comparison signal;

[0024] The first driving module has an input end connected to the output end of the first delay trigger module and is used to amplify and isolate the first weak current on-off control signal and then output a first on-off control signal.

[0025] Optionally, the first delay trigger module includes:

[0026] A first comparison module, whose first input terminal receives the first control signal and whose second input terminal receives the reference voltage, is used to delay the conversion of the first control signal into a first acquisition voltage, and output a first trigger signal when the first acquisition voltage is greater than the reference voltage;

[0027] A first trigger module, whose first input end receives the first comparison signal and whose second input end is connected to the output end of the first comparison module, is used to output the first weak current on-off control signal according to the first trigger signal.

[0028] Optionally, the second control submodule includes:

[0029] a second delay trigger module, whose first input terminal receives the second control signal, whose second input terminal receives the reference voltage, and whose third input terminal receives the first comparison signal, and is used to output a second weak current on-off control signal according to the second control signal, the reference voltage and the first comparison signal;

[0030] The second driving module has an input end connected to the output end of the second delay trigger module, and is used to amplify and isolate the output second weak current on-off control signal and then output a second on-off control signal.

[0031] Optionally, the second delay trigger module includes:

[0032] A second comparison module, whose first input terminal receives the second control signal and whose second input terminal receives the reference voltage, is used to delay the conversion of the second control signal into a second acquisition voltage, and output a second trigger signal when the second acquisition voltage is greater than the reference voltage;

[0033] The second trigger module has a first input end receiving the first comparison signal and a second input end connected to the output end of the second comparison module, and is used for outputting the second weak current on-off control signal according to the second trigger signal.

[0034] Optionally, the first switch module is a first MOS tube, the second switch module is a second MOS tube, the first control signal includes a first voltage control signal and a second voltage control signal, and the second control signal includes a third voltage control signal and a fourth voltage control signal;

[0035] The resonant LLC switching power converter further comprises:

[0036] a first voltage dividing unit, wherein a first end of the first voltage dividing unit receives the first voltage control signal, a second end of the first voltage dividing unit is connected to the gate of the first MOS tube, a third end of the first voltage dividing unit is connected to the source of the first MOS tube and then receives the second voltage control signal, and controls the on and off of the first MOS tube according to the first voltage control signal and the second voltage control signal;

[0037] a second voltage dividing unit, wherein a first end of the second voltage dividing unit receives the third voltage control signal, a second end of the second voltage dividing unit is connected to the gate of the second MOS tube, a third end of the second voltage dividing unit is connected to the source of the second MOS tube and then receives the fourth voltage control signal, and controls the on and off of the second MOS tube according to the third voltage control signal and the fourth voltage control signal;

[0038] The second control module is used to control the third switch module to start conducting when it is detected that the output current is greater than the preset current and a high level signal is formed between the first voltage control signal and the second voltage control signal or a high level signal is formed between the second voltage control signal and the ground terminal of the second control module;

[0039] The second control module is further configured to control the fourth switch module to start conducting when the output current is greater than a preset current and a high level signal is formed between the third voltage control signal and the fourth voltage control signal.

[0040] A second aspect of an embodiment of the present invention provides a control method for the resonant LLC switching power converter according to claim 1, wherein the control method is applied to the control module and comprises:

[0041] Outputting a first control signal to the first switch module to control the on / off state of the first switch module, and outputting a second control signal to the second switch module to control the on / off state of the second switch module;

[0042] The output current of the resonant LLC switching power converter is obtained, the output current is compared with a preset current, and the on-off state of the third switch module is controlled according to the comparison result and the state of the first control signal, and the on-off state of the fourth switch module is controlled according to the comparison result and the state of the second control signal.

[0043] Optionally, obtaining the output current of the resonant LLC switching power converter includes:

[0044] The output current of the resonant LLC switching power converter is obtained by detecting the effective value of the AC component of the voltage across the second capacitor module.

[0045] Optionally, controlling the on / off state of the third switch module according to the current comparison result and the state of the first control signal includes:

[0046] When the output current is greater than the preset current and the first control signal is switched from the first level signal to the second level signal, the third switch module is controlled to switch from the first switch state to the second switch state with a delay, and the second switch state is maintained;

[0047] When the output current is not greater than the preset current and the first control signal is switched from the first level signal to the second level signal, the third switch module is controlled to switch from the second switch state to the first switch state with delay, and the first switch state is maintained.

[0048] Optionally, controlling the on / off state of the fourth switch module according to the current comparison result and the state of the second control signal includes:

[0049] When the output current is greater than the preset current and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is controlled to switch from the first switch state to the second switch state with a delay, and the second switch state is maintained;

[0050] When the output current is not greater than the preset current and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is delayed to control the switch from the second switch state to the first switch state and maintain the first switch state.

[0051] The technical effect of the embodiment of the present invention is: by real-time monitoring of the output current and comparing it with the preset current, dynamic adjustment of the resonant LLC switching power converter is achieved. When the load state changes, the control module can intelligently adjust the on-off state of the third switch module and the fourth switch module to ensure zero voltage shutdown or conduction under high current load, and disconnect the capacitor module under low current load to reduce power loss. This design effectively avoids the problem of efficiency reduction caused by the increase of hard-off loss of the first switch module and the second switch module under high load current in the prior art, and the problem that the drive waveform output by the control circuit increases the hard-on loss in standby and low load conditions, which is an intermittent pulse group and a skip cycle state. The circuit of the present invention greatly reduces the hard-on and hard-off losses of the first switch module and the second switch module, can maintain low standby power consumption and high-efficiency operation, and significantly improves the overall efficiency and adaptability of the power converter. BRIEF DESCRIPTION OF THE DRAWINGS

[0052] In order to more clearly illustrate the technical solutions of the embodiments of the present invention, the accompanying drawings required for use in the description of the embodiments of the present invention will be briefly introduced below. Obviously, the accompanying drawings in the following description are only some embodiments of the present invention. For ordinary technicians in this field, other accompanying drawings can be obtained based on these accompanying drawings without paying creative labor.

[0053] Figure 1 It is a structural schematic diagram of a resonant LLC switching power converter provided by the prior art;

[0054] Figure 2 This is a first structural schematic diagram of a resonant LLC switching power converter provided by Embodiment 1 of the present invention;

[0055] Figure 3 This is a second structural schematic diagram of a resonant LLC switching power converter provided by the first embodiment of the present invention;

[0056] Figure 4 This is a first structural schematic diagram of a second control module in a resonant LLC switching power converter provided in Embodiment 1 of the present invention;

[0057] Figure 5 It is a structural schematic diagram of a first control submodule in a second control module in a resonant LLC switching power converter provided in Embodiment 1 of the present invention;

[0058] Figure 6 It is a structural schematic diagram of a first delay trigger module in a first control submodule in a second control module in a resonant LLC switching power converter provided in Embodiment 1 of the present invention;

[0059] Figure 7 It is a structural schematic diagram of a second control submodule in a second control module in a resonant LLC switching power converter provided in Embodiment 1 of the present invention;

[0060] Figure 8 It is a structural schematic diagram of a second delay trigger module in a second control submodule in a second control module in a resonant LLC switching power converter provided in Embodiment 1 of the present invention;

[0061] Fig. 9 is a circuit diagram of a resonant LLC switching power converter provided by Embodiment 1 of the present invention;

[0062] Fig.10 This is a schematic diagram of the timing sequence of transistor drive voltage waveforms output by a first control module and a second control module of a resonant LLC switching power converter provided in Embodiment 1 of the present invention;

[0063] Fig.11 This is a second structural schematic diagram of a second control module in a resonant LLC switching power converter provided in the first embodiment of the present invention;

[0064] Fig.12 is a circuit diagram of a second control module in a resonant LLC switching power converter provided by Embodiment 1 of the present invention;

[0065] Fig.13 It is a comparison diagram of the driving voltage waveform of the main switch transistors Q1 and Q2 and the current waveform of the main components on the same time axis in a resonant LLC switching power converter provided by the first embodiment of the present invention;

[0066] Fig.14 It is a comparison diagram of the driving voltage waveform of the main switch transistors Q1 and Q2 and the voltage waveform of the main components on the same time axis in a resonant LLC switching power converter provided by the first embodiment of the present invention;

[0067] Fig.15 yes Fig.13 , Fig.14 Detailed expansion diagram of the voltage and current waveforms of the four working modes identified in the same time axis comparison diagram;

[0068] Fig.16 is a flow chart of a control method of a resonant LLC switching power converter provided by Embodiment 2 of the present invention;

[0069] Fig.17 is a specific flow chart of step S20 in a control method of a resonant LLC switching power converter provided in Embodiment 2 of the present invention;

[0070] Fig.18 is another specific flow chart of step S20 in a control method for a resonant LLC switching power converter provided in the second embodiment of the present invention;

[0071] In the figure: 101, the first switch module; 102, the second switch module; 103, the third switch module; 104, the fourth switch module; 105, the first capacitor module; 106, the second capacitor module; 107, the third capacitor module; 108, the energy storage and isolation module; 110, the control module; 111, the first control module; 112, the second control module; 201, the current detection module; 202, the reference voltage module; 203, the first delay trigger module; 204, the first drive module; 205, the second delay trigger module; 206, the second drive module; 211, the first control submodule; 212, the second control submodule; 241, the first comparison module; 242, the first trigger module; 243, the second comparison module; 244, the second trigger module. DETAILED DESCRIPTION

[0072] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0073] It should be understood that the present invention can be implemented in different forms and should not be construed as being limited to the embodiments set forth herein. On the contrary, providing these embodiments will make the disclosure thorough and complete and fully convey the scope of the present invention to those skilled in the art. In the accompanying drawings, the sizes and relative sizes of layers and regions may be exaggerated for clarity. The same reference numerals throughout represent the same elements.

[0074] It should be understood that when an element or layer is referred to as being "on, adjacent to, connected to or coupled to other elements or layers, it may be directly on, adjacent to, connected to or coupled to other elements or layers, or there may be intervening elements or layers. In contrast, when an element is referred to as being "directly on, directly adjacent to, directly connected to or directly coupled to other elements or layers, there may be no intervening elements or layers. It should be understood that, although the terms first, second, third, etc. may be used to describe various elements, components, regions, layers and / or parts, these elements, components, regions, layers and / or parts should not be limited by these terms. These terms are only used to distinguish one element, component, region, layer or part from another element, component, region, layer or part. Therefore, without departing from the teachings of the present invention, the first element, component, region, layer or part discussed below may be represented as a second element, component, region, layer or part.

[0075] In order to fully understand the present invention, detailed structures and steps will be presented in the following description to illustrate the technical solutions proposed by the present invention. The preferred embodiments of the present invention are described in detail below, but in addition to these detailed descriptions, the present invention may also have other implementations.

[0076] Embodiment 1

[0077] This embodiment 1 provides a resonant LLC switching power converter, such as Figure 2 As shown, including:

[0078] The resonant LLC switching power converter includes a first switch module 101, a second switch module 102, a third switch module 103, a fourth switch module 104, a first capacitor module 105, a second capacitor module 106, a third capacitor module 107, an energy storage and isolation module 108 and a control module 110;

[0079] The first end of the first switch module 101 is connected to the first end of the second capacitor module 106, the second end of the first switch module 101 is respectively connected to the first end of the second switch module 102, the second end of the third switch module 103, the first end of the third capacitor module 107 and the first end of the energy storage and isolation module 108, the second end of the second capacitor module 106 is connected to the first end of the third switch module 103, the second end of the second switch module 102 is respectively connected to the second end of the fourth switch module 104 and the first end of the first capacitor module 105, the second end of the first capacitor module 105 is connected to the second end of the energy storage and isolation module 108, the second end of the third capacitor module 107 is connected to the first end of the fourth switch module 104, and the control module 110 is respectively connected to the control end and the second end of the first switch module 101, the control end and the second end of the second switch module 102, the control end and the second end of the third switch module 103 and the control end and the second end of the fourth switch module 104;

[0080] The control module 110 is used to output a first control signal to the first switch module 101 to control the on / off state of the first switch module 101, and output a second control signal to the second switch module 102 to control the on / off state of the second switch module 102;

[0081] The control module 110 is also used to obtain the output current of the resonant LLC switching power converter, compare the output current with a preset current, and control the on-off state of the third switch module 103 according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module 104 according to the comparison result and the state of the second control signal.

[0082] Among them, the first switch module 101 is controlled by the first control signal output by the control module 110. When receiving the start control signal, the first switch module 101 is turned on to make the current flow to the energy storage and isolation module 108. When receiving the shutdown control signal, the first switch module 101 is disconnected and the current path is interrupted. The second switch module 102 works in conjunction with the first switch module 101 to provide a switch signal of another phase to form a complete bridge topology. The second switch module 102 is controlled by the second control signal output by the control module 110. When the first switch module 101 is turned off, the second switch module 102 is turned on to ensure the continuity of the current. The third switch module 103 and the fourth switch module 104 are dynamically adjusted according to the output current state to achieve zero voltage soft opening and reduce loss and interference. The on and off of the third switch module 103 and the fourth switch module 104 are strictly controlled by the load current and the main switch state to ensure no interference to the main circuit and maintain high-efficiency operation. The first capacitor module 105 forms an oscillation circuit with the energy storage and isolation module 108 when the circuit is turned on. The second capacitor module 106, as part of the resonant circuit, provides the resonant capacitance required by the circuit at the transition moment of the switch module switching, and forms a resonant circuit with the energy storage and isolation module 108 and the switch module to generate a resonant signal, store energy and release energy during the resonance process, and reduce the turn-off loss of the first switch module 101. The third capacitor module 107, as part of the resonant circuit, provides the resonant capacitance required by the circuit at the transition moment of the first switch module 101 switching, and forms a resonant circuit with the energy storage and isolation module 108 and the first switch module 101 to generate a resonant signal, store energy and release energy during the resonance process, and reduce the turn-off loss of the second switch module 102. The energy storage and isolation module 108 includes a transformer or an inductor element for energy storage, energy transfer and input-output isolation. When the switch module is turned on, the energy storage module stores energy, and the electrical isolation of the input and output is realized through the transformer. The energy storage module releases the stored energy to the output at the resonant frequency. The control module 110 is used to generate a switch control signal and adjust the switch state according to the feedback signal of the output current. The control module 110 obtains the output current of the resonant LLC power converter, compares the output current with the preset current, and adjusts the state of the third switch module 103 and the fourth switch module 104 according to the result. A high-frequency control signal is output to the first switch module 101 and the second switch module 102 to ensure that the entire circuit operates within a reliable frequency range. The control module 110 monitors the output current of the resonant LLC switching power converter in real time. The output current reflects the load state and working condition of the power supply and is a key parameter for adjusting the power conversion efficiency and control strategy. The control module 110 compares the real-time collected output current with the preset reference current. The preset current is an ideal current value set according to design requirements or load characteristics, which is used to determine whether the system is in a high current load state or a low current load state.According to the comparison result between the output current and the preset current, the control module 110 determines whether the state of the third switch module 103 needs to be adjusted. If the output current exceeds the preset value and is determined to be in a high current load state, the control module 110 will adjust the on-off of the third switch module 103 based on the state of the first control signal (such as a level change) to ensure that the third switch module 103 is turned off or turned on with zero voltage. If the output current does not exceed the preset value and is determined to be in a low current load state, the control module 110 will adjust the on-off of the third switch module 103 based on the state of the first control signal (such as a level change) to disconnect the second capacitor module 106 from the circuit. Similarly, the control module 110 will also control the on-off state of the fourth switch module 104 based on the same comparison result between the output current and the preset current, combined with the state of the second control signal. If the output current exceeds the preset value and is determined to be in a high current load state, the control module 110 will adjust the on-off of the fourth switch module 104 based on the state of the second control signal (such as a level change) to ensure that the fourth switch module 104 is turned off or turned on with zero voltage. If the output current does not exceed the preset value and is determined to be in a low current load state, the control module 110 will adjust the on / off of the fourth switch module 104 based on the state (such as level change) of the second control signal to disconnect the third capacitor module 107 from the circuit.

[0083] The technical effect of the technical solution provided in the first embodiment is that the dynamic adjustment of the resonant LLC switching power converter is realized by real-time monitoring of the output current and comparing it with the preset current. When the load state changes, the control module can intelligently adjust the on-off state of the third switch module and the fourth switch module to ensure zero voltage shutdown or conduction under high current load, and disconnect unnecessary capacitor modules under low current load to reduce power loss. This design effectively avoids the problem of efficiency reduction caused by the increase of hard-off loss of the first switch module and the second switch module under high load current in the prior art, and the problem that the drive waveform output by the control circuit increases the hard-on loss in standby and small load conditions, which is an intermittent pulse group and a skip cycle state. The circuit of the present invention greatly reduces the hard-on and hard-off losses of the first switch module and the second switch module, can maintain low standby power consumption and high-efficiency operation, and significantly improves the overall efficiency and adaptability of the power converter.

[0084] As an implementation method, Figure 3 As shown, the control module 110 includes:

[0085] A first control module 111, which is connected to a control end of the first switch module 101 and a control end of the second switch module 102, and is used to output a first control signal to the first switch module 101 to control the on / off state of the first switch module 101, and output a second control signal to the second switch module 102 to control the on / off state of the second switch module 102;

[0086] The second control module 112 is respectively connected to the control end of the first switch module 101, the control end of the second switch module 102, the control end of the third switch module 103 and the control end of the fourth switch module 104, and is used to obtain the output current of the resonant LLC switching power converter, compare the output current with the preset current, and control the on-off state of the third switch module 103 according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module 104 according to the comparison result and the state of the second control signal.

[0087] Among them, the first control module 111 controls the on-off state of the main switch module (the first switch module 101 and the second switch module 102). Generate a first control signal and a second control signal to drive the first switch module 101 and the second switch module 102 to work alternately. The first control module 111 generates the first control signal and the second control signal according to the system operation state, outputs the first control signal to the first switch module 101 to control its on-off state, and outputs the second control signal to the second switch module 102 to control its on-off state. The first control signal and the second control signal are alternately turned on to ensure that the first switch module 101 and the second switch module 102 work alternately. The second control module 112 realizes dynamic control of the auxiliary switch module (the third switch module 103 and the fourth switch module 104). Based on the output current signal collected in real time, combined with the state of the first control signal and the second control signal, the on-off state of the third switch module 103 and the fourth switch module 104 is adjusted to achieve zero voltage conduction of the first switch module 101 and the second switch module 102.

[0088] The second control module 112 acquires the output current of the resonant LLC converter in real time, and compares it with the preset current to generate a comparison result. According to the comparison result between the output current and the preset current, combined with the state of the first control signal, the on-off state of the third switch module 103 is controlled, and according to the comparison result between the output current and the preset current, combined with the state of the second control signal, the on-off state of the fourth switch module 104 is controlled. Specifically: when the output current is greater than the preset current, and the first control signal switches from the first level signal to the second level signal, the third switch module 103 is delayed to switch from the first switch state to the second switch state, and the second switch state is maintained; when the output current is greater than the preset current, and the second control signal switches from the first level signal to the second level signal, the fourth switch module 104 is delayed to switch from the first switch state to the second switch state, and the second switch state is maintained. The second switch state is the on state, which realizes zero voltage conduction or shutdown of the first switch module 101 and the second switch module 102 under large current conditions. When the output current is not greater than the preset current, and the first control signal is switched from the first level signal to the second level signal, the third switch module 103 is delayed to switch from the second switch state to the first switch state, and the first switch state is maintained. When the output current is not greater than the preset current, and the second control signal is switched from the first level signal to the second level signal, the fourth switch module 104 is delayed to switch from the second switch state to the first switch state, and the first switch state is maintained. The first switch state is the off state, which realizes that the third switch module 103 and the fourth switch module 104 are disconnected under low current conditions, thereby making the second capacitor module 106 and the third capacitor module 107 in an open circuit state, avoiding the operating frequency f w The problem of increasing the hard-turn-on power loss of the LLC switching power converter during the intermittent pulse group and cycle skipping states.

[0089] The technical effect of this embodiment is that through the coordinated control of the first control module and the second control module, the resonant LLC switching power converter can achieve zero voltage turn-on and turn-off under large current load and efficient operation under small current load. The first control module ensures the alternating conduction of the main switch module to achieve stable resonant operation; the second control module dynamically adjusts the state of the auxiliary switch module through real-time current feedback, optimizes energy conversion at high current, reduces switching losses, and avoids additional hard-on power losses caused by frequency discontinuity at low current. The overall design effectively improves the efficiency, stability and adaptability of the converter.

[0090] As an implementation method, Figure 4 As shown, the second control module 112 includes:

[0091] A reference voltage module 202, used for outputting a reference voltage;

[0092] A current detection module 201, whose first input end is connected to the second end of the first capacitor module 105, and whose second input end is connected to the output end of the reference voltage module 202, is used to collect the output current of the resonant LLC switching power converter, convert the output current into a collection voltage, and compare the collection voltage with the reference voltage to output a first comparison signal;

[0093] A first control submodule 211, whose first input terminal receives a first control signal, whose second input terminal is connected to the output terminal of the reference voltage module 202, whose third input terminal is connected to the output terminal of the current detection module 201, whose output terminal is connected to the control terminal of the third switch module 103, outputs a first on-off control signal to the third switch module 103 according to the first control signal, the reference voltage and the first comparison signal;

[0094] The second control submodule 212 has a first input terminal receiving a second control signal, a second input terminal connected to the output terminal of the reference voltage module 202, a third input terminal connected to the output terminal of the current detection module 201, and an output terminal connected to the control terminal of the fourth switch module 104, and outputs a second on-off control signal to the fourth switch module 104 according to the second control signal, the reference voltage and the first comparison signal.

[0095] Among them, the reference voltage module 202 outputs a stable reference voltage as a reference value for the current detection module 201, the first control submodule 211, and the second control submodule 212, and is used to determine the judgment criteria for current control and comparison in the resonant LLC switching power converter. The reference voltage module 202 generates and outputs a fixed reference voltage, which is provided to the current detection module 201 for comparison with the acquisition voltage, and is also provided to the first control submodule 211 and the second control submodule 212 to participate in generating the on-off control signal. The current detection module 201 collects the output current signal of the resonant LLC switching power converter, converts the collected output current into a corresponding acquisition voltage, compares the acquisition voltage with the reference voltage, and generates a first comparison signal as a basis for subsequent control. The current detection module 201 receives the output current signal from the second end of the first capacitor module 105, converts the output current signal into an acquisition voltage through an internal conversion circuit, and compares the magnitude relationship between the acquisition voltage and the reference voltage in real time. When the acquisition voltage exceeds the reference voltage, a first comparison signal is output to indicate whether the current current state meets the trigger condition. The first control submodule 211 generates a first on-off control signal according to the first control signal, the reference voltage and the first comparison signal, controls the on-off operation of the third switch module 103, and participates in the dynamic adjustment of the resonant circuit at the switching transition moment of the first switch module 101. The first control submodule 211 receives the first control signal (used to indicate the starting point of the on-off logic), the reference voltage (as a reference value) and the first comparison signal (indicating the current current state), determines whether the first control signal reaches the trigger condition, and performs logic processing on the reference voltage and the first comparison signal at the same time, and generates the first on-off control signal according to the comprehensive judgment. The signal is output to the control end of the third switch module 103, triggering the on-off action of the third switch module 103, so that the switching loss of the main switch module 101 in different working states is greatly reduced. The second control submodule 212 generates a second on-off control signal according to the second control signal, the reference voltage and the first comparison signal. Control the on-off operation of the fourth switch module 104, and participate in the dynamic adjustment of the resonant circuit at the switching transition moment of the second switch module 102. The second control submodule 212 receives the second control signal, the reference voltage and the first comparison signal, determines whether the second control signal reaches the trigger condition, and performs logic processing on the reference voltage and the first comparison signal. According to the comprehensive judgment, a second on-off control signal is generated, and the signal is output to the control end of the fourth switch module 104 to trigger the on-off action of the fourth switch module 104, so that the switching loss of the main switch module 101 in different working states is greatly reduced.

[0096] The technical effect of this embodiment is that through the coordinated work of the reference voltage module, the current detection module, the first control submodule and the second control submodule, the switching loss of the main switch module of the resonant LLC switching power converter is dynamically controlled and greatly reduced. The reference voltage module provides a stable reference voltage, and the current detection module collects the current in real time and generates a comparison signal to provide a basis for subsequent control. The first control submodule and the second control submodule generate accurate on-off control signals according to the input control signal, the reference voltage and the current comparison signal, and respectively drive the actions of the third switch module and the fourth switch module, thereby dynamically adjusting and reducing the switching loss of the main switch module. This design effectively ensures that the system operates efficiently under different load conditions, while effectively reducing circuit interference.

[0097] As an implementation method, Figure 5 As shown, the first control submodule 211 includes:

[0098] A first delay trigger module 203, whose first input terminal receives a first control signal, whose second input terminal receives a reference voltage, and whose third input terminal receives a first comparison signal, is used to output a first weak current on-off control signal according to the first control signal, the reference voltage and the first comparison signal;

[0099] The first driving module 204 has an input end connected to the output end of the first delay trigger module 203 and is used to amplify and isolate the first weak current on-off control signal and then output the first on-off control signal.

[0100] Among them, the first delay trigger module 203 performs delay processing on the first control signal, combines the reference voltage and the first comparison signal, and outputs the first weak current on-off control signal, which is used to realize the precise on-off control of the output weak current. The first delay trigger module 203 receives three input signals, the first control signal is used as the core control basis to indicate the starting point of the trigger logic, and the reference voltage is used to compare with the internally generated signal to determine the trigger condition. The first comparison signal is used to assist in determining whether the trigger condition is met, delay processing is performed on the first control signal, and the first acquisition voltage is generated, and the magnitude relationship between the first acquisition voltage and the reference voltage is monitored in real time. When the first acquisition voltage exceeds the reference voltage and the first comparison signal meets the trigger condition, the first weak current on-off control signal is output to control the operation of the subsequent circuit module. If the condition is not met, no signal is output to ensure the safe operation of the circuit. The first driving module 204 receives the first weak current on-off control signal output by the first delay trigger module 203, amplifies and electrically isolates the first weak current on-off control signal, and outputs the first on-off control signal to drive the subsequent circuit. The control signal is electrically isolated from the power circuit through the isolation circuit to avoid interference of high voltage or large current on the control circuit, and the first on-off control signal is output to directly drive the on-off operation of the power device to achieve the control target.

[0101] The technical effect of this embodiment is that through the coordinated work of the first delay trigger module and the first drive module, accurate on-off control and safe driving of the third switch module are achieved. The first delay trigger module is based on delay processing and multi-signal judgment mechanism, and only outputs reliable weak current on-off control signals when the trigger conditions are met, ensuring the accuracy and safety of control. The first drive module amplifies and electrically isolates the weak current signal, converting it into a high current or high voltage signal that can drive the power device, while avoiding interference of high voltage or large current on the control circuit. This design effectively improves the driving ability, anti-interference and stability of the control signal, and realizes efficient and safe circuit operation.

[0102] As an implementation method, Figure 6 As shown, the first delay trigger module 203 includes:

[0103] A first comparison module 241, whose first input terminal receives a first control signal and whose second input terminal receives a reference voltage, is used to delay the first control signal and convert it into a first acquisition voltage, and output a first trigger signal when the first acquisition voltage is greater than the reference voltage;

[0104] The first trigger module 242 has a first input terminal receiving the first comparison signal and a second input terminal connected to the output terminal of the first comparison module 241 , and is configured to output a first weak current on / off control signal according to the first trigger signal.

[0105] Among them, the first comparison module 241 delays the first control signal, generates a first acquisition voltage, and monitors the magnitude relationship between the first acquisition voltage and the reference voltage in real time. When the first acquisition voltage exceeds the reference voltage, the first comparison module 241 outputs a first trigger signal. The first trigger module 242 detects the state of the first trigger signal, and activates the trigger logic when a valid first trigger signal is received. According to the trigger logic, the first weak current on-off control signal is output to instruct the subsequent circuit module to perform the on-off control of the weak current. When the trigger signal is invalid or does not meet the conditions, the control signal is not output to ensure that the circuit is in a safe state.

[0106] The technical effect of this embodiment is that through the collaborative work of the first comparison module and the first trigger module, the delay processing and precise trigger control of the first control signal are realized. The first comparison module converts the first control signal into a first acquisition voltage, and monitors its magnitude relationship with the reference voltage in real time, and outputs the first trigger signal only when the trigger condition is met. The first trigger module generates a first weak current on-off control signal according to the state of the trigger signal to ensure that the subsequent circuit module performs the on-off operation of the weak current at the appropriate time. This design has technical effects such as accurate delay, rapid response and high safety, which effectively avoids false triggering and ensures the stability and reliability of the circuit.

[0107] As an implementation method, Figure 7 As shown, the second control submodule 212 includes:

[0108] A second delay trigger module 205, whose first input terminal receives a second control signal, whose second input terminal receives a reference voltage, and whose third input terminal receives a first comparison signal, is used to output a second weak current on-off control signal according to the second control signal, the reference voltage and the first comparison signal;

[0109] The second driving module 206 has an input end connected to the output end of the second delay trigger module, and is used to amplify and isolate the output second weak current on-off control signal and then output the second on-off control signal.

[0110] Among them, the main function of the second delay trigger module 205 is to output the second weak current on-off control signal at an appropriate time according to the input second control signal, the reference voltage and the first comparison signal. The second control signal is used to indicate the trigger logic, which is usually triggered when the level state changes. The reference voltage is used as a reference voltage to compare with the internally generated voltage to ensure that the circuit operates within a predetermined voltage range. The first comparison signal is used to assist in determining whether the trigger condition is met, delay the second control signal, generate the second acquisition voltage, and monitor the magnitude relationship between the second acquisition voltage and the reference voltage in real time. When the second acquisition voltage exceeds the reference voltage and the first comparison signal meets the trigger condition, the second weak current on-off control signal is output to control the operation of the subsequent circuit module. The main function of the second driver module 206 is to receive the second weak current on-off control signal output by the second delay trigger module 205, amplify and electrically isolate the signal, and output the second on-off control signal to drive the subsequent circuit. The second driver module 206 outputs the processed second on-off control signal to control the on-off operation of the fourth switch module 104.

[0111] The technical effect of this embodiment is that through the coordinated work of the second delay trigger module and the second drive module, the precise on-off control and safe driving of the fourth switch module are realized. The second delay trigger module is based on the delay processing and multi-signal judgment mechanism, and only outputs a reliable weak current on-off control signal when the trigger condition is met, ensuring the accuracy and safety of the control. The second drive module amplifies and electrically isolates the weak current signal, converting it into a high current or high voltage signal that can drive the power device, while avoiding the interference of high voltage or large current on the control circuit. This design effectively improves the driving ability, anti-interference and stability of the control signal, and realizes efficient and safe circuit operation.

[0112] As an implementation method, Figure 8 As shown, the second delay trigger module 205 includes:

[0113] A second comparison module 243, whose first input terminal receives the second control signal and whose second input terminal receives the reference voltage, is used to delay the conversion of the second control signal into the second collection voltage, and output a second trigger signal when the second collection voltage is greater than the reference voltage;

[0114] The second trigger module 244 has a first input terminal receiving the first comparison signal and a second input terminal connected to the output terminal of the second comparison module, and is used to output a second weak current on-off control signal according to the second trigger signal.

[0115] Among them, the main function of the second comparison module 243 is to compare the input second control signal with the reference voltage, and generate a second acquisition voltage according to the comparison result, which is used to trigger the subsequent control signal output. The second control signal is used as the input signal controlled by the second switch module 102 to indicate whether a delay trigger is required, which is usually triggered when the level state changes. The reference voltage provides a reference voltage for comparison with the voltage converted from the second control signal. The second comparison module 243 converts the received second control signal into the second acquisition voltage through delay processing. The delay processing is used to ensure the timing accuracy of the system and avoid control instability caused by too fast signal changes. The converted second acquisition voltage is compared with the reference voltage. If the second acquisition voltage is greater than the reference voltage, it is considered that the system meets the trigger condition, and the module will output the second trigger signal. If the condition is not met, the subsequent operation will not be triggered. The main function of the second trigger module 244 is to further output a second weak current on-off control signal for controlling the subsequent circuit according to the second trigger signal and the first comparison signal output by the second comparison module 243, which is used to control the on-off state of the fourth switch module 104 in the subsequent circuit.

[0116] The technical effect of this embodiment is that the delay processing and precise trigger control of the second control signal are realized through the coordinated work of the second comparison module and the second trigger module. The second comparison module converts the second control signal into the second acquisition voltage, and monitors its magnitude relationship with the reference voltage in real time, and outputs the second trigger signal only when the trigger condition is met; the second trigger module generates a second weak current on-off control signal according to the state of the trigger signal to ensure that the subsequent circuit module performs the on-off operation of the weak current at the appropriate time. This design has the technical effects of precise delay, rapid response and high safety, effectively avoiding false triggering and ensuring the stability and reliability of the circuit.

[0117] As an implementation manner, the first switch module 101 is a first MOS tube, the second switch module 102 is a second MOS tube, the first control signal includes a first voltage control signal and a second voltage control signal, and the second control signal includes a third voltage control signal and a fourth voltage control signal;

[0118] The resonant LLC switching power converter further includes:

[0119] A first voltage dividing unit, wherein a first end thereof receives a first voltage control signal, a second end thereof is connected to the gate of the first MOS tube, a third end thereof is connected to the source of the first MOS tube and then receives a second voltage control signal, and controls the on and off of the first MOS tube according to the first voltage control signal and the second voltage control signal;

[0120] A second voltage dividing unit, wherein a first end thereof receives a third voltage control signal, a second end thereof is connected to the gate of the second MOS tube, a third end thereof is connected to the source of the second MOS tube and then receives a fourth voltage control signal, and controls the on and off of the second MOS tube according to the third voltage control signal and the fourth voltage control signal;

[0121] The second control module 112 is used to control the third switch module to start conducting when it is detected that the output current is greater than the preset current and a high level signal is formed between the first voltage control signal and the second voltage control signal or a high level signal is formed between the second voltage control signal and the ground terminal of the second control module;

[0122] The second control module 112 is further configured to control the fourth switch module to start conducting when the output current is greater than a preset current and a high level signal is formed between the third voltage control signal and the fourth voltage control signal.

[0123] Among them, the first voltage divider unit is used to control the gate voltage of the first MOS tube according to the input first voltage control signal and the second voltage control signal, thereby adjusting the on-off state of the first MOS tube. The first end of the first voltage divider unit receives the first voltage control signal as a control input. The first voltage control signal and the second voltage control signal work together to adjust the gate voltage of the first MOS tube through the voltage division principle and the current discharge effect of the voltage divider resistor. When a high-level signal is formed between the first voltage control signal and the second voltage control signal, the control signal triggers the first MOS tube to turn on; when the control signal becomes a low level, the first MOS tube is disconnected; when the control circuit is not powered on, the voltage divider resistor reliably discharges the gate capacitor voltage of the MOS tube to 0V. The second voltage divider unit adjusts the gate voltage of the second MOS tube according to the input third voltage control signal and the fourth voltage control signal, thereby controlling the on-off state of the second MOS tube. The first end of the second voltage divider unit receives the third voltage control signal as a control input. The third voltage control signal and the fourth voltage control signal work together to control the gate voltage of the second MOS tube through the voltage division method and the current discharge effect of the voltage divider resistor. When a high level signal is formed between the third voltage control signal and the fourth voltage control signal, the control signal triggers the second MOS tube to turn on; when the control signal becomes a low level, the second MOS tube is disconnected; when the control circuit is not powered on, the voltage divider resistor reliably discharges the gate capacitor voltage of the MOS tube to 0V. According to the above signal, the second voltage divider unit accurately controls the switch state (on or off) of the second MOS tube, and ensures that the MOS tube is in a reliable off state when not powered on. When the output current is greater than the preset current, the second control module 112 controls the conduction state of the third switch module and the fourth switch module according to the level state of each control signal. The second control module 112 first monitors the output current and compares it with the preset reference current. When the output current is greater than the preset current, it is determined that the load is in a high current state and is ready for subsequent dynamic control. When a high level signal is formed between the first voltage control signal and the second voltage control signal, or a high level signal is formed between the second voltage control signal and the ground terminal, the second control module 112 starts the conduction of the third switch module. This control ensures that when the load current is large, the third switch module 103 can work effectively and support the zero voltage conduction or shutdown of the first switch module 101. When a high level signal is formed between the third voltage control signal and the fourth voltage control signal, the second control module 112 will start the conduction of the fourth switch module 104. This operation ensures that the fourth switch module 104 can enter the conduction state under preset conditions and support the zero voltage conduction or shutdown of the second switch module 102.

[0124] The technical effect of this embodiment is that by accurately controlling the gate voltage of the first MOS tube and the second MOS tube, efficient regulation and stability and reliability of the power converter are achieved. The first voltage divider unit and the second voltage divider unit respectively control the switching state of the first MOS tube and the second MOS tube to ensure accurate control of current flow. When the load current is large, the second control module can timely adjust the conduction state of the third switch module and the fourth switch module according to the feedback signal of the current, realize zero voltage conduction and disconnection of the first switch module and the second switch module, reduce the loss in the switching process, and optimize the power conversion efficiency.

[0125] The embodiment of the present invention is described in detail below through a specific circuit structure:

[0126] The first switch module 101 is a main switch transistor Q1, the second switch module 102 is a main switch transistor Q2, the third switch module 103 is an auxiliary switch transistor Q3, the fourth switch module 104 is an auxiliary switch transistor Q4, the first capacitor module 105 is a resonant capacitor C2, the second capacitor module 106 is an auxiliary capacitor C4, the third capacitor module 107 is an auxiliary capacitor C5, the energy storage isolation module includes an inductor L1 and a transformer T1, and the resonant LLC switching power converter also includes: a resistor R1, a resistor R2, a resistor R3, a resistor R4, a resistor R5, a resistor R6, a resistor R7, a resistor R8, a capacitor C1, a capacitor C3, a diode D1 and a diode D2.

[0127] One end of the auxiliary capacitor C4 is connected to the drain D of the auxiliary switch transistor Q3, the source S of the auxiliary switch transistor Q3, the source S of the main switch transistor Q1, the drain D of the main switch transistor Q2, the positive electrode of the auxiliary capacitor C5, one end of the resistor R2, one end of the resistor R6, one end of the inductor L1, the second output end of the first control module 111 (output drive signal VS-Q1), the second input end of the second control module 112 (drive signal VS-Q1 voltage sampling end), the second auxiliary DC power supply negative electrode VCC2-GND, the second output end of the second control module 112 (output drive signal VS-Q3), the seventh input end of the second control module 112 (second power supply negative end), and the twelve points are connected together. The negative electrode of the auxiliary capacitor C5 is connected to the drain electrode D of the auxiliary switch transistor Q4, the other end of the resistor R2 is connected to the gate electrode G of the switch transistor Q1 and one end of the resistor R1, the other end of the resistor R6 is connected to the gate electrode G of the auxiliary switch transistor Q3 and one end of the resistor R5, the other end of the resistor R1 is connected to the first output end (output drive signal VG-Q1) of the first control module 111, and the other end of the resistor R5 is connected to the first output end (output drive signal VG-Q3) of the second control module 112; the other end of the inductor L1 is connected to one end of the primary coil of the transformer T1, and the other end of the primary coil of the transformer T1, one end of the resonant capacitor C2, the second input end (IS current sampling end) of the first control module 111, and the first input end (IS current sampling end) of the second control module 112 are connected together; the other end of the resonant capacitor C2, the source electrode S of the main switch transistor Q2, the source electrode S of the auxiliary switch transistor Q4, one end of the resistor R4, one end of the resistor R8, the negative electrode of the input capacitor C1, and the input DC The negative voltage electrode VI-GND, the negative electrode of the first auxiliary DC power supply VCC1-GND, the fourth output terminal of the first control module 111 (output drive signal VS-Q2), the fourth output terminal of the second control module 112 (output drive signal VS-Q4), the fifth input terminal of the first control module 111 (first power supply negative terminal), the fourth input terminal of the first control module 111 (first power supply negative terminal), and the fifth input terminal of the second control module 112 (first power supply negative terminal) are connected together; the other end of the resistor R4 is connected to the gate electrode G of the main switch transistor Q2 and one end of the resistor R3, and the other end of the resistor R8 is connected to the gate electrode G of the auxiliary switch transistor Q4 and one end of the resistor R7; the other end of the resistor R3 is connected to the third output terminal VG-Q2 of the first control module 111 and the third input terminal (VG-Q2 voltage sampling terminal) of the second control module 112, and the other end of the resistor R7 is connected to the third output terminal (output drive signal VG-Q4) of the second control module 112. The first auxiliary DC power positive electrode VCC1, the third input terminal of the first control module 111 (the first auxiliary DC power positive electrode VCC1), and the fourth input terminal of the second control module 112 (the first power supply positive terminal) are connected together.The positive electrode VCC2 of the second auxiliary DC power supply is connected to the sixth input terminal (the positive terminal of the second power supply) of the second control module 112. The first end of the secondary coil of the transformer T1 is connected to the positive electrode of the diode D1, the negative electrode of the diode D1, the negative electrode of the diode D2, the positive electrode of the output capacitor C3, the positive electrode VO of the output terminal of the switching power converter, and the fifth input terminal (the positive electrode of the output voltage sampling terminal) of the first control module 111 are connected together; the second end of the secondary coil of the transformer T1 is connected to the positive electrode of the diode D2, the negative electrode of the output capacitor C3, the third end of the secondary coil of the transformer T1 (the center tap of the coil), the negative electrode VO-GND of the output terminal of the switching power converter, and the sixth input terminal (the negative electrode of the output voltage sampling terminal) of the first control module 111, and the four points are connected together.

[0128] in, Fig. 9 and Fig.10The meanings of the signals in are as follows: U(GS-Q*) is the voltage difference waveform between the gate G and source S of the switching transistor Q*, and * represents a digital label; U(VS-Q1) is the voltage difference waveform between the source S circuit node A of the switching transistor Q1 (the signal received by this node is the drive signal VS-Q1) and VCC1-GND (0 point reference point); VG-Q1 is the first positive drive signal output by the first control module 111, which is connected to the gate G of the main switching transistor Q1 through the resistor R1; VS-Q1 is the first positive drive signal output by the first control module 111, which is connected to the gate G of the main switching transistor Q1 through the resistor R1; A first negative driving signal output by a control module 111 is connected to a source S of the main switching transistor Q1, which has the same potential as the negative electrode VCC2-GND of the second auxiliary power supply; VG-Q2 is a second positive driving signal output by the first control module 111, which is connected to a gate G of the main switching transistor Q2 through a resistor R3; VS-Q2 is a second negative driving signal output by the first control module 111, which is connected to a source S of the main switching transistor Q2, which has the same potential as the negative electrode VCC1-GND of the first auxiliary power supply. VG-Q3 is the third positive driving signal output by the second control module 112, which is connected to the gate G of the auxiliary switch transistor Q3 through the resistor R5; VS-Q3 is the third negative driving signal output by the second control module 112, which is connected to the source S of the auxiliary switch transistor Q3, and has the same potential as the first negative driving signal VS-Q1 and the second auxiliary power supply negative electrode VCC2-GND; VG-Q4 is the fourth positive driving signal output by the second control module 112, which is connected to the gate G of the auxiliary switch transistor Q4 through the resistor R7; VS-Q4 is the fourth negative driving signal output by the second control module 112, which is connected to the source S of the auxiliary switch transistor Q4, and has the same potential as the first auxiliary power supply negative electrode VCC1-GND; I(VO) is the current value flowing out of the output terminal VO of the switching power converter; VP is the output voltage signal waveform of the current detection module; IH represents a larger current value, and the driving pulse signal output by the switching power converter is a continuous pulse; IS represents a set current value. When the output current value I(VO) of the innovative resonant LLC switching power converter of the present invention is less than the set value IS, it operates in a small current load state 1. At this time, the drive pulse signal output by the first control module 111 is a continuous pulse or a discontinuous pulse group with a higher frequency. At this time, the voltage difference U(GS-Q3) and U(GS-Q3) between the G pole and the S pole of the auxiliary switching transistors Q3 and Q4 are both low levels (close to 0V); when the output current value I(VO) is greater than the set value IS, it operates in a large current load state 2. The drive pulse signal output by the first control module 111 is a continuous pulse with a lower frequency. At this time, the voltage difference U(GS-Q3) and U(GS-Q3) between the G pole and the S pole of the auxiliary switching transistors Q3 and Q4 are both high levels (close to their auxiliary power supply voltage values).IL represents a smaller current value, at which time the drive pulse signal output by the first control module 111 is a discontinuous pulse group or a continuous pulse with a higher frequency. VH represents a high signal level. Ton is the high level period of the drive signal voltage in a switching cycle T, that is, the on period of the switching transistor in a cycle T. Toff is the low level period of the drive signal voltage in a switching cycle T, that is, the off period of the switching transistor in a cycle T. 180° is the phase difference of the waveform of the first drive signal voltage difference U (GS-Q1) output by the first control module 111 and the second drive signal voltage difference U (GS-Q2) is 180°.

[0129] like Fig.11 As shown, the second control module 112 includes a current detection module 201, a reference voltage module 202, a first delay trigger module 203, a second delay trigger module 205, a first drive module 204 and a second drive module 206, wherein the input and output signals of each module are as follows:

[0130] The input signals of the current detection module 201 include: the voltage waveform IS across the resonant capacitor C2 (the amplitude of the AC component in this voltage waveform is proportional to the maximum current value flowing through the resonant capacitor C2), and the reference voltage V-REF, both of which use the first auxiliary DC power supply negative electrode VCC1-GND as the reference 0 point. Its output signal is the switching power converter output power state signal VP, which uses the first auxiliary DC power supply negative electrode VCC1-GND as the reference 0 point.

[0131] The output signal of the reference voltage module 202 includes: a reference voltage source V-REF generated by the positive electrode of the first auxiliary DC power supply VCC1 and stabilized by the reference voltage, and the negative electrode of the first auxiliary DC power supply VCC1-GND is used as the reference 0 point.

[0132] The input signals of the first delay trigger module 203 include: a reference voltage V-REF, a voltage difference between a signal VS-Q1 at a circuit node A of the source S of the main switch transistor Q1 and the negative electrode VCC1-GND of the first auxiliary DC power supply, and a power state signal VP output by the current detection module 201. The signals VS-Q1 and VP both use the negative electrode VCC1-GND of the auxiliary DC power supply as a reference 0 point. The output signal is a small current drive level signal V-Q3 of the auxiliary switch transistor Q3, which uses the negative electrode VCC1-GND of the first auxiliary DC power supply as a reference 0 point. The positive electrode of the working power supply is VCC1, and the negative electrode of the working power supply is VCC1-GND.

[0133] The input signals of the second delay trigger module 205 include: reference voltage V-REF, driving voltage signal VG-Q2 of the main switch transistor Q2, output power state signal VP of the current detection module 201, and both signal VG-Q2 and signal VP use the negative electrode VCC1-GND of the first auxiliary DC power supply as reference 0 point. Its output signal is a small current driving level signal V-Q4 of the auxiliary switch transistor Q4, using the negative electrode VCC1-GND of the first auxiliary DC power supply as reference 0 point. Its working power supply positive electrode is VCC1, and its working power supply negative electrode is VCC1-GND.

[0134] The input signal of the first driving module 204 is the small current driving level signal V-Q3 of the auxiliary switching transistor Q3, and its output signal is the large current driving voltage signal VG-Q3 of the auxiliary switching transistor Q3, with the negative electrode VCC2-GND of the second auxiliary DC power supply (or circuit nodes VS-Q3, VS-Q1) as the 0 point reference point.

[0135] The input signal of the second driving module 206 is a small current driving signal level signal of the auxiliary switch transistor Q4, and the output signal is a large current driving voltage signal VG-Q4 of the auxiliary switch transistor Q4, with the first auxiliary DC power supply negative electrode VCC1-GND as the 0 point reference point.

[0136] Among them, VCC1 is the positive electrode of the first auxiliary DC power supply; VCC1-GND is the negative electrode of the first auxiliary DC power supply; VCC2 is the positive electrode of the second auxiliary DC power supply; VCC2-GND is the negative electrode of the second auxiliary DC power supply.

[0137] exist Fig. 9 Circuit and Fig.11 In the circuit block diagram shown, the positive electrode of the resonant capacitor C2 is connected to the first input end of the current detection module 201, and the current signal IS is input. The output end of the current detection module 201 is connected to the first input end of the first delay trigger module 203 and the first input end of the second delay trigger module 205, and the voltage signal waveform VP is output.

[0138] The second input end of the first delay trigger module 203 is connected to the source S circuit node of the main switch transistor Q1, and receives the first negative drive signal VS-Q1; the third input end of the first delay trigger module 203, the third input end of the second delay trigger module 205, and the output end of the reference voltage module 202 are connected together, and receive the reference voltage signal V-REF; the output end of the first delay trigger module 203 is connected to the first input end of the first drive module 204, and outputs a small current drive level signal V-Q3; the first output end of the first drive module 204 is connected to one end of the resistor R5, and the other end of the resistor R5 is connected to the gate G of the auxiliary switch transistor Q3, and outputs the third positive drive signal VG-Q3; the second output end of the first drive module 204 is connected to the source S of the auxiliary switch transistor Q3, and outputs the third negative drive signal VS-Q3, the positive pole of the second auxiliary power supply is connected to the second input end of the first drive module 204, and outputs the power supply signal VCC2; the negative pole of the second auxiliary power supply VCC2-GND is connected to the third input end of the first drive module 204.

[0139] The second input end of the second delay trigger module 205 is connected to the main switch transistor Q2 driving voltage output end of the first control module 111, and receives the second positive driving signal VG-Q2; the output end of the second delay trigger module 205 is connected to the first input end of the second driving module 206, and outputs a small current driving level signal V-Q4; the first output end of the second driving module 206 is connected to one end of the resistor R7, and outputs the fourth positive driving signal VG-Q4, and the other end of the resistor R7 is connected to the gate G of the auxiliary switching transistor Q4, and the second output end of the second driving module 206 is connected to the negative electrode VI-GND of the input DC voltage, the negative electrode VCC1-GND of the auxiliary power supply, and the source S of the auxiliary switching transistor Q4, and receives the fourth negative driving signal VS-Q4.

[0140] The positive pole of the first auxiliary power supply is connected to the second input terminal of the current detection module 201, the first input terminal of the reference voltage module 202, the fourth input terminal of the first delay trigger module 203, the fourth input terminal of the second delay trigger module 205, the fourth input terminal of the first driving module 204, and the second input terminal of the second driving module 206, and outputs a power signal.

[0141] The negative pole VCC-GND of the first auxiliary power supply is connected to the third input terminal of the current detection module 201, the second input terminal of the reference voltage module 202, the fifth input terminal of the first delay trigger module 203, the fifth input terminal of the second delay trigger module 205, the fifth input terminal of the first driving module 204, and the third input terminal of the second driving module 206.

[0142] The components in the current detection module 201 include: a comparator IC201 , a resistor R201 , a resistor R202 , a resistor R203 , a resistor R204 , a resistor R205 , a resistor R206 , a capacitor C201 , a capacitor C202 , and a capacitor C203 . The connection method is as follows: one end of capacitor C201 is connected to the current signal IS, the other end of capacitor C201 is connected to five points, one end of resistor R201, one end of resistor R202, one end of resistor R203, and one end of capacitor C202; the other end of resistor R201 is connected to three points, one end of resistor R204 and the non-inverting input terminal of comparator IC201; the other end of resistor R202 is connected to the negative electrode of the auxiliary first auxiliary power supply VCC1-GND (reference 0 potential point); the other end of resistor R203 and one end of resistor R205 are connected to the reference voltage V-REF; the other end of resistor R205 is connected to three points, one end of resistor R206 and the inverting input terminal of comparator IC201; the other end of resistor R204 is connected to the cathode of diode D201; the anode of diode D201 is connected to the output terminal of comparator IC201, and outputs signal VP. The positive power terminal of the comparator IC201 and one end of the capacitor C203 are connected to the first auxiliary power positive electrode and receive the power signal VCC1. The negative power terminal of the comparator IC201 and the other end of the capacitor C203 are connected to the first auxiliary power negative electrode VCC1-GND.

[0143] The components in the reference voltage module 202 include: a resistor R207, a three-terminal voltage regulator IC202, and a capacitor C204. The connection method is as follows: one end of the resistor R207 is connected to the positive electrode of the first auxiliary power supply to receive the power supply signal VCC1, the other end of the resistor R207, the cathode of the three-terminal voltage regulator IC202, and one end of the capacitor C204 are connected together to output the reference voltage signal V-REF, and the other end of the capacitor C204 and the anode of IC202 are connected together to the negative electrode of the first auxiliary power supply VCC1-GND.

[0144] The components in the first delay trigger module 203 include: comparator IC204, D trigger IC203, resistor R208, resistor R209, resistor R210, capacitor C205, capacitor C206, and capacitor C207. The connection method is as follows: one end of the resistor R208 receives the first negative driving signal VS-Q1, the other end of the resistor R208 is connected to the resistor R209, the resistor R210, one end of the capacitor C205, and the in-phase input end of the comparator IC204, the resistor R209 and the other end of the capacitor C205, the trigger IC203 set 0 end R and set 1 end S, the negative power supply of IC203, one end of the capacitor C206, one end of the capacitor C207, and the negative power supply of the comparator IC204 are connected to the first auxiliary power supply negative electrode VCC1-GND, and the other end of the resistor R210 is connected to the diode D2 The cathode of D202, the anode of diode D202, the output terminal of comparator IC204 and the clock input terminal C of D flip-flop IC203 are connected together to receive the signal V-T1. The data input terminal D of D flip-flop IC203 is connected to the output terminal of comparator IC201 in current detection module 201 to receive the signal VP. The other end of capacitor C206, the other end of C207 and the positive power supply terminals of comparator IC204 and D flip-flop IC203 are connected together to the positive pole of the first auxiliary power supply to receive the power supply signal VCC1. The data output terminal Q of D flip-flop IC203 is connected together to output a small current drive level signal V-Q3.

[0145] The components in the first driving module 204 include: resistors R211 , R212 , R213 , R214 , R215 , R216 , R217 , NPN switch transistors Q203 , Q205 , PNP switch transistors Q204 , Q206 , and capacitor C208 . The connection method is as follows: one end of the resistor R211 receives the small current driving level signal V-Q3, the other end of the resistor R211 is commonly connected to three points: one end of the resistor R212 and the base B of the NPN type switching transistor Q205, the emitter E of the NPN type switching transistor Q205 and the other end of the resistor R211 are commonly connected to the negative electrode VCC1-GND of the first auxiliary power supply, the collector C of the NPN type switching transistor Q205 is connected to one end of the resistor R213, the other end of the resistor R213 is commonly connected to three points: the base B of the PNP type switching transistor Q206 and one end of the resistor R214, the other end of the resistor R214, the emitter E of the PNP type switching transistor Q206, the collector C of the NPN type switching transistor Q203, and one end of the capacitor C208 are commonly connected To the positive electrode of the second auxiliary power supply, receiving the power supply signal VCC2, the collector C of the PNP switching transistor Q206 is connected to one end of the resistor R215, the other end of the resistor R215, the base B of the PNP switching transistor Q204, the base B of the NPN switching transistor Q203, and one end of the resistor R216 are connected together, the emitter E of the PNP switching transistor Q204 and the emitter E of the NPN switching transistor Q203 and one end of the resistor R217 are connected together, the other end of the resistor R217 is the output end, and the third positive driving signal VG-Q3 is output, the collector C of the PNP switching transistor Q204 and the other end of the resistor R216 are connected together to the negative electrode VCC2-GND of the second auxiliary power supply, and the third negative driving signal VS-Q3 is output.

[0146] The components in the second delay trigger module 205 include: comparator IC205, D trigger IC206, resistor R218, resistor R219, resistor R220, capacitor C209, capacitor C210, and capacitor C211. The connection method is as follows: one end of the resistor R218 receives the second positive driving signal VG-Q2, the other end of the resistor R218 is connected to five points, namely, one end of the resistor R219, one end of the resistor R220, one end of the capacitor C209, and the in-phase input end of the comparator IC205; the other end of the resistor R219, the other end of the capacitor C209, one end of the capacitor C210, one end of C211, the negative power supply of the comparator IC205, the set 0 end R and the set 1 end S of the D trigger IC206, and the negative power supply of IC206 are connected to the negative pole of the first auxiliary power supply VCC1-GND, and the other end of the resistor R220 is connected to The cathode of the diode D203 is connected, and the anode of the diode D203, the output terminal of the comparator IC205, and the clock input terminal C of the D flip-flop IC206 are connected together to output the signal V-T2. The data input terminal D of the D flip-flop IC206 is connected to the output terminal of the comparator IC201 in the current detection module 201 to receive the signal VP. The other end of the capacitor C210 and the other end of C211, the positive power supply terminals of the comparator IC205 and the D flip-flop IC206 are connected together to the positive terminal of the first auxiliary power supply to receive the power supply signal VCC1. The data output terminal Q of the D flip-flop IC206 outputs a small current drive level signal V-Q4.

[0147] The components in the second driving module 206 include: a resistor R221 , a resistor R222 , a resistor R223 , an NPN switch transistor Q201 , a PNP switch transistor Q202 , and a capacitor C212 . The connection method is as follows: one end of the resistor R221 receives the small current driving level signal V-Q4, the other end of the resistor R221, the base B of the PNP switching transistor Q202, the base B of the NPN switching transistor Q201, and one end of the resistor R222 are connected together, the emitter E of the NPN switching transistor Q201, the emitter E of the PNP switching transistor Q202 and one end of the resistor R223 are connected together, the other end of the resistor R223 is connected to output the fourth positive driving signal VG-Q4, the collector C of the NPN switching transistor Q201 is connected to the positive electrode of the first auxiliary power supply, and receives the power supply signal VCC1, the collector C of the PNP switching transistor Q202 and the other end of the resistor R216 are connected to the negative electrode VCC1-GND of the first auxiliary power supply, and output the fourth negative driving signal VS-Q4.

[0148] The first control module 111 in the innovative resonant LLC switching power converter of the present invention outputs driving signals of two main switching transistors Q1 and Q2, and the reference 0 potentials of the two main driving signals are different. The first positive driving signal is VG-Q1, and the first negative driving signal is VS-Q1, which is the same as the potential of the negative electrode VCC2-GND of the second auxiliary DC power supply; the second positive driving signal is VG-Q2, and the second negative driving signal is VS-Q2, which is the same as the potential of the negative electrode VCC1-GND of the first auxiliary DC power supply.

[0149] The second control module 112 innovatively added in the present invention outputs drive signals of two auxiliary switching transistors Q3 and Q4, and the reference 0 potentials of these two auxiliary drive signals are different, wherein the third positive drive signal is VG-Q3, and the third negative drive signal is VS-Q3, which has the same potential as the first negative drive signal VS-Q1 and the second auxiliary DC power supply negative electrode VCC2-GND; wherein the fourth positive drive signal is VG-Q4, and the fourth negative drive signal is VS-Q4, which has the same potential as the second auxiliary DC power supply negative electrode VCC1-GND.

[0150] like Fig.10 As shown, the first drive signal waveform U(GS-Q1) (i.e., the voltage difference between the first positive drive signal VG-Q1 and the first negative drive signal VS-Q1) output by the first control module 111 and the second drive signal waveform U(GS-Q2) (i.e., the voltage difference between the second positive drive signal VG-Q2 and the second negative drive signal VS-Q2) have a phase difference of 180° within one switching cycle.

[0151] like Fig. 9 , Fig.10 As shown, the third driving signal waveform U(GS-Q3) output by the second control module 112 is converted from a low level to a high level, and at the same time, the auxiliary switch tube Q3 is triggered to turn on, which requires two necessary conditions:

[0152] The first condition: the LLC power converter output current value I(VO) is a large current value IH, such as Fig.10 As shown, the current value IH must meet the condition: IH is greater than the set current value IS, that is, IH>IS. At this time, the drive signal waveforms U(GS-Q1) and U(GS-Q2) output by the first control module 111 are continuous square waves.

[0153] The second condition: the first drive signal waveform U(GS-Q1) is at a high level or the main switch transistor Q1 source drive signal VS-Q1 is at a high voltage (reference 0 potential is VCC1-GND). This condition is only used as the starting trigger condition for U(GS-Q3) to convert from a low level to a high level. When U(GS-Q1) maintains a high level, this second condition is no longer used as a basis for judgment.

[0154] Same as above, Fig.10 As shown, the fourth drive signal waveform U(GS-Q4) output by the second control module 112 is converted from a low level to a high level, and at the same time, the auxiliary switch tube Q4 is triggered to turn on, which requires two necessary conditions:

[0155] The first condition: the converter output current value I(VO) is a large current value IH, such as Figure 4 As shown, the current value IH must meet the condition: IH is greater than the set current value IS, that is, IH>IS. At this time, the drive signal waveforms U(GS-Q1) and U(GS-Q2) output by the first control module 111 are continuous square wave waveforms.

[0156] The second condition: The second drive signal waveform U(GS-Q2) is at a high level. This condition is only used as a trigger condition for the start of the transition of U(GS-Q4) from a low level to a high level. When U(GS-Q4) maintains a high level, this second condition is no longer used as a basis for judgment.

[0157] As mentioned above, when the power converter enters the high current load state 2 from the low current load state 1, the auxiliary switch transistors Q3 and Q4 cannot be turned on immediately. At this time, the second control module 112 also needs to judge the state of the main switch transistors Q1 and Q2, and wait for the main switch transistors Q1 and Q2 to enter a stable on state before immediately triggering the auxiliary switch transistors Q3 and Q4 to turn on. After the auxiliary switch transistors Q3 and Q4 are triggered to turn on, they will be locked in the on state. At this time, only when the following conditions for triggering the auxiliary switch transistors Q3 and Q4 to turn off are met can they be switched to the off state. Therefore, the auxiliary switch transistors Q3 and Q4 both realize a high-efficiency zero-voltage soft turn-on function, and the turning on of the auxiliary switch transistors Q3 and Q4 does not interfere with the main circuit.

[0158] like Fig.10 As shown, the third driving signal waveform U(GS-Q3) output by the second control module 112 is converted from a high level to a low level, and at the same time, the auxiliary switch tube Q3 is triggered to turn off, which requires two necessary conditions:

[0159] The first condition: the output current value I(VO) of the power converter is a small current value IL, such as Fig.10As shown, the current value I-L needs to satisfy the condition that I-L is less than the set current value I-S, i.e., I-L < I-S. At this time, the driving signal waveforms U(GS-Q1) and U(GS-Q2) output by the first control module 111 are intermittent square wave groups (i.e., the case where the low-level time interval repeatedly appears in the square wave and exceeds one square wave period) or continuous square waves with a relatively high frequency.

[0160] Second condition: The first driving signal waveform U(GS-Q1) is at a high level or the source potential VS-Q1 of the switching transistor Q1 is at a high voltage (referring to the reference 0 potential as VCC1-GND). This condition only serves as the trigger condition for the starting moment when U(GS-Q3) changes from a high level to a low level. When U(GS-Q1) maintains a low level, this second condition is no longer used as the judgment basis.

[0161] Similarly, as Fig.10 shown, for the fourth driving signal waveform U(GS-Q4) output by the second control module 112 to change from a high level to a low level and at the same time the auxiliary switching transistor Q4 to be triggered to turn off, two necessary conditions are required:

[0162] First condition: The output current value I(VO) of the power converter is a small current value I-L. As Fig.10 shown, the current value I-L needs to satisfy the condition that I-L is less than the set current value I-S, i.e., I-L < I-S. At this time, the driving signal waveforms U(GS-Q1) and U(GS-Q2) output by the first control module 111 are intermittent square wave groups (i.e., the case where the low-level time interval repeatedly appears in the square wave and exceeds one square wave period) or continuous square waves with a relatively high frequency.

[0163] Second condition: The second driving signal waveform U(GS-Q2) is at a high level. This condition only serves as the trigger condition for the starting moment when U(GS-Q4) changes from a high level to a low level. When U(GS-Q4) maintains a low level, this second condition is no longer used as the judgment basis.

[0164] As above, when the power converter enters the "small current load state 1" from the "large current load state 2", it cannot immediately turn off the auxiliary switching transistors Q3 and Q4. At this time, the second control module 112 still needs to judge the states of the main switching transistors Q1 and Q2 and wait until the main switching transistors Q1 and Q2 enter a stable on state before immediately triggering the auxiliary switching transistors Q3 and Q4 to turn off. After the auxiliary switching transistors Q3 and Q4 are triggered to turn off, they will be locked in the off state. At this time, only when the above conditions for triggering the auxiliary switching transistors Q3 and Q4 to conduct are met can they be switched to the on state. Therefore, both the auxiliary switching transistors Q3 and Q4 achieve a high-efficiency zero-voltage soft turn-off function, and the turn-off of the auxiliary switching transistors Q3 and Q4 does not cause any interference to the main circuit.

[0165] The meaning of the component symbols in the above schematic diagram includes the series and parallel combinations of physical single components and physical components of the same nature.

[0166] Some components in the schematic diagram can use extreme values, for example:

[0167] a. The resistance values ​​of resistors R2, R4, R6, R8, R204, R206, R209, R210, R212, R214, R216, R219, R220, and R222 can be infinite, which is equivalent to an open circuit and can be removed. The resistor symbols do not need to be drawn in the schematic diagram.

[0168] b. The resistance values ​​of resistors R1, R3, R5, R7, R201, R205, R208, R215, R217, R218, and R223 can be 0Ω, and the symbols of the resistors can be replaced by short-circuited connecting wires.

[0169] c. Diodes D201, D202, and D203 can be removed, which is equivalent to a short circuit, and replaced with connecting wires.

[0170] d. The capacitance of capacitor C203, capacitor C204, capacitor C205, capacitor C206, capacitor C207, capacitor C208, capacitor C209, capacitor C210, capacitor C211, and capacitor C212 can be 0F. In this case, the capacitor is equivalent to an open circuit and the capacitor symbol may not be drawn in the schematic diagram.

[0171] A resonant LLC switching power converter innovatively provided by the present invention is Fig. 9 , Fig.11 , Fig.12 The working principle of the implementation circuit shown is described as follows:

[0172] like Fig. 9 As shown, the core control integrated circuit IC of the first control module 111 in the innovative resonant LLC switching power converter of the present invention can be implemented by using an existing LLC switching power converter dedicated control IC, or by writing a software program using an existing general-purpose digital signal processor DSP.

[0173] like Fig.12 As shown, the present invention is based on Fig.11 The principle analysis of a detailed implementation circuit of the second control module 112 of the block diagram is as follows:

[0174] (1) The working principle of the current detection module 201 is described as follows:

[0175] Since the effective value of the AC component in the voltage waveform across the resonant capacitor C2 of the resonant LLC switching power converter is proportional to the effective value of the current flowing through the capacitor C2, the current value of the LLC switching power converter resonant cavity can be expressed by detecting the effective value of the AC component of the voltage across the capacitor C2, as shown in Formula 1:

[0176] When U(GS-Q1) is high level and U(GS-Q2) is low level:

[0177] I Q1-ON =I L1 =I T1-1 =I C2 =C2*dV C2 / dt Formula 1

[0178] When U(GS-Q1) is low level and U(GS-Q2) is high level:

[0179] I Q2-ON =I L1 =I T1-1 =I C2 =C2*dV C2 / dt Formula 1

[0180] The meaning of symbols in formula 1: C* is the capacitance value of capacitor C*; * is any positive integer or letter; I Q1-ON is the instantaneous value of the current flowing through the drain D and source S of the main switch transistor Q1 when the voltage U(GS-Q1) is at a high level and the main switch transistor Q1 is turned on, with the current direction flowing into the drain D being the positive direction; I Q2-ON is the instantaneous value of the current flowing through the drain D and source S of the main switch transistor Q2 when the voltage U(GS-Q2) is at a high level and the main switch transistor Q2 is turned on, with the current direction flowing into the drain D being the positive direction; I L1 is the instantaneous value of the current flowing through the inductor L1, with the current flowing out from one end of the inductor L1 and into one end of the primary coil of the transformer T1 as the positive direction; I T1-1 is the instantaneous value of the current flowing through the primary coil of transformer T1, with the current flowing out from one end of inductor L1 and flowing into one end of the primary coil of transformer T1 as the positive direction; I C2 is the instantaneous value of the current flowing through capacitor C2, with the current flowing out of one end of the primary coil of transformer T1 and flowing into the positive voltage end of capacitor C2 as the positive direction; dV C2 / dt is the rate of change of the instantaneous value of the voltage across capacitor C2. An increase in voltage is a positive rate of change, and a decrease in voltage is a negative rate of change.

[0181] Applying Fourier analysis in the above formula 1, the resonant cavity loop current I C2 It can be approximately expressed as a sine wave I C2-MAX*sin(ωt), and substituting it into Formula 1, we can get the approximate expression of the AC component waveform in the voltage waveform across capacitor C2, as shown in Formula 2:

[0182]

[0183] From formula 2, we can know that the maximum voltage amplitude V of the AC component in the voltage waveform across capacitor C2 is C2-AC-MAX The maximum current amplitude I of the current waveform of the resonant cavity loop C2-MAX A fixed ratio relationship.

[0184] like Fig.12 As shown, the capacitor C201, the capacitor C202, and the resistor R202 in the current detection circuit 201 form an AC voltage divider circuit, wherein the capacitive reactances of the capacitor C201 and the capacitor C202 are X C201 =1 / (2πf W *C201),X C202 =1 / (2πf W *C202), this voltage divider circuit makes the AC voltage amplitude V generated at both ends of capacitor C202 C202-AC-MAX The AC component amplitude V in the voltage waveform across capacitor C2 C2-AC-MAX A fixed ratio relationship.

[0185] like Fig.12 As shown, the resistor R215 and the resistor R202 form a DC reference voltage divider circuit. Since the resistor R202 is connected in parallel with the capacitor C202, this DC voltage is the DC component of the voltage waveform across the capacitor C202.

[0186] Therefore, as mentioned above, the voltage across capacitor C202 (resistor R202) is formed by the superposition of DC component voltage and AC component voltage, and the composite voltage is calculated as shown in the following formula 3:

[0187]

[0188] Meaning of symbols in formula 2-3: X C201 is the capacitor C201 at the operating switching frequency f W Capacitive reactance value; X C202 is the capacitor C202 at the operating switching frequency f W The capacitive reactance value; f W is the operating switching frequency value of the resonant LLC switching power converter; ΔV C2 is the instantaneous value of the AC component of the voltage waveform across capacitor C2; ΔV C2-MAX V is the maximum value of the AC component of the voltage waveform across capacitor C2; C202-MAX is the maximum amplitude of the voltage waveform across capacitor C202; V C202-DCV is the DC component voltage value of the voltage waveform across capacitor C202; C202-AC-MAX V is the maximum voltage amplitude of the AC component of the voltage waveform across capacitor C202; C202-MAX is the maximum amplitude of the voltage waveform across capacitor C202; I C2-MAX is the current flowing through capacitor C2, i.e., the maximum amplitude of the resonant cavity current waveform of the resonant LLC switching power converter; U V-REF It is the voltage difference between the positive terminal of the reference voltage V-REF and the reference 0 point voltage VCC1-GND.

[0189] Since there is no current detection element connected in series in the main resonant cavity loop, the above current detection method does not increase the significant power consumption, and the volume is smaller, and the printed circuit board PCB layout is easier, so that this current detection method is widely used. Various other current detection methods in the prior art, such as the series resistance direct current detection method, the current transformer isolation detection method, etc., can also be applied to the second control module 112 of the innovative resonant LLC switching power converter of the present invention, which will not be repeated here.

[0190] The working principle of the reference voltage module 202 is described as follows: the positive electrode VCC of the auxiliary power supply supplies power to the three-terminal voltage regulator IC202 through the resistor R207, the reference electrode R of the three-terminal voltage regulator IC202 is connected to the cathode K, and the anode A of the three-terminal voltage regulator IC202 is connected to the negative electrode VCC1-GND (reference 0 potential) of the auxiliary power supply, so that the cathode K of the three-terminal voltage regulator IC202 generates a reference voltage V-REF that has been stabilized and regulated relative to the reference 0 potential VCC1-GND.

[0191] The working principle of the first delay trigger module 203 is as follows: Figure 3 , Figure 6 As shown, the high voltage input signal U between the node VS-Q1 of the resonant LLC switching power converter circuit and the 0 potential reference point VCC1-GND of the present invention is VS-Q1 After the voltage is divided by resistors R208 and R209, a low voltage waveform is formed in proportion, as calculated in formula 4:

[0192] U R209 =U VS-Q1 *R 209 (R 208 +R 209 ) Formula 4

[0193] like Fig.10 and Fig.12 As shown, when the first driving signal waveform U(GS-Q1) output by the first control module 111 is converted from a low level to a high level, the main switch transistor Q1 is converted from an off state to an on state. At this time, U VS-Q1 ≈UVI By selecting appropriate resistance values ​​of resistors R208 and R209, the voltage across resistor R209 meets the condition: U VCC >U R209 >U V-REF At this time, the output voltage V-T1 of the comparator IC204 is converted from a low level to a high level, and the trigger IC203 starts to trigger at the rising edge of the V-T1 voltage waveform, transmits the level state of the signal VP input at its input end to the output end and outputs a small current drive level signal V-Q3.

[0194] When the signal VP is at a low level, the small current drive level signal V-Q3 is also converted to a low level at this time, indicating that the power converter is in a small current load state 1. When the signal VP is at a high level, the small current drive level signal V-Q3 is also converted to a high level at this time, indicating that the converter is in a large current load state 2. And before the next rising edge of the V-T1 signal appears, the current output state of the small current drive level signal V-Q3 will be locked.

[0195] When the first driving signal waveform U(GS-Q1) output by the first control module 111 changes from a high level to a low level, the main switch transistor Q1 changes from an on state to an off state. VS-Q1 ≈0V, U R209 ≈0V. At this time, the voltage of the signal V-T1 outputted from the output terminal of the comparator IC204 is converted from a high level to a low level, and at this time, the trigger IC203 still locks the current output state of the small current driving level signal V-Q3.

[0196] Resistor R210 and diode D202 form a positive feedback hysteresis circuit of comparator IC204. When the output signal V-T1 of IC204 is high, the current flows through the positive and negative electrodes of diode D202 and the two ends of resistor R210 into the two ends of resistor R209, making the voltage value U R209 The comparator IC204 can stabilize the high level state of its output signal V-T1 even when it is disturbed.

[0197] The working principle of the first driving module 204 is described as follows: Fig. 9 , Fig.12As shown, when the low current driving level signal V-Q3 is converted from a low level to a high level, the current flows out from the output pin of the trigger IC203 in the first delay trigger module 203, and is divided into two paths after passing through the resistor R211, one of which flows into the base B of the NPN switch transistor Q205, and then flows into the negative electrode of the first auxiliary power supply VCC1-GND through the collector C; the other flows into the negative electrode of the first auxiliary power supply VCC1-GND through the resistor R212. At this time, the NPN switch transistor Q205 is converted from the off state to the on state.

[0198] As described above, due to the current amplification function of the NPN switch transistor Q205, a current larger than the current flowing into its base B flows out from the positive electrode of the second auxiliary power supply and is divided into two paths, one path flows into the emitter E of the PNP switch transistor Q206, and then flows into the resistor R213 through its base B; the other path flows into the resistor R213 through the resistor R214. This current flows into the collector C of the NPN switch transistor Q205 through the resistor R213, and then flows into the negative electrode VCC1-GND of the first auxiliary power supply through its emitter E. At this time, the PNP switch transistor Q206 is switched from the off state to the on state.

[0199] As described above, due to the current amplification function of the PNP switch transistor Q206, a current larger than the current flowing into its base B flows out from the second auxiliary power positive electrode VCC2, and flows into the base B of the NPN switch transistor Q203 through the emitter E of the PNP switch transistor Q206, the collector C of the PNP switch transistor Q206, and the resistor R215 in sequence, and then is divided into two paths, one of which flows into the resistor R217 through the emitter E, and the other flows into the second auxiliary power negative electrode VCC2-GND through the resistor R216. At this time, the NPN switch transistor Q203 is converted from the off state to the on state. Due to the current amplification function of the NPN switch transistor Q203, a current larger than the current flowing into its base B flows out from the second auxiliary power positive electrode VCC2, and flows into the resistor R217 through the collector C of the NPN switch transistor Q203 and the emitter E of the NPN switch transistor Q203, and outputs the third positive drive signal VG-Q3 through the resistor R217.

[0200] As mentioned above, the third positive driving signal VG-Q3 flows into the gate G of the NMOSFET auxiliary switch transistor Q3 and one end of the resistor R6 through the resistor R5, and then flows into the common connection point of the output terminal VS-Q3 and the negative electrode of the second auxiliary power supply VCC2-GND through the source S of the auxiliary switch transistor Q3 and the other end of the resistor R6. When the charging voltage value of the parasitic capacitance between the gate G of Q3 and its source S exceeds the turn-on threshold voltage Vth value of Q3, the auxiliary switch transistor Q3 is converted from the off state to the on state. When the state conversion is stable, since its input voltage V-Q3 is maintained at a high level, the voltage difference U(GS-Q3) between the positive driving signal VG-Q3 and the negative driving signal VS-Q3 at its output end will also be maintained at a high level, and the auxiliary switch transistor Q3 will be maintained in the on state.

[0201] like Fig. 9 , Fig.12 As shown, when the low current driving level signal V-Q3 is converted from a high level to a low level, the current flowing into the base B of the NPN switch transistor Q205 drops to 0A, and at this time, the NPN switch transistor Q205 is converted from an on state to an off state. Therefore, the current flowing through the base B of the PNP switch transistor Q206 drops to about 0A, and the PNP switch transistor Q206 is also converted from an on state to an off state. Therefore, the current flowing into the base B of the NPN switch transistor Q203 drops to about 0A, and the NPN switch transistor Q203 is also converted from an on state to an off state.

[0202] As mentioned above, since the NPN switch transistor Q203 is in the off state, the high-level driving voltage provided by it to the gate G and source S of the auxiliary switch transistor Q3 through the resistor R217 and the resistor R5 disappears. At this time, the charge stored in the parasitic capacitance between the gate G and the source S of the auxiliary switch transistor Q3 flows out from its gate G, and then is divided into two paths, one of which flows back to its source S (i.e., circuit node VS-Q3) through the resistor R6, and the other flows into the emitter E of the PNP switch transistor Q204 through the resistor R5 and the resistor R217 in turn, flows out from its base B, and then flows back to the source S of the auxiliary switch transistor Q3 (i.e., circuit node VS-Q3) through the resistor R216. At this time, due to the current amplification function of the PNP switch transistor Q204, a current larger than that of its base B will flow out from its collector C and flow back to the source S of the auxiliary switch transistor Q3 (i.e., circuit node VS-Q3). When the discharge voltage value of the parasitic capacitance between the gate G and the source S of the auxiliary switch transistor Q3 is lower than the turn-on threshold voltage Vth of the auxiliary switch transistor Q3, the auxiliary switch transistor Q3 is switched from the on state to the off state. When the state transition is stable, since its input voltage V-Q3 is maintained at a low level, the voltage difference U(GS-Q3) between the positive drive signal VG-Q3 and the negative drive signal VS-Q3 at its output end will also be maintained at a low level, and the auxiliary switch transistor Q3 will be maintained in the off state.

[0203] The working principle of the second delay trigger module 203 is described as follows: the circuit structure and internal working principle of the second delay trigger module 203 are exactly the same as those of the first driving module 204. The input reference voltage signal V-REF and the current state signal VP of the second delay trigger module 203 are the same as the corresponding input signals of the first driving module 204. The positive pole VCC1 and the negative pole VCC1-GND of the working power supply of the second delay trigger module 203 are exactly the same as those of the working power supply of the first driving module 204.

[0204] The input voltage signal VG-Q2 of the second delay trigger module 203 is the positive electrode of the driving voltage signal of the main switch transistor Q2, which is different from the circuit node of the corresponding input voltage signal VS-Q3 of the first driving module 204. The signal output terminal connection point VS-Q4 of the second delay trigger module 203 is different from the signal output terminal connection point V-Q3 of the first driving module 204.

[0205] As described above, the working principle analysis process of the second delay trigger module 203 is similar to the working principle analysis process of the first driving module 204, and will not be repeated here.

[0206] The working principle of the second driving module 206 is described as follows: Fig. 9 , Fig.12As shown, when the small current driving level signal V-Q4 is converted from a low level to a high level, the current flows out from the output end of the trigger IC206 in the second delay trigger module 205, and the current is divided into two paths after passing through the resistor R221, one of which flows into the base B of the NPN switching transistor Q201, and then merges into the resistor R223 through its emitter E, and the other flows into the negative electrode VCC1-GND of the first auxiliary power supply through the resistor R222. At this time, the NPN switch transistor Q201 is converted from the off state to the on state. Due to the current amplification function of the NPN switch transistor Q201, a current larger than the current flowing into its base B flows out from the positive electrode VCC1 of the first auxiliary power supply, and flows into the resistor R223 through the collector C of the NPN switch transistor Q201 and the emitter E of the NPN switch transistor Q201, flows to the output end through the resistor R223, and then flows into the gate G of the auxiliary switch transistor Q4 and one end of the resistor R8 through the resistor R7, and flows into the common connection point of the output end and the negative electrode VCC1-GND of the first auxiliary power supply through the source S of the auxiliary switch transistor Q4 and the other end of the resistor R8. When the charging voltage value of the parasitic capacitance between the gate G of the auxiliary switch transistor Q4 and its source S exceeds the turn-on threshold voltage Vth value of the auxiliary switch transistor Q4, the auxiliary switch transistor Q4 is converted from the off state to the on state. When the state transition is stable, since its small current drive level signal V-Q4 is maintained at a high level, the voltage difference U(GS-Q4) between its output positive drive signal VG-Q4 and negative drive signal VS-Q4 will be maintained at a high level, and the auxiliary switching transistor Q4 will be maintained in the on state.

[0207] like Fig. 9 , Fig.12 As shown, when the low current driving level signal V-Q4 is converted from high level to low level, the current flowing into the base B of the NPN switch transistor Q201 drops to 0 A. At this time, the NPN switch transistor Q201 is converted from the on state to the off state.

[0208] As mentioned above, since the NPN switch transistor Q201 is in the off state, the high-level driving voltage provided by it to the gate G and source S of the auxiliary switch transistor Q4 through the resistor R223 and the resistor R7 disappears. At this time, the charge stored in the parasitic capacitance between the gate G and the source S of the auxiliary switch transistor Q4 flows out from its gate G and is divided into two paths, one of which flows back to its source S (i.e., circuit node VS-Q4) through the resistor R8, and the other flows into the emitter E of the PNP switch transistor Q202 through the resistors R7 and R223 in turn, flows out from its base B, and then flows back to the source S of the auxiliary switch transistor Q4 (i.e., circuit node B, the received signal is the driving signal VS-Q4) through the resistor R222. At this time, due to the current amplification function of the PNP switch transistor Q202, a current larger than that of its base B will flow out from its collector C and flow back to the source S of the auxiliary switch transistor Q4 (i.e., circuit node B). When the discharge voltage value of the parasitic capacitance between the gate G and the source S of the auxiliary switch transistor Q4 is lower than the turn-on threshold voltage Vth of the auxiliary switch transistor Q4, the auxiliary switch transistor Q4 is switched from the on state to the off state. When the state transition is stable, since its input voltage V-Q4 is maintained at a low level, the voltage difference U(GS-Q4) between the positive drive signal VG-Q4 and the negative drive signal VS-Q4 at its output end will be maintained at a low level, and the auxiliary switch transistor Q4 will be maintained in the off state.

[0209] like Fig.10 , Fig.11 , Fig.12 As shown, the innovative resonant LLC switching power converter of the present invention is divided into two working states, namely, a small current load state 1 and a large current load state, by detecting the magnitude of the resonant cavity current. In the small current load state 1, the current detection module 201 outputs a signal VP of a low level, and in the large current load state 2, the current detection module 201 outputs a signal VP of a high level, wherein each working state has four working modes.

[0210] The working process and principle of the innovative resonant LLC switching power converter under different load states and different working modes are analyzed as follows:

[0211] ⑴ Low current load state 1:

[0212] like Fig.10 As shown, when the innovative resonant LLC switching power converter of the present invention is converted from the high current load state 2 to the low current load state 1, the output signal VP of the current detection module 201 is converted from a high level to a low level.

[0213] At this time, according to the above analysis of the first driving module 204, the voltage difference U(GS-Q3) between the driving signal VG-Q3 outputted from the positive output terminal of the first driving module 204 and the driving signal VS-Q3 outputted from the negative output terminal will be converted from a high level to a low level, and the auxiliary switch transistor Q3 will be converted from an on state to an off state. When the state conversion is stable, the voltage difference U(GS-Q3) will be maintained at a low level, and the auxiliary switch transistor Q3 will be maintained in an off state.

[0214] At this time, according to the above analysis of the second driving module 206, the voltage difference U(GS-Q4) between the driving signal VG-Q4 outputted from the positive output terminal of the second driving module 206 and the driving signal VS-Q4 outputted from the negative output terminal will be converted from a high level to a low level, and the auxiliary switch transistor Q4 will be converted from an on state to an off state. When the state conversion is stable, the voltage difference U(GS-Q4) will be maintained at a low level, and Q4 will be maintained in an off state.

[0215] As described above, in the low current load state 1, the auxiliary switch transistors Q3 and Q4 in the innovative resonant LLC switching power converter of the present invention are both maintained in the off state. At this time, its working mode is the same as that of the LLC switching power converter in the prior art, which will not be repeated here.

[0216] ⑵High current load state 2:

[0217] like Fig.10 As shown, when the power converter switches from the small current load state 1 to the large current load state 2, the output signal VP of the current detection module 201 switches from a low level to a high level.

[0218] At this time, according to the analysis of the first driving module 204, the voltage difference U(GS-Q3) between the third positive driving signal VG-Q3 and the third negative driving signal VS-Q3 of the first driving module 204 will be converted from a low level to a high level within a high-frequency switching cycle T period, and the auxiliary switch transistor Q3 will be converted from an off state to an on state. When the conversion state is stable, the voltage difference U(GS-Q3) will be maintained at a high level, and the auxiliary switch transistor Q3 will be maintained in an on state.

[0219] At this time, according to the analysis of the second driving module 206, the voltage difference U(GS-Q4) between the fourth positive driving signal VG-Q4 and the fourth negative driving signal VS-Q4 output by the second driving module 206 will be converted from a low level to a high level within a high-frequency switching cycle T period, and the auxiliary switch transistor Q4 will be converted from an off state to an on state. When the conversion state is stable, the voltage difference U(GS-Q4) will be maintained at a high level, and the auxiliary switch transistor Q4 will be maintained in an on state.

[0220] As mentioned above, in the high current load state 2, the auxiliary switch transistors Q3 and Q4 in the innovative resonant LLC switching power converter of the present invention are both maintained in the on state. At this time, the power converter of the present invention has the following 4 working modes, and its principle and process are analyzed as follows:

[0221] Working mode 1:

[0222] like Fig.10 As shown, in the mode 1 period in the high current load state 2, the first output drive voltage U(GS-Q1) of the control circuit module 10 is converted from a low level to a high level, the output second drive voltage U(GS-Q2) is maintained at a low level, the main switch transistor Q1 is converted from an off state to an on state, and the main switch transistor Q2 is turned off. The output drive voltages U(GS-Q3) and U(GS-Q4) of the second control module 112 are maintained at a high level, the auxiliary switch transistor Q3 is maintained on, and the auxiliary switch transistor Q4 is maintained on. As described in working mode 4, the turn-on process of the main switch transistor Q1 is zero voltage turn-on, so the turn-on power loss is close to 0W.

[0223] like Fig.10 As shown in FIG. 1 , since the on-resistance Rds(on) of the auxiliary switch transistors Q3 and Q4 is extremely low, one end of the auxiliary capacitor C4 is equivalent to being connected to the source S of the main switch transistor Q1, and the other end of the auxiliary capacitor C4 is directly connected to the drain D of the main switch transistor Q1. Therefore, the auxiliary capacitor C4 is equivalent to being connected in parallel between the drain D and the source S of the main switch transistor Q1.

[0224] Similarly, one end of the auxiliary capacitor C5 is equivalent to being connected to the source S of the main switch transistor Q2, and the other end of the auxiliary capacitor C5 is directly connected to the drain D of the main switch transistor Q2. Therefore, the auxiliary capacitor C5 is equivalent to being connected in parallel between the drain D and the source S of the main switch transistor Q1.

[0225] At this time, the current flows out from the positive electrode VI of the input DC voltage, through the drain D of the N-channel insulated gate field effect transistor Q1, the source S of the main switch transistor Q1, the two ends of the inductor L1, the two ends of the primary coil of the transformer T1, the two ends of the capacitor C2, and flows into the negative electrode VI-GND of the input DC voltage. At this time, the capacitor C2 is charged; the forward (here defined: the current flowing out of the inductor L1 and entering the transformer T1 is the forward current) energy storage of the inductor L1 increases. At this time, the voltage U across the auxiliary capacitor C4 is C4 ≈U Q1-DS-ON ≈0V, the voltage across the auxiliary capacitor C5 is U C5 ≈U Q2-DS-OFF ≈VI.

[0226] At this time, the output current of the positive end of the induced voltage of the secondary coil of transformer T1 is divided into two paths through the positive and negative electrodes of diode D1. One path of current flows back to the negative end of the voltage of the secondary coil of T1 through the positive and negative electrodes of capacitor C3, and the other path of current flows back to the negative end of the voltage of the secondary coil of T1 through the positive end of the output DC voltage VO, the positive and negative ends of the load, and the negative end of the output DC voltage VO-GND. At this time, capacitor C3 is charged; diode D2 is reversely turned off.

[0227] Working mode 2:

[0228] like Fig.10 As shown, in the mode 2 period in the high current load state 2, the first output drive voltage U(GS-Q1) of the control circuit module 10 is converted from the high level of the working mode 1 to the low level, and its second output drive voltage U(GS-Q2) is maintained at a low level, the main switch transistor Q1 is converted from the on state to the off state, and the main switch transistor Q2 is maintained at the off state. The output drive voltages U(GS-Q3) and U(GS-Q4) of the second control module 112 are maintained at a high level, the auxiliary switch transistor Q3 is maintained on, and the auxiliary switch transistor Q4 is maintained on.

[0229] At this time, after the current flows out from the positive electrode VI of the input DC voltage, it is divided into two paths. One path passes through the drain D of the main switch transistor Q1, the semiconductor channel resistance between the drain D and the source S of the main switch transistor Q1, and the source S of the main switch transistor Q1, and is merged into the circuit node VS-Q1. The other path passes through the positive electrode of the auxiliary capacitor C4, the negative electrode of C4, the drain D of the auxiliary switch transistor Q3, and the source S of the auxiliary switch transistor Q3, and is also merged into the circuit node VS-Q1, and then passes through both ends of the inductor L1, both ends of the primary coil of the transformer T1, and the positive electrode of the capacitor C2, and is merged into the common connection point of the negative electrode of the capacitor C2 and the source S of the main switch transistor Q2, and finally flows back to the negative electrode of the DC voltage VI-GND.

[0230] As described above, when the current flows through the common connection node of the drain D of the main switch transistor Q1 and the positive electrode of the auxiliary capacitor C4, it is divided into two branches. One branch flows through the semiconductor channel resistance between the drain D and the source S of the main switch transistor Q1 and merges into the circuit node VS-Q1. This resistance value changes rapidly from nearly zero ohms when the main switch transistor Q1 is turned on (the transition period is generally hundreds of nanoseconds) to several megaohms when the main switch transistor Q1 is turned off. The other branch current flows into the positive electrode of the auxiliary capacitor C4, and through the negative electrode of the auxiliary capacitor C4, the drain D of the auxiliary switch transistor Q3, and the source S of the auxiliary switch transistor Q3, it also merges into the circuit node VS-Q1. At this time, the auxiliary capacitor C4 is charged, and the voltage across it is charged from nearly 0V in mode 1 to U C4 ≈U Q1-DS-OFF ≈VI.

[0231] When the main switch transistor Q1 completes the shutdown process and the auxiliary capacitor C4 also completes the charging process, since the current in the inductor L1 cannot change suddenly, the freewheeling current flows out from the positive terminal of the self-inductance voltage of the inductor L1 (the positive direction current defined above), passes through the two ends of the primary coil of the transformer T1, the two ends of the capacitor C2, the source S of the main switch transistor Q2, the positive and negative poles of the parasitic reverse body diode between the source S and the drain D of the main switch transistor Q2, and the drain D of the main switch transistor Q2, and flows back to the negative terminal of the self-inductance voltage of the inductor L1.

[0232] At this time, the voltage value between the source S and the drain D of the main switch transistor Q2 is about -1V, and the main switch transistor Q2 meets the zero voltage turn-on condition.

[0233] By selecting appropriate specifications of the inductor L1 and the transformer T1, the energy stored in the inductor L1 is released quickly, and the current I L1 =I T1-1 =I C2 Rapidly decreases, and capacitor C2 continues to charge. L1 Less than or equal to the excitation current I of transformer T1 T1-M When the diode D1 is turned on, it quickly changes to the reverse off state. At this time, the diode D2 is still in the reverse off state, and the secondary coil of the transformer T1 also quickly stops charging the capacitor C3, and the capacitor C3 quickly changes from the charging state to the discharging state.

[0234] At this time, the current flows out from the positive electrode of the capacitor C3, passes through the output DC voltage positive terminal VO, the positive and negative terminals of the load, and the output DC voltage negative terminal VO-GND, and flows back to the negative electrode of the capacitor C3.

[0235] As described above, the current flowing out from VI is divided into two branches after passing through the common node of the drain D of the main switch transistor Q1 and the positive electrode of the capacitor C4. By selecting a suitable capacity of the capacitor C4 and increasing the turn-off speed of the main switch transistor Q1, during the turn-off process time period of the main switch transistor Q1 in working mode 2, the branch current value of one branch passing through the semiconductor channel between the drain D and the source S of the main switch transistor Q1 is extremely small and the time is short, while the branch current value of the other branch passing through the auxiliary capacitor C4 is large and the time is long. In this way, the main switch transistor Q1 can be softly turned off with a current close to zero, thereby greatly reducing the turn-off loss of the main switch transistor Q1 and further improving the efficiency of the switching power converter to meet the growing requirements of today's social economy for energy saving and carbon reduction.

[0236] Working mode 3:

[0237] like Fig.10As shown, in the mode 3 period in the high current load state 2, the first output drive voltage U (GS-Q1) of the control circuit module 10 is maintained at a low level, and the output second drive voltage U (GS-Q2) is converted from the low level in the working mode 2 period to a high level, the main switch transistor Q1 is maintained off, and the main switch transistor Q2 is converted from the off state to the on state. The output drive voltages U (GS-Q3) and U (GS-Q4) of the second control module 112 are maintained at a high level, the auxiliary switch transistor Q3 is maintained on, and the auxiliary switch transistor Q4 is maintained on. As described in working mode 2, the turn-on process of the main switch transistor Q2 is zero voltage turn-on, so the turn-on power loss is close to 0W.

[0238] By properly selecting the inductance value of the inductor L1 and the excitation inductance value of the primary coil of the transformer T1, the forward energy storage of the inductor L1 can be quickly released in the short period at the beginning of working mode 2 and mode 3. At this time, the current flows out from the positive electrode of the capacitor C2, through the two ends of the primary coil of the transformer T1, the two ends of the inductor L1, the drain D of the main switch transistor Q2, and the source S of the main switch transistor Q1, and flows back to the negative electrode of the capacitor C2. At this time, the capacitor C2 is discharged; the reverse (here defined as: the current flowing out of the transformer T1 and flowing into the inductor L1 is the reverse current) energy storage of the inductor L1 increases.

[0239] At this time, the output current of the positive end of the induced voltage of the secondary coil of transformer T1 is divided into two paths through the positive and negative electrodes of diode D2. One path of current flows back to the negative end of the voltage of the secondary coil of T1 through the positive and negative electrodes of capacitor C3, and the other path of current flows back to the negative end of the voltage of the secondary coil of T1 through the positive end of the output DC voltage VO, the positive and negative ends of the load, and the negative end of the output DC voltage VO-GND. At this time, capacitor C3 is charged; diode D1 is reversely turned off.

[0240] Working mode 4:

[0241] like Fig.10 As shown, in the mode 4 period in the high current load state 2, the first output drive voltage U(GS-Q1) of the control circuit module 10 is maintained at a low level, the output second drive voltage U(GS-Q2) is converted from the high level in the working mode 3 period to the low level, the main switch transistor Q1 is maintained off, and the main switch transistor Q2 is converted from the on state to the off state. The output drive voltages U(GS-Q3) and U(GS-Q4) of the second control module 112 are maintained at a high level, the auxiliary switch transistor Q3 is maintained on, and the auxiliary switch transistor Q4 is maintained on.

[0242] At this time, the current flows out from the positive terminal of the capacitor C2, passes through the two ends of the primary coil of the transformer T1, the two ends of the inductor L1, and the common connection node of the drain D of the main switch transistor Q2 and the positive electrode of the auxiliary capacitor C5, and then is divided into two branches. One branch passes through the drain D of the main switch transistor Q2, the semiconductor channel resistance between the drain D and the source S of the main switch transistor Q2, and the source S of the main switch transistor Q2, and is merged into the negative DC voltage electrode VI-GND. The other branch passes through the positive electrode of the auxiliary capacitor C5, the negative electrode of the auxiliary capacitor C5, the drain D of the auxiliary switch transistor Q3, and the source S of the auxiliary switch transistor Q3, and is also merged into the negative DC voltage electrode VI-GND.

[0243] As mentioned above, when the current flows through the common connection node of the drain D of the main switch transistor Q2 and the positive electrode of the auxiliary capacitor C5, it is divided into two branches. One branch flows through the semiconductor channel resistance between the drain D and the source S of the main switch transistor Q2, and merges into the negative electrode of the DC voltage VI-GND. This resistance value changes rapidly from nearly zero ohms when the main switch transistor Q2 is turned on (the transition period is generally hundreds of nanoseconds) to several megaohms when the main switch transistor Q2 is turned off. The other branch current flows into the positive electrode of the auxiliary capacitor C5, through the negative electrode of the auxiliary capacitor C5, the drain D of the auxiliary switch transistor Q4, and the source S of the auxiliary switch transistor Q4, and merges into the negative electrode of the input DC voltage VI-GND. At this time, the auxiliary capacitor C5 is charged, and the voltage across it is charged from nearly 0V in mode 3 to the maximum value U C5 ≈U Q2-DS-OFF ≈VI. At the same time, the auxiliary capacitor C4 is discharged, and the voltage across it changes from U C4 ≈U Q1-DS-OFF ≈VI, the minimum value is discharged to close to 0V.

[0244] As mentioned above, during the charging process of the auxiliary capacitor C5, when the auxiliary capacitor C5 voltage U C5 Greater than the voltage U across capacitor C2 C2 When the voltage across the inductor L1 is reversed, the stored energy of the inductor L1 will be gradually released, and the self-induced electromotive force of the inductor L1 will continue to maintain the direction of the above current (the reverse current defined above), but its current value will continue to decrease. The current flows out from the positive end of the self-induced voltage of the inductor L1, through the common connection point of the source S of the main switching transistor Q1 and the negative electrode of the auxiliary capacitor C4, the positive and negative electrodes of the parasitic reverse body diode between the source S and the drain D of the main switching transistor Q1, the common connection point of the drain D of the main switching transistor Q1 and the positive electrode of the auxiliary capacitor C4, the positive electrode of the capacitor C1, the negative electrode of C1, the negative electrode of the capacitor C2, the positive electrode of C2, and the two ends of the primary coil of the transformer T1, and flows into the negative end of the self-induced voltage of the inductor L1.

[0245] At this time, the voltage value between the source S and the drain D of the main switch transistor Q1 is about -1V, and the main switch transistor Q1 meets the zero voltage turn-on condition. The main switch transistor Q1 will achieve zero voltage soft turn-on in the working mode 1 of the next cycle.

[0246] At this time, the energy stored in the inductor L1 is released quickly, the capacitor C2 continues to discharge, and the diode D2 quickly switches from the on state to the reverse off state. At this time, the diode D1 is still in the reverse off state, and the secondary coil of the transformer T1 also quickly stops charging the capacitor C3, and the capacitor C3 quickly switches from the charging state to the discharging state.

[0247] At this time, the current flows out from the positive electrode of the capacitor C3, passes through the output DC voltage positive terminal VO, the positive and negative terminals of the load, and the output DC voltage negative terminal VO-GND, and flows back to the negative electrode of the capacitor C3.

[0248] As described above, in the innovative resonant LLC switching power converter of the present invention, as can be seen from the above working mode analysis, in the mode 2 and mode 4 in the large current load state 2 of the present invention, due to the current bypass effect of the auxiliary capacitors C4 and C5, the main switch transistors Q1 and Q2 achieve extremely low current soft shutdown during the shutdown process. Therefore, the present invention further reduces the power loss of the LLC switching power converter and further improves the conversion efficiency by an innovative method.

[0249] In summary, the innovative resonant LLC switching power converter of the present invention meets the economic and social development direction of energy saving, consumption reduction and low carbon, and has broad application prospects.

[0250] An implementation circuit of the innovative resonant LLC switching power converter of the present invention is exemplified as follows: Fig. 9 As shown, the parameters of each component of the circuit in the example are selected as follows:

[0251] Resistance: R1 = R3 = R5 = R7 = 10Ω, R2 = R4 = R6 = R8 = 10KΩ;

[0252] Main switch transistor Q1, Q2 model: N-channel field effect transistor SPP20N60C3 (20A, 600V);

[0253] Auxiliary switch transistor Q1, Q2 model: N-channel field effect transistor SPP02N60C3 (2A, 600V);

[0254] Inductor L1 = 235uH; Capacitor C2 = 15nH; Transformer T1 primary coil excitation inductor LP1 = 1mH;

[0255] DC input voltage: U(VI)=380V;

[0256] Steady-state DC output voltage: U(VO)=24V; output power: 230W;

[0257] Capacitor: C1 = 220uF, C3 = 4400uF;

[0258] Auxiliary power supply voltage: U(VCC1, VCC1-GNG)=12V;

[0259] The switching frequency of the switching converter at full output load is 83.3kHz.

[0260] The dead time of the first driving signal waveform U(GS-Q1) and the second driving signal waveform U(GS-Q2) output by the first control module 111 is 900 nS.

[0261] When the switching power converter using the innovative technology of the present invention is operated at full load (belonging to the above-mentioned high current load state 2), the measured Fig. 9 In the circuit shown, the driving voltage waveform of the main switch transistors Q1 and Q2 and the current waveform of the main components are compared on the same time axis, as shown in FIG. Fig.13 As shown above, it can be measured as follows Fig. 9 The driving voltage waveform of the main switching transistors Q1 and Q2 in the circuit shown is compared with the voltage waveform of the main components on the same time axis, as shown in FIG. Fig.14 As shown above. Fig.13 , Fig.14 The detailed expansion diagram of the voltage and current waveforms of the four working modes identified in the same time axis comparison diagram is as follows: Fig.15 As shown:

[0262] Figure 13-15 Meaning of symbols: U(GS-Q*) is the voltage waveform of the gate G and source S of the NMOSFET transistor Q*, and the * represents any number or letter number; U(DS-Q*) is the voltage waveform of the drain D and source S of the NMOSFET transistor Q*; I(DS-Q*) is the current waveform flowing through the drain D and source S of the NMOSFET transistor Q*; I(C*) is the current waveform of the capacitor, and the positive direction of measurement is the current flowing out of the positive electrode of the capacitor. U(L1) is the voltage waveform across the inductor L1, and the end of the inductor L1 connected to the drain D of the transistor Q2 is the positive voltage terminal for measurement; I(L1) is the current waveform flowing through the inductor L1, and the direction of measurement of the current flowing into the positive voltage terminal of L1 is the positive direction of measurement; U(VO) is the voltage waveform of the output terminal VO of the switching power converter, and the positive direction of measurement is the current flowing into the positive voltage terminal of L1. Fig. 9 The circuit node VO-GND in is the reference 0 level. U(VI) is the voltage waveform at the input terminal VI of the switching power converter. Fig. 9 The circuit node VI-GND in the circuit is referenced to the 0 level. Fig. 9 , Fig.10It can be seen that when the innovative resonant LLC switching power converter of the present invention works in the high current load state 2, because the driving voltages U(GS-Q3) and U(GS-Q4) of the auxiliary switch transistors Q3 and Q4 are both maintained at a high level, the auxiliary switch transistors Q3 and Q4 are turned on and the on-resistance is extremely low. At this time, the auxiliary capacitor C3 connected in series with the auxiliary switch transistor Q3 is equivalent to being connected in parallel across the drain D and source S of the main switch transistor Q1, and the auxiliary capacitor C4 connected in series with the auxiliary switch transistor Q4 is equivalent to being connected in parallel across the drain D and source S of the main switch transistor Q2.

[0263] like Figure 13-15 As shown, in the process of the main switch transistor Q1 switching from the on state to the off state during the working mode 2 period, the current I(DS-Q1) flowing through the drain D and source S of the main switch transistor Q1 immediately drops from about 500mA to less than 100mA. Since the main switch transistor Q1 is completely turned off at this time, the current of about 100mA is the current flowing through the parasitic capacitance between the drain D and source S of the main switch transistor Q1, and its loss is extremely low and can be ignored. At this time, the current I(C3) flowing through the auxiliary capacitor C3 immediately rises from about 50mA to a peak of about 400mA.

[0264] like Figure 13-15 As shown in FIG. 1 , when the driving voltage U(GS-Q1) of the main switch transistor Q1 has dropped to nearly 0V and the main switch transistor Q1 has been completely turned off, the drain D and source S voltage U(DS-Q1) of the main switch transistor Q1 rises from nearly 0V to about 380V at a relatively slow slope. Therefore, in working mode 2, there is no crossover period in which the values ​​of the two parameters of the drain D and source S current waveform I(DS-Q2) and the voltage waveform U(GS-Q2) of Q2 are both significantly greater than 0 at the same time.

[0265] In the half-bridge LLC switching power converter, the turn-off loss calculation formula of the NMOSFET main switch transistor Q1 is as follows:

[0267]

[0268] Therefore, it can be known that the innovative resonant LLC switching power converter of the present invention realizes the delay-free conversion of the path of the power converter resonant cavity current I(L1) during the rapid shutdown process of the main switching transistor Q1, and I(L1) switches without delay from the semiconductor conduction channel between the drain D and the source S of the main switching transistor Q1 (operating mode 1) to the auxiliary capacitor C3 branch and the parasitic capacitor branch of the main switching transistor Q1 (operating mode 2), thereby causing the main switching transistor Q1 current I(DS-Q1) to drop significantly and immediately without delay during its shutdown process.

[0269] T in Formula 5ON-OFF-Q1 The parameter is the voltage value U between the drain D and the source S of the main switch transistor Q1 during the shutdown process. DS-Q1 With the current value I DS-Q1 At the same time, the crossover time period is greater than zero, such as Figure 13-15 As shown and mentioned above, it can be seen that T ON-OFF-Q1 ≈0nS. Therefore, according to Formula 5, E ON-OFF-Q1 ≈0J, so the innovative resonant LLC switching power converter of the present invention makes the turn-off loss of the main switch transistor Q1 very small, and realizes the zero-current soft turn-off function of the main switch transistor Q1.

[0270] like Figure 13-15 As shown, in the process of the main switch transistor Q2 switching from the on state to the off state during the working mode 4 period, the current I(DS-Q2) flowing through the drain D and source S of the main switch transistor Q2 immediately drops from about 500mA to less than 100mA. Since the main switch transistor Q2 is completely turned off at this time, the current of about 100mA is the current flowing through the parasitic capacitance between the drain D and source S of the main switch transistor Q2, and its loss is extremely low and can be ignored. At this time, the current I(C3) flowing through the auxiliary capacitor C3 immediately rises from about 50mA to a peak of about 400mA.

[0271] like Figure 13-15 As shown in FIG. 4 , when the driving voltage U(GS-Q2) of the main switch transistor Q2 has dropped to nearly 0V and the main switch transistor Q2 has been completely turned off, the drain D and source S voltage U(DS-Q2) of the main switch transistor Q2 rises from nearly 0V to about 380V at a relatively slow slope. Therefore, in working mode 4, the drain D and source S current waveform I(DS-Q2) and the voltage waveform U(GS-Q2) of the main switch transistor Q2 do not have a crossover period in which the values ​​of the two parameters are significantly greater than 0 at the same time.

[0272] The hard turn-off loss calculation formula of the NMOSFET auxiliary switch transistor Q4 in the half-bridge LLC switching power converter is as follows:

[0273]

[0274] Meaning of symbols in formula 5-6: U VS-Q1 It is the instantaneous value of the voltage between the circuit node VS-Q1 and the negative pole of the input DC voltage VI-GND at the 0-point reference point; U R* is the instantaneous value of the voltage across the resistor R*; T ON-OFF-Q* The voltage U between the drain D and source S of the switch tube Q* during the transition period when the switch tube Q* changes from the fully on state to the fully off state. DS-Q* With current I DS-Q*The time period when the value is significantly greater than 0; W is the operating switching frequency of the main switching transistors Q1 and Q2 in the resonant LLC switching power converter; E ON-OFF-Q* U is the power loss value generated during the transition period when the switch tube Q* changes from a fully on state to a fully off state; DS-Q* is the instantaneous voltage value between the drain D and source S of the NMOSFET switch tube Q*; I DS-Q* is the instantaneous value of the current flowing through the semiconductor channel between the drain D and the source S of the NMOSFET switch tube Q*; I DS-MAX-Q* It is the peak current flowing through the semiconductor channel between the drain D and the source S of the NMOSFET switch tube Q*; * represents a number or letter label.

[0275] Therefore, it can be known that the innovative resonant LLC switching power converter of the present invention realizes the delay-free conversion of the path of the power converter resonant cavity current I(L1) during the rapid shutdown process of the main switching transistor Q2, and I(L1) switches without delay from the semiconductor conduction channel between the drain D and the source S of the main switching transistor Q2 (working mode 3) to the parasitic capacitance branch (working mode 4) flowing into the auxiliary capacitor C4 branch and the main switching transistor Q2, so that the main switching transistor Q2 current I(DS-Q2) drops significantly without delay during its shutdown process.

[0276] T in Formula 6 ON-OFF-Q2 The parameter is the voltage value U between the drain D and the source S of the main switch transistor Q2 during the shutdown process. DS-Q2 With the current value I DS-Q2 At the same time, the crossover time period is greater than zero, such as Figure 13-15 As shown and mentioned above, it can be seen that T ON-OFF-Q2 ≈0nS. Therefore, according to Formula 6, E ON-OFF-Q2 ≈0J, so the innovative resonant LLC switching power converter of the present invention makes the turn-off loss of the main switch transistor Q2 very small, and realizes the soft turn-off function of the main switch transistor Q2.

[0277] like Fig.13 As shown, when the voltage U(DS-Q1) between the drain D and the source S of the main switch transistor Q1 drops to nearly 0V, its driving voltage U(GS-Q1) rises from nearly 0V to about 12V. Therefore, the innovative resonant LLC switching power converter of the present invention makes the turn-on loss of the main switch transistor Q1 very small, and realizes the zero-voltage soft turn-on function of the main switch transistor Q1.

[0278] like Fig.13As shown, when the voltage U(DS-Q2) between the drain D and the source S of the main switch transistor Q2 drops to nearly 0V, its second drive voltage U(GS-Q2) rises from nearly 0V to about 12V. Therefore, the innovative resonant LLC switching power converter of the present invention makes the turn-on loss of the main switch transistor Q2 very small, and realizes the zero-voltage soft turn-on function of the main switch transistor Q2.

[0279] In summary, the innovative resonant LLC switching power converter of the present invention realizes the zero voltage soft turn-on and zero current soft turn-off functions of the main switch transistors Q1 and Q2, making the switching loss very small, further improving the efficiency of the power converter. Therefore, the present invention meets the growing demand for high efficiency, energy saving and carbon reduction in today's social economy, has the prospect of large-scale application, and has broad social and economic benefits.

[0280] like Figure 1-15 As shown, the half-bridge arm switching circuit in the innovative resonant LLC switching power converter of the present invention is innovatively composed of two main switching transistors Q1, Q2 and two auxiliary switching transistors Q3, Q4 and two auxiliary capacitors C4, C5, and their innovative connection method.

[0281] During the working mode 2 period of the innovative resonant LLC switching power converter in the high current load state 2, during the rapid shutdown process of the main switch transistor Q1, the delay-free conversion of the path of the power converter resonant cavity current I(L1) is achieved, and I(L1) switches without delay from the semiconductor conduction channel between the drain D and the source S of Q1 (working mode 1) to the parasitic capacitance branch flowing into the auxiliary capacitor C3 branch and the main switch transistor Q1 (working mode 2), thereby causing the main switch transistor Q1 current I(DS-Q1) to drop significantly without delay during its shutdown process.

[0282] T in Formula 5 ON-OFF-Q1 The parameter is the voltage value U between the drain D and the source S of the main switch transistor Q1 during the shutdown process. DS-Q1 With the current value I DS-Q1 At the same time, the crossing time period is greater than zero. As mentioned above, it can be seen that in the innovative resonant LLC switching power converter of the present invention, T ON-OFF-Q1 ≈0nS, so according to Formula 5, E ON-OFF-Q1 ≈0J, so the innovative resonant LLC switching power converter of the present invention makes the turn-off loss of the main switch transistor Q1 very small, and realizes the zero-current soft turn-off function of the main switch transistor Q1.

[0283] During the working mode 4 period of the innovative resonant LLC switching power converter in the high current load state 2, during the rapid shutdown process of the main switch transistor Q2, the path of the power converter resonant cavity current I(L1) is converted without delay, and I(L1) switches without delay from the semiconductor conduction channel between the drain D and the source S of the main switch transistor Q2 (working mode 3) to the parasitic capacitance branch of the auxiliary capacitor C4 branch and the main switch transistor Q2 (working mode 4), so that the main switch transistor Q2 current I(DS-Q2) drops significantly and immediately without delay during its shutdown process.

[0284] T in Formula 6 ON-OFF-Q2 The parameter is the voltage value U between the drain D and the source S of the main switch transistor Q2 during the shutdown process. DS-Q2 With the current value I DS-Q2 At the same time, the crossover time period is greater than zero. As mentioned above, it can be seen that in the innovative resonant LLC switching power converter of the present invention, T ON-OFF-Q2 ≈0nS, so according to Formula 6, E ON-OFF-Q2 ≈0J. Therefore, the innovative resonant LLC switching power converter of the present invention makes the turn-off loss of the main switch transistor Q2 very small, and realizes the zero-current soft turn-off function of the main switch transistor Q2.

[0285] like Fig.14 As shown, when the voltage U(DS-Q1) between the drain D and source S of the main switch transistor Q1 drops to nearly 0V, its driving voltage U(GS-Q1) rises from nearly 0V to about 12V. Therefore, the innovative resonant LLC switching power converter of the present invention makes the turn-on loss of Q1 very small, and realizes the zero-voltage soft turn-on function of the main switch transistor Q1.

[0286] like Fig.14 As shown, when the voltage U(DS-Q2) between the drain D and the source S of the main switch transistor Q2 drops to nearly 0V, its second drive voltage U(GS-Q2) rises from nearly 0V to about 12V. Therefore, the innovative resonant LLC switching power converter of the present invention makes the turn-on loss of the main switch transistor Q2 very small, and realizes the zero-voltage soft turn-on function of the main switch transistor Q2.

[0287] In summary, Figure 1 Compared with the high-current hard-off state of the switch module in the prior art shown in the figure, the innovative resonant LLC switching power converter of the present invention realizes the zero-voltage soft-on and zero-current soft-off functions of the main switch transistors Q1 and Q2 in the high-current load state 2, so that its switching loss becomes very small. Compared with the prior art, the technology of the present invention further improves the efficiency of the power converter.

[0288] like Fig.10As shown, in the innovative resonant LLC switching power converter of the present invention, in the small current load state 1, the output drive signals U(GS-Q3) and U(GS-Q4) of the second control module 112 are both at a low level, and the auxiliary switch transistors Q3 and Q4 are both maintained in the off state. At this time, the auxiliary capacitors C4 and C5 are disconnected from the half-bridge arm main switch transistors Q1 and Q2.

[0289] A prior art has the following problem: because a prior art directly connects a capacitor in parallel between the drain D and the source S of the bridge arm switch tube, when its driving pulse is a discontinuous pulse group output, the switch tube is in a hard-on state at the first pulse after the driving pulse is interrupted. At this time, the above-mentioned auxiliary capacitor added to the switch tube will increase the switching loss. Because under normal circumstances, when the resonant LLC switching power converter is in no-load standby and extremely light load, the control module adjusts the operating frequency to increase to the set maximum safe frequency value in order to stabilize the output voltage, but it still cannot achieve the voltage stabilization requirement. At this time, in order to stabilize the output voltage, the control module will enter a state of continuously interrupted driving pulse group output. Therefore, the above-mentioned prior art method will increase the loss of the LLC switching power converter in standby and light load, and reduce efficiency.

[0290] As described above, the innovative resonant LLC switching power converter of the present invention does not have the problem of increased switch tube turn-on loss in the above-mentioned prior art. Therefore, the present invention can not only realize the zero current shutdown and zero voltage turn-on functions of the above-mentioned main switch transistor Q1 or Q2 in the large current load state 2, but also can meet the requirements of not increasing the turn-on power consumption of the main switch transistor Q1 or Q2 in the small current load state 1, and realize low standby power consumption and high efficiency under small load.

[0291] In summary, the innovative resonant LLC switching power converter of the present invention can further reduce losses and improve efficiency within the full load range and all operating states. Therefore, it can fully meet the social and economic development trend that places higher demands on energy saving, carbon reduction, low standby power consumption, and high efficiency under small loads, and is conducive to the large-scale popularization and application of this technology in more emerging application scenarios and under higher requirements.

[0292] like Fig.10 As shown, the input-output control logic definition of the innovative second control module 112 in the innovative resonant LLC switching power converter of the present invention is as follows: the output drive signal U (GS-Q3) needs two necessary conditions to be converted from a low level to a high level or from a high level to a low level.

[0293] like Fig.10As shown, when the power converter enters the high current load state 2 from the low current load state 1, the auxiliary switch transistors Q3 and Q4 cannot be turned on immediately. At this time, the second control module 112 also needs to judge the state of the main switch transistors Q1 and Q2, and wait for the main switch transistors Q1 and Q2 to enter a stable on state before immediately triggering the auxiliary switch transistors Q3 and Q4 to turn on. After the auxiliary switch transistors Q3 and Q4 are triggered to turn on, they will be locked in the on state. At this time, only when the following conditions for triggering the auxiliary switch transistors Q3 and Q4 to turn off are met can they be switched to the off state. Therefore, the auxiliary switch transistors Q3 and Q4 both realize a high-efficiency zero-voltage soft turn-on function, and the turning on of the auxiliary switch transistors Q3 and Q4 does not interfere with the main circuit.

[0294] like Fig.10 As shown, when the LLC power converter enters the low current load state 1 from the high current load state 2, the auxiliary switch transistors Q3 and Q4 cannot be turned off immediately. At this time, the second control module 112 also needs to judge the state of the main switch transistors Q1 and Q2, and wait for the main switch transistors Q1 and Q2 to enter a stable on state before immediately triggering the auxiliary switch transistors Q3 and Q4 to turn off. After the auxiliary switch transistors Q3 and Q4 are triggered to turn off, they will be locked in the off state. At this time, only when the above-mentioned conditions for triggering the auxiliary switch transistors Q3 and Q4 to turn on are met can they be switched to the on state. Therefore, the auxiliary switch transistors Q3 and Q4 both realize a high-efficiency zero-voltage soft turn-off function, and the turning off of the auxiliary switch transistors Q3 and Q4 does not interfere with the main circuit.

[0295] and Figure 1 Compared with the prior art shown, the innovative resonant LLC switching power converter of the present invention not only realizes the high-efficiency functions of zero-voltage soft turn-on and zero-current soft turn-off of the main switching transistors Q1 and Q2 in the large current load state 2, but also realizes the high-efficiency functions of zero-voltage turn-on and zero-voltage turn-off of the auxiliary switching transistors Q3 and Q4 in all working states.

[0296] The innovative second control module 112 and its circuit composition and connection method in the innovative resonant LLC switching power converter of the present invention are all independent newly added circuits without changing the control circuit of the prior art. The first control module 111 used in the technology of the present invention is compatible with the control module of the prior art, so the total R&D and update cost of the power supply system can be reduced. Therefore, the low-efficiency resonant LLC switching power converter of the prior art can be upgraded to the resonant LLC switching power converter innovated by the present invention at a very low cost, and the present invention can achieve high efficiency and energy saving requirements in any working current state.

[0297] The half-bridge arm switch circuit of the present invention is innovatively composed of two main switch transistors Q1, Q2 and two auxiliary switch transistors Q3, Q4 and two auxiliary capacitors C4, C5, and an innovative connection method. The same two sets of the innovative half-bridge arm circuits of the present invention can be applied to the full-bridge power converter to meet the needs of different application scenarios.

[0298] In summary, the innovative resonant LLC switching power converter technology of the present invention simultaneously meets the growing demand of today's social economy for high efficiency under large loads and high efficiency under extremely small loads, as well as energy saving, carbon reduction, and low standby power consumption. In addition, the technology of the present invention has strong versatility, low updating cost, and high reliability, and meets the technical requirements of a large number of traditional and emerging application fields. Therefore, the technology of the present invention has the prospect of large-scale application and considerable social and economic benefits.

[0299] The above-mentioned inventive points of the innovative resonant LLC switching power converter of the present invention can also be applied to LLC switching power supplies with a full-bridge topology structure. The innovative resonant full-bridge LLC switching power converter of the present invention is composed of the half-bridge arm switching circuit innovated by the above-mentioned inventive points.

[0300] The above-mentioned inventive points of the innovative resonant LLC switching power supply converter of the present invention can also be applied to a full-bridge switching power supply with a phase shifting control method. The innovative phase-shifting full-bridge switching power supply converter of the present invention is composed of two groups of half-bridge arm switching circuits innovated by the above-mentioned inventive points.

[0301] The resonant LLC switching power converter, resonant full-bridge LLC switching power converter and phase-shifted full-bridge switching power converter innovated by the present invention can be applied to high-frequency switching combination power supplies of any appearance and structural form.

[0302] The resonant LLC switching power converter, resonant full-bridge LLC switching power converter and phase-shifted full-bridge switching power converter innovated by the present invention can be implemented in any printed circuit board layout connection mode.

[0303] The components in the resonant LLC switching power converter, resonant full-bridge LLC switching power converter and phase-shifted full-bridge switching power converter innovated by the present invention can be components of the same principle and performance in any package and their series and parallel combinations.

[0304] Embodiment 2

[0305] This second embodiment provides a control method for a resonant LLC switching power converter based on the first embodiment, such as Fig.16 As shown, the control method is applied to the control module, including:

[0306] Step S10: Output a first control signal to the first switch module to control the on / off state of the first switch module, and output a second control signal to the second switch module to control the on / off state of the second switch module.

[0307] Step S20. Obtain the output current of the resonant LLC switching power converter, compare the output current with the preset current, and control the on-off state of the third switch module according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module according to the comparison result and the state of the second control signal.

[0308] Further, in step S20, the output current of the resonant LLC switching power converter is obtained, including:

[0309] The output current of the resonant LLC switching power converter is obtained by detecting the effective value of the AC component of the voltage across the second capacitor module.

[0310] Further, such as Fig.17 As shown, in step S20, controlling the on / off state of the third switch module according to the current comparison result and the state of the first control signal includes:

[0311] Step S201. When the output current is greater than the preset current and the first control signal is switched from the first level signal to the second level signal, the third switch module is controlled to switch from the first switch state to the second switch state by delay, and the second switch state is maintained.

[0312] Step S202: When the output current is not greater than the preset current and the first control signal is switched from the first level signal to the second level signal, the third switch module is controlled to switch from the second switch state to the first switch state by delay, and the first switch state is maintained.

[0313] Among them, in step S201, when the output current is greater than the preset current, it is judged that the load is in a relatively large current state, and at this time, the third switch module needs to be controlled to adapt to the high current working requirements. When the first control signal switches from the first level signal to the second level signal, it indicates that the control needs to be switched. In this case, the third switch module is switched from the first switch state (e.g., the disconnected state) to the second switch state (e.g., the on state) through delay control. Once the third switch module enters the second switch state, it will maintain its on state to ensure that the current can flow stably and reach the predetermined working mode of the system to meet the requirements under high load current.

[0314] In step S202, when the output current is not greater than the preset current, it is determined that the load is in a relatively low current state. At this time, the third switch module does not need to remain in the on state to avoid unnecessary power loss.

[0315] When the first control signal switches from the first level signal to the second level signal, the control system still needs to perform delay adjustment. In this case, through delay control, the third switch module will switch from the second switch state (on state) back to the first switch state (off state), ensuring that the flow of current is cut off, thereby reducing power loss at low load. When the third switch module is maintained in the first switch state, it ensures that the system is in a low-power operation mode, which meets the requirements of efficient operation under light load conditions.

[0316] The technical effect of this embodiment is that through the control strategy of steps S201 and S202, the third switch module can dynamically adjust its on-off state according to the size of the load current, thereby achieving efficient power consumption management and system optimization. Under high current load, the switch is switched to the on state through delay control and remains on, ensuring stable current flow, meeting high load working requirements, while achieving zero voltage conduction and reducing switching losses. Under low current load, the switch is switched to the off state through delay control, reducing power consumption and ensuring efficient operation of the system under light load conditions. The overall design effectively improves the operating efficiency of the power converter, optimizes the power conversion performance, and significantly reduces switching losses.

[0317] Further, such as Fig.18 As shown, controlling the on-off state of the fourth switch module according to the current comparison result and the state of the second control signal includes:

[0318] Step S203: When the output current is greater than the preset current and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is controlled to switch from the first switch state to the second switch state with delay, and the second switch state is maintained.

[0319] Step S204: When the output current is not greater than the preset current and the second control signal switches from the first level signal to the second level signal, delay controlling the fourth switch module to switch from the second switch state to the first switch state and maintain the first switch state.

[0320] The control process of step S203 and step S204 is similar to the control process of step S201 and step S202, and will not be repeated here.

[0321] The above embodiments are only used to illustrate the technical solutions of the present invention, rather than to limit the same. Although the present invention has been described in detail with reference to the aforementioned embodiments, those skilled in the art should understand that the technical solutions described in the aforementioned embodiments may still be modified, or some of the technical features may be replaced by equivalents. Such modifications or replacements do not deviate the essence of the corresponding technical solutions from the spirit and scope of the technical solutions of the embodiments of the present invention, and should be included in the protection scope of the present invention.

Claims

1. A resonant LLC switching power converter, characterized in that: include: The resonant LLC switching power converter includes a first switch module, a second switch module, a third switch module, a fourth switch module, a first capacitor module, a second capacitor module, a third capacitor module, an energy storage and isolation module, and a control module; The first end of the first switch module is connected to the first end of the second capacitor module, the second end of the first switch module is respectively connected to the first end of the second switch module, the second end of the third switch module, the first end of the third capacitor module and the first end of the energy storage and isolation module, the second end of the second capacitor module is connected to the first end of the third switch module, the second end of the second switch module is respectively connected to the second end of the fourth switch module and the first end of the first capacitor module, the second end of the first capacitor module is connected to the second end of the energy storage and isolation module, the second end of the third capacitor module is connected to the first end of the fourth switch module, and the control module is respectively connected to the control end and the second end of the first switch module, the control end and the second end of the second switch module, the control end and the second end of the third switch module and the control end and the second end of the fourth switch module; The control module is used to output a first control signal to the first switch module to control the on / off state of the first switch module, and output a second control signal to the second switch module to control the on / off state of the second switch module; The control module is also used to obtain the output current of the resonant LLC switching power converter, compare the output current with a preset current, and control the on-off state of the third switch module according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module according to the comparison result and the state of the second control signal.

2. The resonant LLC switching power converter according to claim 1, characterized in that: The control module comprises: a first control module, connected to a control end of the first switch module and a control end of the second switch module, respectively, and configured to output a first control signal to the first switch module to control the on / off state of the first switch module, and output a second control signal to the second switch module to control the on / off state of the second switch module; a second control module, which is respectively connected to the control end of the first switch module, the control end of the second switch module, the control end of the third switch module and the control end of the fourth switch module, and is used to obtain the output current of the resonant LLC switching power converter, compare the output current with a preset current, and control the on-off state of the third switch module according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module according to the comparison result and the state of the second control signal.

3. The resonant LLC switching power converter according to claim 2, characterized in that: The second control module includes: A reference voltage module, used for outputting a reference voltage; a current detection module, wherein a first input end thereof is connected to the second end of the first capacitor module, and a second input end thereof is connected to the output end of the reference voltage module, and is used to collect the output current of the resonant LLC switching power converter, convert the output current into a collection voltage, and compare the collection voltage with the reference voltage to output a first comparison signal; a first control submodule, wherein a first input end thereof receives the first control signal, a second input end thereof is connected to an output end of the reference voltage module, a third input end thereof is connected to an output end of the current detection module, an output end thereof is connected to a control end of the third switch module, and a first on-off control signal is output to the third switch module according to the first control signal, the reference voltage and the first comparison signal; A second control submodule, whose first input end receives the second control signal, whose second input end is connected to the output end of the reference voltage module, whose third input end is connected to the output end of the current detection module, and whose output end is connected to the control end of the fourth switch module, outputs a second on-off control signal to the fourth switch module according to the second control signal, the reference voltage and the first comparison signal.

4. The resonant LLC switching power converter according to claim 3, characterized in that: The first control submodule includes: a first delay trigger module, whose first input terminal receives the first control signal, whose second input terminal receives the reference voltage, and whose third input terminal receives the first comparison signal, and is used to output a first weak current on-off control signal according to the first control signal, the reference voltage and the first comparison signal; The first driving module has an input end connected to the output end of the first delay trigger module and is used to amplify and isolate the first weak current on-off control signal and then output a first on-off control signal.

5. The resonant LLC switching power converter according to claim 4, characterized in that: The first delay trigger module includes: A first comparison module, whose first input terminal receives the first control signal and whose second input terminal receives the reference voltage, is used to delay the conversion of the first control signal into a first acquisition voltage, and output a first trigger signal when the first acquisition voltage is greater than the reference voltage; A first trigger module, whose first input end receives the first comparison signal and whose second input end is connected to the output end of the first comparison module, is used to output the first weak current on-off control signal according to the first trigger signal.

6. The resonant LLC switching power converter according to claim 3, characterized in that: The second control submodule includes: a second delay trigger module, whose first input terminal receives the second control signal, whose second input terminal receives the reference voltage, and whose third input terminal receives the first comparison signal, and is used to output a second weak current on-off control signal according to the second control signal, the reference voltage and the first comparison signal; The second driving module has an input end connected to the output end of the second delay trigger module, and is used to amplify and isolate the output second weak current on-off control signal and then output a second on-off control signal.

7. The resonant LLC switching power converter according to claim 6, characterized in that: The second delay trigger module includes: A second comparison module, whose first input terminal receives the second control signal and whose second input terminal receives the reference voltage, is used to delay the conversion of the second control signal into a second acquisition voltage, and output a second trigger signal when the second acquisition voltage is greater than the reference voltage; The second trigger module has a first input end receiving the first comparison signal and a second input end connected to the output end of the second comparison module, and is used for outputting the second weak current on-off control signal according to the second trigger signal.

8. The resonant LLC switching power converter according to claim 1, characterized in that: The first switch module is a first MOS tube, the second switch module is a second MOS tube, the first control signal includes a first voltage control signal and a second voltage control signal, and the second control signal includes a third voltage control signal and a fourth voltage control signal; The resonant LLC switching power converter further comprises: a first voltage dividing unit, wherein a first end of the first voltage dividing unit receives the first voltage control signal, a second end of the first voltage dividing unit is connected to the gate of the first MOS tube, a third end of the first voltage dividing unit is connected to the source of the first MOS tube and then receives the second voltage control signal, and controls the on and off of the first MOS tube according to the first voltage control signal and the second voltage control signal; a second voltage dividing unit, wherein a first end of the second voltage dividing unit receives the third voltage control signal, a second end of the second voltage dividing unit is connected to the gate of the second MOS tube, a third end of the second voltage dividing unit is connected to the source of the second MOS tube and then receives the fourth voltage control signal, and controls the on and off of the second MOS tube according to the third voltage control signal and the fourth voltage control signal; The second control module is used to control the third switch module to start conducting when it is detected that the output current is greater than the preset current and a high level signal is formed between the first voltage control signal and the second voltage control signal or a high level signal is formed between the second voltage control signal and the ground terminal of the second control module; The second control module is further configured to control the fourth switch module to start conducting when the output current is greater than a preset current and a high level signal is formed between the third voltage control signal and the fourth voltage control signal.

9. A control method for the resonant LLC switching power converter according to claim 1, characterized in that: The control method is applied to the control module, comprising: Outputting a first control signal to the first switch module to control the on / off state of the first switch module, and outputting a second control signal to the second switch module to control the on / off state of the second switch module; The output current of the resonant LLC switching power converter is obtained, the output current is compared with a preset current, and the on-off state of the third switch module is controlled according to the comparison result and the state of the first control signal, and the on-off state of the fourth switch module is controlled according to the comparison result and the state of the second control signal.

10. The control method according to claim 9, characterized in that: The obtaining the output current of the resonant LLC switching power converter comprises: The output current of the resonant LLC switching power converter is obtained by detecting the effective value of the AC component of the voltage across the second capacitor module.

11. The control method according to claim 9, characterized in that: The controlling the on / off state of the third switch module according to the current comparison result and the state of the first control signal comprises: When the output current is greater than the preset current and the first control signal is switched from the first level signal to the second level signal, the third switch module is controlled to switch from the first switch state to the second switch state with a delay, and the second switch state is maintained; When the output current is not greater than the preset current and the first control signal is switched from the first level signal to the second level signal, the third switch module is controlled to switch from the second switch state to the first switch state with delay, and the first switch state is maintained.

12. The control method according to claim 9, characterized in that: The controlling the on-off state of the fourth switch module according to the current comparison result and the state of the second control signal comprises: When the output current is greater than the preset current and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is controlled to switch from the first switch state to the second switch state with a delay, and the second switch state is maintained; When the output current is not greater than the preset current and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is controlled to switch from the second switch state to the first switch state with delay, and the first switch state is maintained.

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