Weak magnetic speed regulation control method and device during high-speed operation of IPMSM

Abstract

The invention relates to a flux-weakening speed regulation control method and a flux-weakening speed regulation control device during high-speed operation of an IPMSM. The method comprises the following steps: establishing a mathematical model of an IPMSM in a d-axis and q-axis rotating coordinate system; based on the mathematical model of the IPMSM in the d-axis and q-axis rotating coordinate system, performing field weakening control by using a gradient descent algorithm, determining a field weakening operation region where the driving motor is located, and correcting d-axis and q-axis stator current set values of the corresponding field weakening control region; and based on the mathematical model of the IPMSM in the d-axis and q-axis rotating coordinate system, the weak magnetic operation region where the driving motor is located and the corrected d-axis and q-axis stator current set values, performing integral saturation current closed-loop control on the driving motor. The defects in the prior art can be overcome, and the dynamic and static response performance of the motor driving system is improved.

Description

Technical Field

[0001] The present invention relates to the technical field of electric vehicles, and in particular to a magnetic field weakening speed regulation control method and device during high-speed operation of an IPMSM. Background Art

[0002] At present, major vehicle manufacturers are committed to breakthroughs in the speed of drive motors, and the continuous increase in speed means that the vehicle has a higher driving speed. For IPMSM (Interior Permanent Magnet Synchronous Motor), due to its advantages such as salient pole effect caused by the asymmetric d- and q-axis magnetic circuit structure and high mechanical strength, it is more conducive to realizing weak magnetic speed control, thereby widening the driving speed range of electric vehicles.

[0003] At present, the mainstream of IPMSM drive control is space vector control. For weak magnetic field control above the base speed of the drive motor, there are gradient descent algorithm, table lookup method and single current regulator control. The core idea is to re-plan the stator current vector trajectory under the dual constraints of the motor's voltage limit ellipse and current limit circle, that is, to change the magnitude of the current components on the d and q axes to weaken the excitation flux and realize the weak magnetic field speed increase process of IPMSM.

[0004] The existing electric vehicle IPMSM will face the following problems during the weak magnetic field acceleration process:

[0005] (1) The vehicle's load capacity decreases: During the weak magnetic field acceleration process, the increase in motor speed is accompanied by a decrease in output torque.

[0006] (2) Poor system speed regulation performance: Sudden changes in load torque will cause large fluctuations in motor speed.

[0007] (3) The current loop PI controller is prone to enter the integral saturation state, and the motor output torque fluctuates greatly. Due to the existence of the integral link, the traditional PI controller will continuously accumulate the deviation of the d-axis and q-axis currents when the electric vehicle is frequently accelerated and decelerated in actual operating conditions, causing the current controller to enter the saturation or even deep saturation state, resulting in the poor tracking response performance of the current loop, which is manifested as large current oscillations, and further causes the output torque fluctuation of the drive motor to increase.

[0008] Therefore, in order to improve the dynamic and static response performance of the motor drive system, it is urgent to realize a weak magnetic speed control method and device during high-speed operation of IPMSM. Summary of the invention

[0009] The object of the present invention is to provide a magnetic field weakening speed regulation control method and device during high-speed operation of an IPMSM, which can solve the deficiencies in the prior art and improve the dynamic and static response performance of a motor drive system.

[0010] To achieve the above object, the present invention adopts the following technical solutions:

[0011] A magnetic field weakening speed control method for IPMSM during high-speed operation, the method comprising the following steps:

[0012] S1. Establish the mathematical model of IPMSM in the d and q axis rotating coordinate system.

[0013] S2. Based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, the gradient descent algorithm is used to perform weak magnetic control, determine the weak magnetic operation area where the drive motor is located, and correct the d and q axis stator current setting values ​​in the corresponding weak magnetic control area.

[0014] S3. Based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values, the drive motor is subjected to anti-integral saturation current closed-loop control.

[0015] As a further improvement of the above technical solution, in step S1, the mathematical model of the IPMSM in the d-axis and q-axis rotating coordinate system is established, including:

[0016] S11. Use formula (1) to establish the voltage equation of IPMSM in the d and q axis rotating coordinate system:

[0017]

[0018] In formula (1), u d / u q 、i d / i q are d-axis and q-axis voltage and current respectively; L d / L q is the d and q axis inductance; R s is the three-phase stator resistance, p is the differential operator, ψ f is the permanent magnet flux, ω e is the electrical angular velocity;

[0019] S12. Use formula (2) to establish the electromagnetic torque equation of IPMSM in the d and q axis rotating coordinate system:

[0020]

[0021] In formula (2), n p is the pole pair number, T eis the electromagnetic torque;

[0022] S13. Use formula (3) to establish the motion equation of the drive motor of the IPMSM in the d and q axis rotating coordinate system:

[0023]

[0024] In formula (3), J is the moment of inertia, ω m is the mechanical angular velocity, T L is the load torque.

[0025] As a further improvement of the above technical solution, in step S2, based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, a gradient descent algorithm is used to perform magnetic weakening control, determine the magnetic weakening operation area where the drive motor is located, and correct the d and q axis stator current setting values ​​of the corresponding magnetic weakening control area, including:

[0026] S21, obtaining a torque increasing direction, a constant torque direction, and a voltage decreasing direction of the drive motor, and determining a weak magnetic operation area of ​​the drive motor according to the torque increasing direction, the constant torque direction, and the voltage decreasing direction;

[0027] S22. Determine a corresponding weak magnetic field control area according to the weak magnetic field operation area where the drive motor is located, and correct the set values ​​of the d-axis and q-axis stator currents in the weak magnetic field control area.

[0028] As a further improvement of the above technical solution, in step S21, the step of obtaining the torque increasing direction, the constant torque direction and the voltage decreasing direction of the drive motor, and determining the weak magnetic operation area of ​​the drive motor according to the torque increasing direction, the constant torque direction and the voltage decreasing direction, includes:

[0029] S211, using formula (4) to obtain the torque increasing direction of the drive motor:

[0030]

[0031] S212. Since the constant torque direction of the drive motor is orthogonal to the torque increasing direction, the constant torque direction of the drive motor is obtained using formula (5):

[0032]

[0033] S213, according to formula (5):

[0034]

[0035] The gradient descent algorithm is used to obtain the voltage decreasing direction, and its cost function is set as shown in formula (7):

[0036]

[0037] When the drive motor enters the field weakening control area and runs at high speed, ignoring the stator winding voltage drop, the voltage equation in the d / q coordinate system is shown in equation (8):

[0038]

[0039] Based on equations (6) and (7), equation (9) is used to obtain the decreasing direction of the voltage of the drive motor:

[0040]

[0041] According to formula (8), let

[0042]

[0043] S214, based on equations (5), (6), (9), and (10), use equation (11) to obtain the angle between the constant torque direction of the drive motor and the voltage decreasing direction The cosine value of the angle The cosine value of determines the weak magnetic operation area of ​​the drive motor:

[0044]

[0045] As a further improvement of the above technical solution, in step S214, the angle The cosine value of determines the weak magnetic operation area where the drive motor is located, including:

[0046] If the angle between the constant torque direction of the drive motor and the decreasing voltage direction is Right now Then the drive motor runs in the weak magnetic field zone 1;

[0047] If the angle between the constant torque direction of the drive motor and the decreasing voltage direction is Right now Then the drive motor enters the second field weakening zone and runs along the MTPV trajectory line;

[0048] The angle between the constant torque direction of the drive motor and the voltage decreasing direction The value range of is (0, π).

[0049] As a further improvement of the above technical solution, in step S22, the corresponding weak magnetic field control area is determined according to the weak magnetic field operation area where the drive motor is located, and the d-axis and q-axis stator current setting values ​​in the weak magnetic field control area are corrected, including:

[0050] S221, using formula (12) to obtain the voltage deviation amplitude is the d-axis and q-axis output voltage synthesis vector amplitude |U s |With the inverter limiting voltage Ulim The difference:

[0051] ΔU=U lim -|U s | (12)

[0052] In the formula,

[0053] S222, obtaining the weak magnetic field operation area where the drive motor is located, and determining the correction direction of the current setting value; if the drive motor is operating in the weak magnetic field area 1, the current setting value will be corrected along the constant torque direction; if the drive motor is operating in the weak magnetic field area 2, the current setting value will be corrected along the MTPV curve direction;

[0054] Among them, the MTPV curve is the line connecting the constant torque curve and the tangent point of the voltage limit ellipse, and the expression is shown in formula (13):

[0055]

[0056] Convert equation (13) to equation (14):

[0057] (L q -L d )L q 2 i q 2 -(L q -2L d )L d i d ψ f +L d ψ f 2 =0 (14)

[0058] Using equations (14) and (15), the MTPV direction is obtained, which is the tangent vector of the MTPV curve trajectory:

[0059]

[0060] According to formula (15), we can get:

[0061]

[0062] S223, based on equations (6), (12) and (16), the current setting value correction formulas for the weak magnetic field zone 1 and the weak magnetic field zone 2 are determined as follows, and the d-axis and q-axis stator current setting values ​​of the corresponding weak magnetic field control areas are corrected using the correction formulas;

[0063] 1) The expression for correcting the current setting value in the weak magnetic zone 1 is:

[0064]

[0065] 2) The expression for correcting the current setting value of the weak magnetic zone 2 is:

[0066]

[0067] Where α and β are the correction proportional coefficients of the current setting value.

[0068] As a further improvement of the above technical solution, in step S2, based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, a gradient descent algorithm is used to perform magnetic weakening control, determine the magnetic weakening operation area where the drive motor is located, and correct the d and q axis stator current setting values ​​of the corresponding magnetic weakening control area, and further include:

[0069] S23, when the deviation voltage ΔU>0, the weak magnetic control link does not work, that is, the correction current i d，m and i q，m are all set to 0; since in the weak magnetic process, the d-axis current will increase negatively, and the q-axis current needs to decrease accordingly, the limiting module is added to the q-axis using formula (19):

[0070]

[0071] S24. During the magnetic field weakening process, the input current of the current loop PI controller is obtained using formula (20):

[0072]

[0073] In the formula, i d-MTPA 、i q-MTPA are the d-axis and q-axis current reference values ​​output by the MTPA control module respectively; i d,m 、i q,m is the d and q axis current correction value output by the weak magnetic control module; i d * 、i q * is the corrected d and q axis current setting value.

[0074] As a further improvement of the above technical solution, in step S3, the anti-integral saturation current closed-loop control of the drive motor is performed based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values, including:

[0075] S31, based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values, obtain the input current deviation e1 of the drive motor and the corrected d axis output voltage u d* The relationship between the corrected d-axis output voltage u d * Whether the saturation value U is reached lim .

[0076] S32, according to the input current deviation e1 of the drive motor and the corrected d-axis output voltage u d * The relationship between the corrected d-axis output voltage u d * Whether the saturation value U is reached lim , determine whether to perform anti-integral saturation current closed-loop control.

[0077] As a further improvement of the above technical solution, in step S32, the input current deviation e1 of the driving motor and the corrected d-axis output voltage u d * The relationship between the corrected d-axis output voltage u d * Whether the saturation value U is reached lim , determine whether to perform anti-integral saturation current closed-loop control, including:

[0078] S321, if the corrected d-axis output voltage u of the drive motor d * Not reaching saturation value U lim , and the input current deviation e1 and the corrected d-axis output voltage u d * In the same direction, the selection switch selects the input signal as e1=i d * -i d , the anti-integral saturation link does not work; if the input current deviation e1 and the corrected d-axis output voltage u d * Reverse, the selection switch selects the input signal as e2=i d * -i d -K c (u d * -u d ), where K c The anti-integral saturation gain coefficient, the anti-integral saturation link starts to work, and the integral term is weakened by the anti-saturation gain link to prevent the current controller from entering a deep saturation state;

[0079] S322, if the corrected d-axis output voltage u d * reaches the saturation value, the input current deviation e1 and the corrected d-axis output voltage u d* In the same direction, in order to avoid deep integral saturation of the controller and the effect of the anti-integral saturation link, the integral link input is e2=i d * -i d -K c (u d * -u d ); If the input current deviation e1 and the corrected d-axis output voltage u d * In the reverse direction, the anti-integral link does not work, and the selection switch sets e1 = i d * -i d It is sent to the integration link as input signal.

[0080] The present invention also includes a weak magnetic speed control device for IPMSM when it is running at high speed, the device comprising: a model building module, a weak magnetic region determination and current setting value correction module and an anti-integral saturation current closed-loop control module;

[0081] The model building module is used to build a mathematical model of the IPMSM in the d-axis and q-axis rotating coordinate system.

[0082] The weak magnetic field area determination and current setting value correction module is used to perform weak magnetic field control based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system using a gradient descent algorithm, determine the weak magnetic field operation area where the drive motor is located, and correct the d and q axis stator current setting values ​​of the corresponding weak magnetic field control area.

[0083] The anti-integral saturation current closed-loop control module is used to perform anti-integral saturation current closed-loop control on the drive motor based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values.

[0084] Compared with the prior art, the advantages of the present invention are:

[0085] (1) The present invention adopts a weak magnetic speed control based on a gradient descent algorithm, and rationally plans the d and q current operating points of the motor on the current limit circle and the MTPV (Maximum Torque Per Voltage) control curve, which can achieve the maximum torque output by the motor at the same motor speed, thereby widening the speed regulation range of the motor and improving the torque output capacity of the motor. The weak magnetic control strategy based on the gradient descent algorithm can rationally and efficiently plan the d and q axis current operating points in the weak magnetic zone 1 and the weak magnetic zone 2.

[0086] (2) The present invention combines the advantages of the conditional integration method and the inverse calculation method, and adopts a variable structure anti-integral saturation current control to eliminate the integral saturation phenomenon of the traditional PI current controller. The d and q axis current loops adopt a variable structure anti-integral saturation current controller to eliminate the integral saturation phenomenon of the current controller and suppress the fluctuation of the output torque. BRIEF DESCRIPTION OF THE DRAWINGS

[0087] Figure 1 It is the planning diagram of the d-axis and q-axis current operating points of the driving motor in the weak magnetic zone 1 and the weak magnetic zone 2; T e1 , T e2 , T emax is the output electromagnetic torque of the motor, and the magnitude relationship is T e1 <T e2 <T emax ;

[0088] Figure 2 It is the torque and voltage direction diagram of the drive motor;

[0089] Figure 3 It is a magnetic field weakening control block diagram of the magnetic field weakening speed regulation control method during high-speed operation of the IPMSM described in the present invention;

[0090] Figure 4 It is the variable structure anti-integral saturation control block diagram described in the present invention;

[0091] Figure 5 It is a schematic diagram of a system simulation model of the magnetic field weakening speed regulation control method during high-speed operation of the IPMSM of the present invention;

[0092] Figure 6 It is the speed response curve of the drive motor;

[0093] Figure 7 It is the drive motor q 、i d Current response curve diagram;

[0094] Figure 8 It is the electromagnetic torque response curve of the drive motor. DETAILED DESCRIPTION

[0095] The present invention will be further described below in conjunction with the accompanying drawings:

[0096] like Figure 3 The weak magnetic speed control method for the IPMSM during high-speed operation shown in the figure comprises the following steps:

[0097] S1. Establish a mathematical model of IPMSM (salient-pole permanent magnet synchronous motor) in the d and q axis rotating coordinate system.

[0098] For the mathematical model of the motor, the calculation of the mathematical model in the three-phase stationary coordinate system (A / B / C) is complex. Through the coordinate vector transformation technology, following the principle that the synthetic magnetic motive force before and after the transformation is equal and the power is constant, it is transformed into a two-phase synchronous rotating coordinate system (d / q) for analysis. Generally, the mathematical model of the motor is represented by the voltage equation, electromagnetic torque equation and motion equation for easy analysis.

[0099] In the step S1, the mathematical model of the IPMSM in the d-axis and q-axis rotating coordinate system is established, including:

[0100] S11. Use formula (1) to establish the voltage equation of IPMSM in the d and q axis rotating coordinate system:

[0101]

[0102] In formula (1), u d / u q 、i d / i q are d-axis and q-axis voltage and current respectively; L d / L q is the d and q axis inductance; R s is the three-phase stator resistance, p is the differential operator, ψ f is the permanent magnet flux, ω e is the electrical angular velocity.

[0103] S12. Use formula (2) to establish the electromagnetic torque equation of IPMSM in the d and q axis rotating coordinate system:

[0104]

[0105] In formula (2), n p is the pole pair number, T e is the electromagnetic torque.

[0106] S13. Use formula (3) to establish the motion equation of the drive motor of the IPMSM in the d and q axis rotating coordinate system:

[0107]

[0108] In formula (3), J is the moment of inertia, ω m is the mechanical angular velocity, T L is the load torque.

[0109] S2. Based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, the gradient descent algorithm is used to perform weak magnetic control, determine the weak magnetic operation area where the drive motor is located, and correct the d and q axis stator current setting values ​​in the corresponding weak magnetic control area.

[0110] When the drive motor runs above the rated speed, its speed regulation range can be divided into weak magnetic zone 1 and weak magnetic zone 2. Weak magnetic zone 2 means that the motor can reach a higher speed, also known as the deep weak magnetic zone.

[0111] Figure 1 The AB section in the middle is the weak magnetic zone 1. The d and q axis current operating points are located on the current limit circle, which is the intersection of the motor output torque curve and the current limit circle. Point A corresponds to the peak torque T output by the motor. emax and rated speed ω N As the motor speed continues to increase, the motor output torque will continue to decrease. Point B is the dividing point between the weak magnetic zone 1 and the weak magnetic zone 2. Point B corresponds to the maximum output torque T of the weak magnetic zone 1. e2 and maximum speed ω c The BM segment is the second weak magnetic zone. The d and q axis current operating points are located on the MTPV (Maximum Torqueper Volt) control curve, which is the tangent point between the motor output torque curve and the voltage limit circle. Point C corresponds to the motor output torque T e2 and speed ω1.

[0112] The weak magnetic field control based on the gradient descent algorithm can realize the optimal path planning of the d and q axis currents in the entire weak magnetic field range of the drive motor, so that the maximum torque is output and the motor speed is the highest. The core part of this control strategy can be divided into two points: one is the determination of the weak magnetic field control area, and the other is the correction of the current setting value.

[0113] For speed control above the rated speed, a weak magnetic speed-up control method needs to be adopted. Returning to the d and q axis coordinate system, the magnitude of the current of the d and q axes is jointly limited by the current limiting circle and the voltage limiting ellipse. In order to ensure that the motor has a strong and stable torque output capability during the weak magnetic speed-up process, the present invention adopts a gradient descent algorithm. The gradient descent algorithm can realize the planning and correction of the d and q axis currents of the driving motor in the weak magnetic zone 1 and zone 2, so that iq and id are on the optimal path, which also achieves the minimum d and q axis currents while outputting the maximum torque, and widens the motor speed regulation range.

[0114] When in the weak magnetic field area, i d 、i q Planning along the current limit circle, the gradient descent algorithm can make the d and q axis current vectors corrected according to the constant torque direction, i d Negative increase, i q The current vector and magnitude remain unchanged. When in the weak magnetic zone II, i d 、i q Planning towards the center of the voltage limit ellipse, the gradient descent algorithm can make i d 、i qCorrection is made along the MTPV curve, that is, the tangent point of the voltage limit ellipse and the constant torque. This planning path is the optimal path. While the speed is constantly increasing, a certain torque output capacity can be guaranteed. Reasonable and efficient d and q axis current planning enables the motor to have good speed regulation performance during the weak magnetic field speed increase process.

[0115] In step S2, the method of performing magnetic field weakening control based on the mathematical model of IPMSM in the d-axis and q-axis rotating coordinate systems and using a gradient descent algorithm to determine the magnetic field weakening operation area where the drive motor is located and to correct the d-axis and q-axis stator current setting values ​​in the corresponding magnetic field weakening control area includes:

[0116] S21, obtaining a torque increasing direction, a constant torque direction, and a voltage decreasing direction of the drive motor, and determining a weak magnetic operation area of ​​the drive motor according to the torque increasing direction, the constant torque direction, and the voltage decreasing direction.

[0117] Figure 2 The decreasing direction of the voltage points to the center point M of the voltage limit ellipse, the constant torque direction is the tangent vector of the constant torque curve, and the angle between the two arrows is The torque increasing direction and the constant torque direction are orthogonal to each other.

[0118] Use formula (4) to obtain the torque increasing direction of the drive motor:

[0119]

[0120] Since the constant torque direction of the drive motor is orthogonal to the torque increasing direction, the constant torque direction of the drive motor is obtained using formula (5):

[0121]

[0122] According to formula (5), let:

[0123]

[0124] The voltage decreasing direction is obtained by using the gradient descent algorithm. Assume that its cost function is as shown in formula (7):

[0125]

[0126] When the drive motor enters the weak magnetic control area and runs at high speed, the stator winding voltage drop can be ignored, and the voltage equation in the d / q coordinate system is shown in equation (8):

[0127]

[0128] According to equations (6) and (7), the decreasing direction of the voltage of the drive motor is obtained:

[0129]

[0130] According to formula (8), let

[0131]

[0132] According to equations (5), (6), (9) and (10), equation (11) is used to obtain the angle between the constant torque direction of the drive motor and the voltage decreasing direction: The cosine value of the angle The cosine value of determines the weak magnetic operation area of ​​the drive motor:

[0133]

[0134] when When (i.e. ), the drive motor runs in the weak magnetic field area 1; when When (i.e. ), the drive motor enters the weak magnetic zone 2 and runs along the MTPV trajectory. The size of is (0, π).

[0135] S22. Determine a corresponding weak magnetic field control area according to the weak magnetic field operation area where the drive motor is located, and correct the set values ​​of the d-axis and q-axis stator currents in the weak magnetic field control area.

[0136] After determining the weak magnetic field operation area where the drive motor is located, the d and q axis stator current setting values ​​of the corresponding weak magnetic field area need to be corrected. The magnitude of the current setting value is calculated by combining the voltage decreasing and constant torque direction information with the voltage deviation amplitude. Among them, the voltage deviation amplitude is the d and q axis output voltage composite vector amplitude |U s |With the inverter limiting voltage U lim The difference can be expressed as:

[0137] ΔU=U lim -|U s | (12)

[0138] In the formula,

[0139] When the drive motor runs in the weak magnetic field zone 1, the current setting value will be corrected along the constant torque direction, that is, formula (5).

[0140] When the drive motor runs in the second field weakening zone, the current setting value will be corrected along the MTPV curve. The MTPV curve is the line connecting the constant torque curve and the tangent point of the voltage limit ellipse, which can be written as:

[0141]

[0142] After finishing, we can get:

[0143] (L q -L d )L q 2 i q 2 -(L q -2L d )L d i d ψ f +L d ψ f 2 =0 (14)

[0144] The MTPV direction is the tangent vector of its trajectory, which can be obtained from formula (14):

[0145]

[0146] According to formula (15), we can get:

[0147]

[0148] Therefore, when the drive motor enters the second field weakening zone, the current setting value will be corrected using formula (15).

[0149] By combining equations (6), (12) and (16), the current setting value correction formulas for the weak magnetic field zone 1 and the weak magnetic field zone 2 are obtained as follows:

[0150] 1) The expression for correcting the current setting value in the weak magnetic zone 1 is:

[0151]

[0152] 2) The expression for correcting the current setting value of the weak magnetic zone 2 is:

[0153]

[0154] Where α and β are the correction proportional coefficients of the current setting value.

[0155] When the deviation voltage ΔU>0, the weak magnetic control link does not work, that is, the correction current i d，m and i q，m Both are set to 0. Since the d-axis current will increase negatively during the weak magnetic process, and the q-axis current needs to decrease accordingly, it is necessary to add a limiting module on the q-axis, and its value is:

[0156]

[0157] During the field weakening process, the input current of the current loop PI controller can be expressed as:

[0158]

[0159] In the formula, i d-MTPA 、i q-MTPA are the d-axis and q-axis current reference values ​​output by the MTPA (Maximum Torque Per Ampere) control module respectively; i d，m 、i q，m is the d and q axis current correction value output by the weak magnetic control module; i d * 、i q * is the corrected d and q axis current setting value.

[0160] The weak magnetic field control strategy based on the gradient descent algorithm can reasonably and efficiently plan the stator current vector trajectory of the weak magnetic field zone 2, so that the d-axis and q-axis current operating points are distributed along the MTPV control curve. Therefore, the maximum torque output capacity can be ensured while widening the speed regulation range of the drive motor.

[0161] S3. Based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values, the drive motor is subjected to anti-integral saturation current closed-loop control.

[0162] In order to solve the problem that the traditional PI controller, under the condition of sudden changes in load and speed, the current loop gain is too large due to the existence of the integral link, accompanied by current oscillation, and then the output torque pulsation is large, which affects the performance of the entire speed regulation system, the present invention proposes an anti-integral saturation current PI control, which combines the advantages of the conditional integral method and the inverse calculation method. Specifically, it is determined whether the integral saturation condition is reached through the judgment basis. If it is not reached, the anti-integral saturation function is not turned on; if it is reached, the anti-integral saturation link is input through the Switch. Through the above design, it is intended to prevent the PI current controller from entering a deep integral saturation state.

[0163] When the drive motor enters the weak magnetic control area, the sudden changes in load and speed will cause a large change in the stator current vector trajectory, which will cause a large gain in the d and q axis current loops. Excessive gain will cause a deviation between the reference input and feedback of the current. The traditional PI controller lacks restrictions on integral saturation. The integral link will continue to accumulate this error, eventually causing the current controller to enter integral saturation or even deep saturation, causing the system speed regulation performance to deteriorate further.

[0164] When the current controller enters a deep saturation state, if the deviation signal of the current loop is reversed, the controller will exit saturation, but it requires a certain adjustment time, which will cause the current controller to be in a saturated state for a longer period of time, resulting in poor tracking response performance of the current loop, accompanied by larger current oscillations, which in turn causes the output torque pulsation of the drive motor to increase.

[0165] In order to eliminate the integral saturation phenomenon of the current controller and suppress the pulsation of the output torque, the present invention combines the advantages of the conditional integral method and the inverse calculation method to propose a method such as Figure 4 The variable structure anti-integral saturation current control is shown.

[0166] In step S3, the anti-integral saturation current closed-loop control of the drive motor is performed based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values, including:

[0167] S31, based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values, obtain the input current deviation e1 of the drive motor and the corrected d axis output voltage u d * The relationship between the corrected d-axis output voltage u d * Whether the saturation value U is reached lim .u d * is the corrected d-axis output voltage, which can be seen from equation (1) and Figure 6 ,u d * It is calculated by the PI current controller and needs no special explanation. It is a relatively basic motor modeling.

[0168] S32, according to the input current deviation e1 of the drive motor and the corrected d-axis output voltage u d * The relationship between the corrected d-axis output voltage u d * Whether the saturation value U is reached lim , determine whether to perform anti-integral saturation current closed-loop control.

[0169] S321, if the corrected d-axis output voltage u of the drive motor d * Not reaching saturation value U lim , and the input current deviation e1 and the corrected d-axis output voltage u d * In the same direction, the selection switch selects the input signal as e1=i d* -i d , the anti-integral saturation link does not work; if the input current deviation e1 and the corrected d-axis output voltage u d * Reverse, the selection switch selects the input signal as e2=i d * -i d -K c (u d * -u d ), where K c The anti-integral saturation gain coefficient is used, and the anti-integral saturation link starts to work. The integral term is weakened by the anti-saturation gain link to prevent the current controller from entering a deep saturation state.

[0170] S322, if the corrected d-axis output voltage u d * reaches the saturation value, the input current deviation e1 and the corrected d-axis output voltage u d * In the same direction, in order to avoid deep integral saturation of the controller and the effect of the anti-integral saturation link, the integral link input is e2=i d * -i d -K c (u d * -u d ); If the input current deviation e1 and the corrected d-axis output voltage u d * In the reverse direction, the anti-integral link does not work, and the selection switch sets e1 = i d * -i d It is sent to the integration link as input signal.

[0171] like Figure 4 As shown in the figure, the anti-integral saturation control is performed by judging the input current deviation e1 of the drive motor and the corrected d-axis output voltage u d * The relationship between the d-axis output voltage u d * Whether the saturation value U is reached lim , to decide whether to open the anti-integral saturation link. If the judgment condition of the Switch module input is greater than zero, e2 is selected to be sent to the integral link; otherwise, the integral link input signal is e1. Table 1 is the action relationship table of the anti-integral saturation link.

[0172] Table 1 Anti-integral saturation link action relationship table

[0173] Integration stage input Judgment basis Anti-integral windup link <![CDATA[e1=i d * -i d ]]> <![CDATA[|u d * | lim And e1·u d * >0]]> ​ No effect <![CDATA[e2=i d * -and d -K c (in d * -in d )]]> <![CDATA[|u d * | lim And i d * ·u d * <0]]> ​ effect <![CDATA[e2=i d * -and d -K c (in d * -in d )]]> <![CDATA[|u d * |≥U lim And i d * ·u d * >0]]> effect <![CDATA[e1=i d * -i d ]]> <![CDATA[|u d * |≥U lim And i d * ·u d * <0]]> No effect

[0174] When the corrected d-axis output voltage u of the drive motor d * The saturation value is not reached, and the input current deviation e1 is different from the corrected d-axis output voltage u d * When the direction is the same, the Switch module selects the input signal as e1=i d * -i d , the anti-integral saturation link does not work; if the input current deviation e1 and the corrected d-axis output voltage u d * Reverse, the Switch module selects the input signal as e2=i d * -i d -K c (u d * -u d ), where K c The anti-integral saturation gain coefficient starts to work, and the integral term is weakened by the anti-saturation gain link, thereby preventing the current controller from entering a deep saturation state.

[0175] When the corrected d-axis output voltage u d * reaches the saturation value, the input current deviation e1 and the corrected d-axis output voltage u d * In the same direction, in order to avoid the deep integral saturation of the controller and the anti-integral saturation link, the integral link input is e2=i d * -i d -K c (u d * -u d ); If the input current deviation e1 and the corrected d-axis output voltage u d * In the reverse direction, the anti-integral link does not work, and the selection switch sets e1 = i d * -i d It is sent to the integration link as input signal.

[0176] The present invention also includes a weak magnetic speed control device for IPMSM when it is running at high speed, the device comprising: a model building module, a weak magnetic region determination and current setting value correction module and an anti-integral saturation current closed-loop control module;

[0177] The model building module is used to build a mathematical model of the IPMSM in the d-axis and q-axis rotating coordinate system.

[0178] The weak magnetic field area determination and current setting value correction module is used to perform weak magnetic field control based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system using a gradient descent algorithm, determine the weak magnetic field operation area where the drive motor is located, and correct the d and q axis stator current setting values ​​of the corresponding weak magnetic field control area.

[0179] The anti-integral saturation current closed-loop control module is used to perform anti-integral saturation current closed-loop control on the drive motor based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values.

[0180] In order to verify the performance of the magnetic field weakening speed control method of the IPMSM during high-speed operation of the present invention, the following Figure 5 The overall simulation model of the algorithm shown in the figure was simulated and analyzed. The drive motor runs at a constant speed and the load torque changes suddenly. The target speed is set to 7000r / min. The motor starts running without load at the beginning. At 0.4s, the load torque suddenly increases to 5N·m. The simulation results are shown in Figure 6 , 7 , as shown in Figure 8.

[0181] Figure 6 , Figure 7 and Figure 8 They are the speed waveform, d-axis and q-axis current waveforms, and the output electromagnetic torque waveform of the drive motor under the Simulink simulation conditions. Under the weak magnetic field control based on the gradient descent algorithm, the motor is in the weak magnetic field speed-up process, such as Figure 6 As shown in the figure, the motor can respond to the given speed of 7000r / min in a short time (0.18s). Figure 6 As shown in the enlarged part, when the load torque suddenly increases to 5N·m in 0.4S, it can be seen that the speed fluctuation amplitude recovers to the given speed after about 8.2r / min. Figure 6 The fluctuation of the red curve in the enlarged part can be obtained. And when the motor runs with a load of 5N·m, its speed fluctuation is small and it can run stably. Figure 7 As shown in Figure 2, when the load torque suddenly increases to 5 N·m at 0.4S, the oscillation amplitude of the d and q axis currents of the drive motor is small during operation. Figure 7 The enlarged part shows that i q 、i d The fluctuation ranges are 1.9~2.5A and -28.5~-27A respectively. It can be seen that under the weak magnetic control described in the present invention, the d and q axis current oscillation phenomena are greatly suppressed, and the current vector amplitude is relatively small. Figure 8 As shown in the figure, when the motor is running with a load of 5N·m, its output torque can respond quickly to the load. Figure 8From the enlarged part, we can see that the torque fluctuation range is around ±0.8N·m.

[0182] In summary, the present invention proposes a method and device for weak magnetic speed control during high-speed operation of IPMSM. The weak magnetic control strategy based on the gradient descent algorithm and the anti-integral saturation controller designed for the current loop adopted by the method and the device can better realize the weak magnetic speed increase of the drive motor and have good speed control performance. In addition, the current and torque oscillation phenomena that occur during the weak magnetic speed increase of the motor can be better suppressed, thereby ensuring that the IPMSM outputs a stable electromagnetic torque while widening the speed regulation range, meeting the needs of actual working conditions such as high-speed driving and overtaking of electric vehicles.

[0183] The above-described embodiments are merely descriptions of preferred implementations of the present invention and are not intended to limit the scope of the present invention. Without departing from the design spirit of the present invention, various modifications and improvements made to the technical solutions of the present invention by ordinary technicians in this field should all fall within the scope of protection determined by the claims of the present invention.

Claims

1. A magnetic field weakening speed control method for IPMSM during high-speed operation, characterized in that: The method comprises the following steps: S1. Establish the mathematical model of IPMSM in the d and q axis rotating coordinate system; S2. Based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, the gradient descent algorithm is used to perform magnetic weakening control, determine the magnetic weakening operation area where the drive motor is located, and correct the d and q axis stator current setting values ​​in the corresponding magnetic weakening control area; S3. Based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values, the drive motor is subjected to anti-integral saturation current closed-loop control.

2. The magnetic field weakening speed regulation control method during high-speed operation of the IPMSM according to claim 1 is characterized in that: In the step S1, the mathematical model of the IPMSM in the d-axis and q-axis rotating coordinate system is established, including: S11. Use formula (1) to establish the voltage equation of IPMSM in the d and q axis rotating coordinate system: In formula (1), u d / u q 、i d / i q are d-axis and q-axis voltage and current respectively; L d / L q is the d and q axis inductance; R s is the three-phase stator resistance, p is the differential operator, ψ f is the permanent magnet flux, ω e is the electrical angular velocity; S12. Use formula (2) to establish the electromagnetic torque equation of IPMSM in the d and q axis rotating coordinate system: In formula (2), n p is the pole pair number, T e is the electromagnetic torque; S13. Use formula (3) to establish the motion equation of the drive motor of the IPMSM in the d and q axis rotating coordinate system: In formula (3), J is the moment of inertia, ω m is the mechanical angular velocity, T L is the load torque.

3. The magnetic field weakening speed regulation control method during high-speed operation of the IPMSM according to claim 1 is characterized in that: In step S2, based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, a gradient descent algorithm is used to perform magnetic weakening control, determine the magnetic weakening operation area where the drive motor is located, and correct the d and q axis stator current setting values ​​of the corresponding magnetic weakening control area, including: S21, obtaining a torque increasing direction, a constant torque direction, and a voltage decreasing direction of the drive motor, and determining a weak magnetic operation area of ​​the drive motor according to the torque increasing direction, the constant torque direction, and the voltage decreasing direction; S22. Determine a corresponding weak magnetic field control area according to the weak magnetic field operation area where the drive motor is located, and correct the set values ​​of the d-axis and q-axis stator currents in the weak magnetic field control area.

4. The magnetic field weakening speed regulation control method during high-speed operation of the IPMSM according to claim 3 is characterized in that: In step S21, the step of acquiring the torque increasing direction, the constant torque direction, and the voltage decreasing direction of the drive motor, and determining the weak magnetic operation area of ​​the drive motor according to the torque increasing direction, the constant torque direction, and the voltage decreasing direction, includes: S211, using formula (4) to obtain the torque increasing direction of the drive motor: S212. Since the constant torque direction of the drive motor is orthogonal to the torque increasing direction, the constant torque direction of the drive motor is obtained using formula (5): S213, according to formula (5): The gradient descent algorithm is used to obtain the voltage decreasing direction, and its cost function is set as shown in formula (7): When the drive motor enters the field weakening control area and runs at high speed, ignoring the stator winding voltage drop, the voltage equation in the d / q coordinate system is shown in equation (8): Based on equations (6) and (7), equation (9) is used to obtain the decreasing direction of the voltage of the drive motor: According to formula (8), let S214, based on equations (5), (6), (9), and (10), use equation (11) to obtain the angle between the constant torque direction of the drive motor and the voltage decreasing direction The cosine value of the angle The cosine value of determines the weak magnetic operation area of ​​the drive motor:

5. The magnetic field weakening speed regulation control method during high-speed operation of IPMSM according to claim 4 is characterized in that: In step S214, the angle The cosine value of determines the weak magnetic operation area where the drive motor is located, including: If the angle between the constant torque direction of the drive motor and the decreasing voltage direction is Right now Then the drive motor runs in the weak magnetic field zone 1; If the angle between the constant torque direction of the drive motor and the decreasing voltage direction is Right now Then the drive motor enters the second field weakening zone and runs along the MTPV trajectory line; The angle between the constant torque direction of the drive motor and the voltage decreasing direction The value range of is (0, π).

6. The magnetic field weakening speed regulation control method during high-speed operation of the IPMSM according to claim 5 is characterized in that: In the step S22, the step of determining the corresponding weak magnetic field control area according to the weak magnetic field operation area where the drive motor is located, and correcting the d-axis and q-axis stator current setting values ​​in the weak magnetic field control area includes: S221, using formula (12) to obtain the voltage deviation amplitude is the d-axis and q-axis output voltage synthesis vector amplitude |U s |With the inverter limiting voltage U lim The difference: ΔU=U lim -|U s | (12) In the formula, S222, obtaining the weak magnetic field operation area where the drive motor is located, and determining the correction direction of the current setting value; if the drive motor is operating in the weak magnetic field area 1, the current setting value will be corrected along the constant torque direction; if the drive motor is operating in the weak magnetic field area 2, the current setting value will be corrected along the MTPV curve direction; Among them, the MTPV curve is the line connecting the constant torque curve and the tangent point of the voltage limit ellipse, and the expression is shown in formula (13): Convert equation (13) to equation (14): (L q -L d )L q 2 i q 2 -(L q -2L d )L d i d ψ f +L d ψ f 2 =0 (14) Using equations (14) and (15), the MTPV direction is obtained, which is the tangent vector of the MTPV curve trajectory: According to formula (15), we can get: S223, based on equations (6), (12) and (16), the current setting value correction formulas for the weak magnetic field zone 1 and the weak magnetic field zone 2 are determined as follows, and the d-axis and q-axis stator current setting values ​​of the corresponding weak magnetic field control areas are corrected using the correction formulas; 1) The expression for correcting the current setting value in the weak magnetic zone 1 is: 2) The expression for correcting the current setting value of the weak magnetic zone 2 is: Where α and β are the correction proportional coefficients of the current setting value.

7. The magnetic field weakening speed regulation control method during high-speed operation of the IPMSM according to claim 6 is characterized in that: In the step S2, based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system, a gradient descent algorithm is used to perform magnetic weakening control, determine the magnetic weakening operation area where the drive motor is located, and correct the d and q axis stator current setting values ​​of the corresponding magnetic weakening control area, which also includes: S23, when the deviation voltage ΔU>0, the weak magnetic control link does not work, that is, the correction current i d，m and i q，m are all set to 0; since in the weak magnetic process, the d-axis current will increase negatively, and the q-axis current needs to decrease accordingly, the limiting module is added to the q-axis using formula (19): S24. During the magnetic field weakening process, the input current of the current loop PI controller is obtained using formula (20): In the formula, i d-MTPA 、i q-MTPA are the d-axis and q-axis current reference values ​​output by the MTPA control module respectively; i d-m 、i q-m is the d and q axis current correction value output by the weak magnetic control module; i d * 、i q * is the corrected d and q axis current setting value.

8. The magnetic field weakening speed regulation control method during high-speed operation of the IPMSM according to claim 7, characterized in that: In step S3, the anti-integral saturation current closed-loop control of the drive motor is performed based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values, including: S31, based on the mathematical model of IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values, obtain the input current deviation e1 of the drive motor and the corrected d axis output voltage u d * The relationship between the corrected d-axis output voltage u d * Whether the saturation value U is reached lim ; S32, according to the input current deviation e1 of the drive motor and the corrected d-axis output voltage u d * The relationship between the corrected d-axis output voltage u d * Whether the saturation value U is reached lim , determine whether to perform anti-integral saturation current closed-loop control.

9. The magnetic field weakening speed regulation control method during high-speed operation of the IPMSM according to claim 8, characterized in that: In step S32, the input current deviation e1 of the driving motor and the corrected d-axis output voltage u d * The relationship between the corrected d-axis output voltage u d * Whether the saturation value U is reached lim , determine whether to perform anti-integral saturation current closed-loop control, including: S321, if the corrected d-axis output voltage u of the drive motor d * Not reaching saturation value U lim , and the input current deviation e1 and the corrected d-axis output voltage u d * In the same direction, the selection switch selects the input signal as e1=i d * -i d , the anti-integral saturation link does not work; if the input current deviation e1 and the corrected d-axis output voltage u d * Reverse, the selection switch selects the input signal as e2=i d * -i d -K c (u d * -u d ), where K c The anti-integral saturation gain coefficient, the anti-integral saturation link starts to work, and the integral term is weakened by the anti-saturation gain link to prevent the current controller from entering a deep saturation state; S322, if the corrected d-axis output voltage u of the drive motor d * reaches the saturation value, the input current deviation e1 and the corrected d-axis output voltage u d * In the same direction, in order to avoid deep integral saturation of the controller and the effect of the anti-integral saturation link, the integral link input is e2=i d * -i d -K c (u d * -u d ); If the input current deviation e1 and the corrected d-axis output voltage u d * In the reverse direction, the anti-integral link does not work, and the selection switch sets e1 = i d * -i d It is sent to the integration link as input signal.

10. A weak magnetic speed control device for IPMSM during high-speed operation, characterized in that: The device comprises: a model building module, a weak magnetic region determination and current setting value correction module and an anti-integral saturation current closed-loop control module; The model building module is used to build a mathematical model of the IPMSM in the d and q axis rotating coordinate system; The weak magnetic region determination and current setting value correction module is used to perform weak magnetic control based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system using a gradient descent algorithm, determine the weak magnetic operation region where the drive motor is located, and correct the d and q axis stator current setting values ​​of the corresponding weak magnetic control region; The anti-integral saturation current closed-loop control module is used to perform anti-integral saturation current closed-loop control on the drive motor based on the mathematical model of the IPMSM in the d and q axis rotating coordinate system, the weak magnetic operation area where the drive motor is located, and the corrected d and q axis stator current setting values.

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