Low-complexity noise-tolerant laser phase recovery method

Abstract

The invention discloses a low-complexity noise-tolerant laser phase recovery method, and belongs to the technical field of signal processing. Two-stage structure joint estimation is used, joint estimation of the frequency offset and the phase of the signal carrier is achieved through feedback stage phase estimation and feedforward stage phase estimation, the problem of phase jump of a traditional Diff-4th algorithm at the low signal-to-noise ratio is solved, and the carrier phase recovery performance is improved. Based on a digital phase-locked loop and Viterbi-Viterbi phase estimation, compared with a traditional Diff-4th method, the calculation complexity is reduced, and calculation resources are reduced. The phase discriminator, the loop filter and the numerically-controlled oscillator form a feedback loop, Doppler frequency shift caused by relative movement of satellites is tracked continuously in real time by combining a feed-forward mechanism, a feedback mechanism and carrier phase estimation rapid adjustment, constellation diagram rotation caused by residual Doppler frequency shift and phase noise in an inter-satellite coherent optical communication system can be eliminated, and the system performance is improved. Compared with a traditional Diff-4th method, the stability and reliability of communication are improved.

Description

Technical Field

[0001] The invention relates to a low-complexity noise-tolerant laser phase recovery method, belonging to the technical field of signal processing. Background Art

[0002] Future space activities will have increasing requirements for data transmission rate and security. Satellite coherent optical communication has the characteristics of large communication capacity and high receiving sensitivity, and has high spectrum efficiency under high-order modulation. However, the phase noise of the laser and the Doppler frequency deviation caused by the relative motion of the satellite will lead to a decrease in the performance of transmission data recovery, so it is necessary to estimate and compensate for the Doppler frequency deviation and phase noise.

[0003] Frequency offset estimation is divided into frequency domain estimation and time domain estimation. Frequency domain estimation estimates the frequency offset by performing Fourier transform on the signal to obtain its frequency information, which is called frequency offset estimation based on fast Fourier transform (FFT-FOE). However, the computational complexity in high-speed communication scenarios is too high and is not suitable for resource-constrained satellite payloads. Time domain estimation estimates the frequency offset using the phase difference information between adjacent symbols, which is called differential frequency offset estimation (Diff-FOE). Block estimation is used to smooth the estimated differential phase to reduce the impact of noise. Diff-FOE has a small amount of computation, but severe burst errors may occur at low signal-to-noise ratios.

[0004] Phase estimation methods mainly include blind estimation and data-assisted estimation methods. Blind estimation methods do not rely on pilot information, have the characteristics of high spectrum efficiency and resource utilization, and because they do not involve the probability calculation required by data-assisted methods, their computational complexity is smaller. Blind estimation methods mainly include Viterbi-Viterbi estimation algorithm (VV-PE) and blind phase search (BPS). BPS quantizes the phase space into a phase test set, and performs the same calculation search for each test phase to minimize the phase with the minimum cost function. The phase estimation accuracy is proportional to the amount of calculation. VV-PE estimates the carrier phase by eliminating the modulation signal, and has the advantages of simple structure and small amount of calculation.

[0005] At present, satellite laser communications mainly use binary phase shift keying (BPSK) and quadrature phase shift keying (QPSK) modulation. Due to the constraints of onboard resources, traditional carrier recovery technology uses Diff-FOE combined with VV-PE (Diff-4th) to estimate the carrier frequency offset and phase, but both are feedforward methods. After cascading, the system performance is severely lost at a low signal-to-noise ratio. Summary of the invention

[0006] In order to solve the problem of serious system performance loss at a low signal-to-noise ratio, the purpose of the present invention is to provide a low-complexity noise-tolerant laser phase recovery method, which can eliminate the constellation rotation caused by residual Doppler frequency offset and phase noise in the intersatellite coherent optical communication system and realize low-complexity noise-tolerant laser phase recovery. The present invention also has the advantages of low computational complexity, reduced requirements on hardware performance, and saving multiplier resources.

[0007] The objectives of the present invention are achieved through the following technical solutions.

[0008] The present invention discloses a low-complexity noise-tolerant laser phase recovery method, which realizes joint estimation of the frequency deviation and phase of a signal carrier through feedback phase estimation and feedforward phase estimation, and includes the following steps:

[0009] Step 1: Laser signal Perform phase detection, phase It is expressed as:

[0010]

[0011] Step 2: Phase of Step 1 and phase detector gain K d Multiplication is the phase error K d θ in , the second-order loop filter is K d θ in Perform noise reduction to obtain a smooth signal θ loop_0 ;

[0012] θ loop_0 The z-domain representation Y(z) is:

[0013] Y(z)=H(z)X(z)

[0014] Where X(z) is K d θ in The z-domain representation of ; H(z) is the transfer function of the second-order loop filter, expressed as:

[0015]

[0016] Among them, K1 and K2 are the gain factors of the loop filter, which affect the response speed of the loop and the ability to filter out noise;

[0017] Step 3: The numerically controlled oscillator is based on the γ loop_0 Calculate the carrier phase estimate

[0018] The z-domain representation Y1(z) is:

[0019] Y1(z)=H1(z)Y(z)

[0020] Where H1(z) is the transfer function of the digital controlled oscillator, expressed as:

[0021]

[0022] Step 4: Phase of Step 1 Subtract the estimate from step 3 Get the error value θ d , phase detector gain K d and error value θ d Multiplication is the phase error K d θ d ; Second-order loop filter for K d θ d Perform noise reduction to obtain a smooth signal θ loop ;

[0023] θ loop The z-domain representation Y′(z) is:

[0024] Y′(z)=H(z)X′(z)

[0025] Where X′(z) is K d θ d The z-domain representation of

[0026] Step 5: According to the θ in step 4 loop Calculate the carrier phase estimate

[0027] The z-domain representation Y1′(z) is:

[0028] Y1′(z)=H1(z)Y′(z)

[0029] Step 6: The phase detector, loop filter and digitally controlled oscillator form a feedback loop, and repeat steps 4 and 5, that is, use step 5 to output Replace step 4 Perform iterative updates and output the carrier phase estimate at each moment The iteration stops when there is no signal input to the phase detector;

[0030] Step 7: According to the phase error K of step 4 d θ d , the phase estimation value of the feedforward stage at the current moment is calculated by the block estimation method

[0031]

[0032] Where k represents the signal sampling point number, and N represents the phase estimation window length;

[0033] Step 8: The phase estimate at the current iteration time of step 6 and the feedforward phase estimate at the current moment in step 7 Add up to get the carrier phase estimate at the current moment

[0034] Step 9: Calculate the signal's carrier phase estimation value in real time by repeating steps 6 to 8.

[0035] Furthermore, the transfer function of the feedback loop in step six is ​​expressed as:

[0036]

[0037] Where K d , K1 and K2 and the characteristic frequency ω of the feedback loop n , the damping factor ξ parameter is related. When the product of the characteristic frequency and the symbol period T is much less than 1, that is, ω n T<<1, three variables K d , K1, K2 and ω n ,ξ is expressed as:

[0038]

[0039]

[0040] The symbol period T is determined by the system design, and the empirical value of ξ is The values ​​of K1 and K2 depend on K d and characteristic frequency ω n ,ω n The empirical conditions are:

[0041]

[0042] Beneficial effects:

[0043] 1. The present invention discloses a low-complexity noise-tolerant laser phase recovery method, which realizes the joint estimation of the frequency deviation and phase of the signal carrier through feedback phase estimation and feedforward phase estimation, solves the problem of phase jump in the traditional Diff-4th algorithm at low signal-to-noise ratio, and improves the carrier phase recovery performance.

[0044] 2. The present invention discloses a low-complexity noise-tolerant laser phase recovery method based on a digital phase-locked loop and Viterbi-Viterbi phase estimation, which reduces the burst errors occurring during the phase recovery process under low signal-to-noise ratio. Compared with the traditional Diff-4th method, it reduces the computational complexity and reduces computing resources.

[0045] 3. The present invention discloses a low-complexity noise-tolerant laser phase recovery method, in which a phase detector, a loop filter and a numerically controlled oscillator form a feedback loop, which combines a feedforward mechanism and a feedback mechanism to achieve rapid adjustment of carrier phase estimation, and continuously and in real time track the Doppler frequency shift caused by the relative motion of satellites. It can eliminate the constellation rotation caused by residual Doppler frequency deviation and phase noise in the inter-satellite coherent optical communication system, and improve the stability and reliability of communication compared with the traditional Diff-4th method. BRIEF DESCRIPTION OF THE DRAWINGS

[0046] Figure 1 It is a schematic diagram of the composition structure and signal processing of the laser communication receiving end in the present invention;

[0047] Figure 2 It is a schematic flow chart of a low-complexity noise-tolerant laser phase recovery method in the present invention;

[0048] Figure 3 This is a comparison diagram of the logarithmic root mean square error of a low-complexity noise-tolerant laser phase retrieval method in the present invention and a traditional Diff-4th method phase estimation;

[0049] Figure 4 This is a BER comparison chart between a low-complexity noise-tolerant laser phase recovery method in the present invention and a traditional Diff-4th method. DETAILED DESCRIPTION

[0050] The invention is further described and described in detail below with reference to the accompanying drawings and examples.

[0051] Embodiment 1:

[0052] Input the signal as Figure 1 In the digital signal processing process shown in FIG, two analog electrical signals I(t) and Q(t) are digitized by ADC, enter FPGA for matched filtering, clock recovery, channel equalization, and then enter the proposed carrier recovery method. For the carrier phase recovery of the signal under QPSK modulation, the satellite coherent optical communication process is simulated, such as Figure 2 As shown, a low-complexity noise-tolerant laser phase recovery method disclosed in this embodiment calculates the input signal phase estimation value by forming a loop through feedback phase estimation and feedforward phase phase estimation. The specific implementation steps are as follows:

[0053] Step 1: The laser signal receiving module receives the laser signal, the laser wavelength is 1550nm, and the modulated signal received by the optical antenna is expressed as:

[0054]

[0055] Among them, E sis the symbol energy within the symbol interval T, the symbol rate is 2.5Gsps, the symbol interval is 4ns, and a k is the amplitude of the kth sampling point, K is the number of sampling points contained in T, h(t-kT) represents the impulse response of the signal, and f c is the carrier frequency, θ c (t) is the phase noise, and n(t) represents the complex white noise with zero mean and variance N0 / 2. c (t) is modeled as a Wiener process, expressed as:

[0056]

[0057] Where G(x) has a mean of zero and a variance of A Gaussian random variable, δ v The 3dB line width of the carrier is 10kHz. The received signal r(t) is related to the local oscillator (LO) signal. Mixing frequency, f LO and θ LO represents the frequency and phase noise of LO respectively. The signal output by the optical mixer is converted into an analog electrical signal through a balanced detector to generate an in-phase (I) component and a quadrature (Q) component expressed as:

[0058]

[0059] Where Δf = f c -f LO is the residual Doppler frequency deviation, n I (t) and n Q (t) is the noise component, and the input signal of the carrier recovery method is expressed as:

[0060]

[0061] Where θ is the carrier phase, expressed as:

[0062]

[0063] Where 2πΔfkT represents the phase introduced by the residual frequency offset, θ c (kT)-θ LO (kT) represents the v Correlated phase noise.

[0064] Step 2: Laser signal Perform phase detection, phase It is expressed as:

[0065]

[0066] Step 3, Phase of Step 2 and phase detector gain Kd Multiplication is the phase error K d θ in , K under QPSK modulation d =4, the second-order loop filter is 4θ in Perform noise reduction to obtain a smooth signal θ loop_0 ;

[0067] θ loop_0 The z-domain representation Y(z) is:

[0068] Y(z)=H(z)X(z)

[0069] Where X(z) is 4θ in The z-domain representation of ; H(z) is the transfer function of the second-order loop filter, expressed as:

[0070]

[0071] Among them, K1 and K2 are the gain factors of the loop filter, which affect the response speed of the loop and the ability to filter out noise;

[0072] Step 4: The numerically controlled oscillator is based on the θ in step 3. loop_0 Calculate the carrier phase estimate

[0073] The z-domain representation Y1(z) is:

[0074] Y1(z)=H1(z)Y(z)

[0075] Where H1(z) is the transfer function of the digital controlled oscillator, expressed as:

[0076]

[0077] Step 5: Phase of Step 2 Subtract the estimate from step 4 Get the error value θ d , the phase detector gain 4 and the error value θ d Multiply by 4θ to get the phase error d ; Second-order loop filter for 4θ d Perform noise reduction to obtain a smooth signal θ loop ;

[0078] θ loop The z-domain representation Y′(z) is:

[0079] Y′(z)=H(z)X′(z)

[0080] Where X′(z) is 4θ d The z-domain representation of

[0081] Step 6: According to θ in step 4 loop Calculate the carrier phase estimate

[0082] The z-domain representation Y1′(z) is:

[0083] Y1′(z)=H1(z)Y′(z)

[0084] Step 7: The phase detector, loop filter and digitally controlled oscillator form a feedback loop, and repeat steps 5 and 6, that is, use step 6 to output Replace step 5 Perform iterative updates and output the carrier phase estimate at each moment The iteration stops when there is no signal input to the phase detector;

[0085] The transfer function of the feedback loop in step 7 is expressed as:

[0086]

[0087] Among them, K1 and K2 are related to the characteristic frequency ω and damping factor ξ parameters of the feedback loop. When the product of the characteristic frequency and the symbol period T is much less than 1, that is, ω n T<<1, two variables K1, K2 and ω n ,ξ is expressed as:

[0088]

[0089] The symbol period T is determined by the system design, and the empirical value of ξ is The values ​​of K1 and K2 depend on the characteristic frequency ω n ,ω n The empirical conditions are:

[0090]

[0091] Step 8: According to the phase error 4θ of step 5 d , the phase estimation value of the feedforward stage at the current moment is calculated by the block estimation method

[0092]

[0093] Where k represents the signal sampling point number, and N represents the phase estimation window length;

[0094] Step 9: The phase estimate at the current iteration time of step 7 and the feedforward phase estimate at the current moment in step eight Add up to get the carrier phase estimate at the current moment

[0095] Step 10: Calculate the signal carrier phase estimation value in real time by repeating steps 7 to 9.

[0096] In a preferred embodiment of the present invention, based on the calculated carrier phase recovery bit error rate results under different signal-to-noise ratios, a bit error rate curve is drawn as follows: Figure 4 As shown, this embodiment shows better estimation continuity when the bit signal-to-noise ratio is lower than 5.2dB, and is significantly better than the traditional algorithm under QPSK modulation and when the bit signal-to-noise ratio is lower than 4.5dB, and the bit error rate is 1 / 10 of that of the traditional algorithm.

[0097] As shown in Table 1, a low-complexity noise-tolerant laser phase recovery method disclosed in this embodiment can reduce the consumption of multipliers by 60% and adders by 99% compared with traditional algorithms.

[0098] Table 1 Complexity comparison table

[0099]

[0100] In Table 1: N1 represents the first-level frequency offset estimation window length, N2 represents the second-level fourth-power phase estimation window length, N represents the fine phase estimation window length in a low-complexity noise-tolerant laser phase recovery method, and K represents the data length.

[0101] The specific description above further illustrates the purpose, technical solutions and beneficial effects of the invention in detail. It should be understood that the above is only a specific embodiment of the present invention and is not intended to limit the scope of protection of the present invention. Any modifications, equivalent substitutions, improvements, etc. made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A low-complexity noise-tolerant laser phase recovery method, characterized in that: The frequency deviation and phase of the signal carrier are jointly estimated by the phase estimation in the feedback stage and the phase estimation in the feedforward stage, including the following steps: Step 1: Laser signal Perform phase detection, phase It is expressed as: Step 2: Phase of Step 1 and phase detector gain K d Multiplication is the phase error K d θ in , the second-order loop filter is K d θ in Perform noise reduction to obtain a smooth signal θ loop_0 ; θ loop_0 The z-domain representation Y(z) is: Y(z)=H(z)X(z) Where X(z) is K d θ in The z-domain representation of ; H(z) is the transfer function of the second-order loop filter, expressed as: Among them, K1 and K2 are the gain factors of the loop filter, which affect the response speed of the loop and the ability to filter out noise; Step 3: The numerically controlled oscillator is based on the θ in step 2. loop_0 Calculate the carrier phase estimate The z-domain representation Y1(z) is: Y1(z)=H1(z)Y(z) Where H1(z) is the transfer function of the digital controlled oscillator, expressed as: Step 4: Phase of Step 1 Subtract the estimate from step 3 Get the error value θ d , phase detector gain K d and error value θ d Multiplication is the phase error K d θ d ; Second-order loop filter for K d θ d Perform noise reduction to obtain a smooth signal θ loop ; θ loop The z-domain representation Y′(z) is: Y′(z)=H(z)X′(z) Where X′(z) is K d θ d The z-domain representation of Step 5: According to the θ in step 4 loop Calculate the carrier phase estimate The z-domain representation Y1′(z) is: Y1′(z)=H1(z)Y′(z) Step 6: The phase detector, loop filter and digital controlled oscillator form a feedback loop; repeat steps 4 and 5, that is, use step 5 to output Replace step 4 Perform iterative updates and output the carrier phase estimate at each moment The iteration stops when there is no signal input to the phase detector; Step 7: According to the phase error K of step 4 d θ d , the phase estimation value of the feedforward stage at the current moment is calculated by the block estimation method Where k represents the signal sampling point number, and N represents the phase estimation window length; Step 8: The phase estimate at the current iteration time of step 6 and the feedforward phase estimate at the current moment in step 7 Add up to get the carrier phase estimate at the current moment Step 9: Calculate the signal's carrier phase estimation value in real time by repeating steps 6 to 8.

2. A low-complexity noise-tolerant laser phase recovery method as claimed in claim 1, characterized in that: The transfer function of the feedback loop in step 6 is expressed as: Where K d , K1 and K2 and the characteristic frequency ω of the feedback loop n , the damping factor ξ parameter is related. When the product of the characteristic frequency and the symbol period T is much less than 1, that is, ω n T<<1, three variables K d , K1, K2 and ω n ,ξ is expressed as: The symbol period T is determined by the system design, and the empirical value of ξ is The values ​​of K1 and K2 depend on K d and characteristic frequency ω n ,ω n The empirical conditions are:

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