Wired and wireless dual-load high-frequency system based on load impedance compression and parameter design method

Abstract

The invention relates to a wired and wireless dual-load high-frequency system based on load impedance compression and a parameter design method. The invention relates to the technical field of load impedance compression design, provides a load impedance compression mechanism, and adds an impedance conversion network to adjust a branch impedance value. The influence brought by the change of the coupling coefficient is compressed from the angle of the impedance track, the complex interlaced track curve among multiple loads is decoupled, the independence of the load power is recovered, and a theoretical basis is provided for further packaging and expansibility design of a multi-load module.

Description

Technical Field

[0001] The invention relates to the technical field of load impedance compression design, in particular to a wired and wireless dual-load high-frequency system and a parameter design method based on load impedance compression. Background Art

[0002] After entering the Industrial Scientific Medical Band (ISM), namely 6.78MHz, 13.56MHz and 27.12MHz, the rectifier input voltage and current no longer present zero phase angle characteristics, such as Figure 1 As shown. Due to the increase in the proportion of parasitic capacitance effect, the input current exceeds the input voltage, and the rectifier exhibits capacitive impedance characteristics, and produces nonlinear changes with power changes. Take the full-bridge rectifier as an example:

[0003]

[0004]

[0005] Among them, Q j is the junction charge stored in the diode junction capacitance. rec_FBR and X rec_FBR R and V represent the real and imaginary input impedance of the full-bridge rectifier respectively. o Corresponding to the rectifier output load and voltage respectively. In low-frequency systems, the load-independent characteristics of LCC / S (or bilateral LCC) obtained by the classical fundamental wave analysis method will change in high-frequency systems. Therefore, the present invention will perform trajectory analysis from a more accurate impedance compression perspective.

[0006] Figure 2 The impedance trajectory changes of the single wired load system and the wired / wireless dual load system under traditional impedance compression design are demonstrated. Figure 3 Shows Figure 2 The output performance of the Class-E power amplifier (PA) used in the analysis. The yellow area is the high efficiency impedance range of the PA, with the PA output impedance Z pa Performing a load-pull sweep, as shown by the red line, the PA efficiency continues to increase as it moves toward the center. The purple curve is the PA power output contour, as shown by the green arrow, the power output gradually decreases. Then, for the single wire load case, the load resistance R L After the rectifier, the capacitive input impedance Z rec ,like Figure 2 (a). Obviously, at this time Z rec It is not within the efficient range of PA, so an impedance matching network (IMN) is needed torec Convert to Z pa On the one hand, it can meet the needs of PA to work efficiently, and on the other hand, it can adjust the power level required by the system. However, when the number of loads begins to increase, such as Figure 2 As shown in (b), the impedance trajectory begins to become complicated. Similarly, the load R L1 , R L2 (purple triangle) becomes capacitive after passing through the rectifier.

[0007] Figure 2 High frequency system impedance trajectory changes; Figure 2 -(a) Single-point impedance compression design of wired load system; Figure 2 -(b) Impedance compression design of high-frequency wired / wireless dual-load system; Figure 2 -(c) Impedance compression design of high-frequency wired / wireless dual-load system based on PA efficiency optimization;

[0008] The wireless branch needs to be able to move freely, so Z rec_1 (Cyan circle) The reflected impedance Z under offset is obtained through coil coupling ref_1 (red triangle). It can be seen that Z ref_1 The impedance variation range is relatively wide, which will lead to the synthetic impedance Z IMN (Green circle) has a larger range of variation. Under the same impedance compression design, Z pa It may not be possible to maintain PA high efficiency. Of course, some dynamic IMN can be used to re-optimize the impedance trajectory to ensure Z pa Falling into the high efficiency area, such as Figure 2 (c) shows. However, this will increase the complexity of the system. More importantly, the impedance trajectory of the system is still relatively chaotic, and the power distribution of each load cannot be accurately obtained. There is power coupling between the loads, and the independent characteristics of each load are not achieved. In this way, the requirements of load scalability cannot be further met. Summary of the invention

[0009] Miniaturization is one of the main trends in the development of power supplies, and high frequency helps to improve the flexibility of the magnetic field space of wireless power transmission technology, making it more suitable for multi-load application scenarios. However, high frequency will also increase the sensitivity of the system and cause power coupling problems in multi-load designs. To this end, the present invention proposes a load impedance compression mechanism for wired / wireless dual-load applications, and adds an impedance conversion network to adjust the branch impedance value. From the perspective of impedance trajectory, the impact of changes in coupling coefficients is compressed, the complex and intertwined trajectory curves between multiple loads are decoupled, and the independence of load power is restored, providing a theoretical basis for further packaging and scalability design of multi-load modules.

[0010] The present invention provides a wired and wireless dual-load high-frequency system and parameter design method based on load impedance compression. The present invention provides the following technical solutions:

[0011] A wired and wireless dual-load high-frequency system based on load impedance compression, the system comprising: a constant AC voltage source, a capacitor C tx 、Inductance L tx 、Inductance L rx , capacitor C rx , diode D1, diode D2, capacitor C r1 , capacitor C r2 、Inductance L 01 、Inductance L 02 , capacitor C0, resistor R0, capacitor C b1 、Inductance L b1 , capacitor C b2 、Inductance L b2 、Diode D F1 、Diode D F2 , capacitor C rF1 , capacitor C rF2 、Inductance L 0F1 、Inductance L 0F2 , capacitor C 0F and resistor R 0F ;

[0012] One end of the constant AC voltage source is connected to a capacitor C tx and capacitor C b1 One end of the capacitor C tx The other end of the inductor L tx One end of the inductor L tx and inductor L rx Mutual inductance; inductance L rx One end of the capacitor C rx One end of the capacitor C rx The other end is connected to diode D1 and capacitor C r1 and inductor L 01 One end of the inductor L 01 The other end of the inductor L 02 , capacitor C0 and one end of resistor R0, the other end of capacitor C0 is connected to the other end of resistor R0 and grounded; diode D1 and capacitor C r1 The other end is connected to diode D2 and capacitor C r2 One end of the diode D2 and capacitor C r2 The other end of the inductor L rx and inductor L 02 The other end of

[0013] The other end of the constant AC voltage source is connected to the inductor L tx and inductor Lb1 The other end of the capacitor C b1 The other end of the inductor L b1 and capacitor C b2 One end of the capacitor C b2 The other end of the inductor L b2 One end of the inductor L b2 The other end of the diode D F1 , capacitor C rF1 and inductor L 0F1 One end of the inductor L 0F1 The other end of the inductor L 0F2 , capacitor C 0F and resistor R 0F One end of the capacitor C 0F The other end of the resistor R 0F The other end of the diode D F1 and capacitor C rF1 The other end is connected to the diode D F2 and capacitor C rF2 One end of the diode D F2 and capacitor C rF2 The other end of the inductor L b1 and inductor L 0F2 the other end.

[0014] Preferably, the load branch reflected impedance Z ref_1 and Z ref_2 Connect in parallel to reduce the synthetic impedance Z IMN , add an impedance conversion network in the wired branch to adjust the reflected impedance Z ref_2 , to achieve fixed power transmission of wired branches.

[0015] A synthetic impedance design method for a wired and wireless dual-load high-frequency system based on load impedance compression, the method specifically comprising:

[0016] For wireless branches, compensate for the reactive components on the secondary side to improve coil efficiency;

[0017] By using the load-pull function, the efficient impedance range of the coil under different coupling coefficients k is depicted with the rectifier input impedance as the reference;

[0018] Select the minimum coupling coefficient for full compensation and design the rectifier parallel capacitor to achieve the maximum efficiency of the coil under this coupling coefficient;

[0019] The impedance trajectory is obtained by analyzing and calculating the impedance values ​​under different coupling coefficients;

[0020] Compare the impedance trajectory to see whether the impedance values ​​under different coupling coefficients are within the high efficiency range of the coil, and adjust the value of the rectifier parallel capacitor;

[0021] When the stable voltage is generated, the wired load power will remain constant, keeping the impedance value unchanged; when ignoring the internal resistance of the line, determine the synthetic impedance Z IMN .

[0022] Preferably, for the wireless branch, the reactive component on the secondary side is compensated to improve the coil efficiency, that is:

[0023]

[0024] When the output power of the wireless branch changes, its corresponding X rec_1 Traditional approach to fully compensated design:

[0025]

[0026] At this time, the reflected impedance Z ref_1 for:

[0027]

[0028] The coil efficiency is expressed as follows:

[0029]

[0030] In the traditional method, the coil efficiency is simplified to:

[0031]

[0032] The peak efficiency is derived as:

[0033]

[0034] The real impedance of the rectifier corresponding to the peak efficiency of the coil is:

[0035]

[0036] Using the load-pull function, the rectifier input impedance Z rec_1 As a benchmark, the efficient impedance range of the coil under different coupling coefficients k is depicted.

[0037] Preferably, for the wired load branch, the reflected impedance Z is reduced by ICN. ref_2 , according to Kirchhoff's voltage / current law:

[0038]

[0039] Among them, u ab is the sinusoidal voltage source generated by PA through IMN, and further derivation yields:

[0040]

[0041] Preferably, the resonant element satisfies the following equation:

[0042]

[0043] The simplified result is:

[0044]

[0045] When u ab After the stable voltage is generated, the wired load power P oF will remain constant, keeping Z rec_2 of unchanged.

[0046] Preferably, the reflected impedance eliminates the reactive component, moves left and right on the zero imaginary impedance horizontal axis, and has parameter flexibility to reduce the synthetic impedance Z IMN The range of change requires Z ref_2 Approaching the left on the horizontal axis.

[0047] Preferably, ignoring the internal resistance of the line, Z IMN equal:

[0048] .

[0050] A computer-readable storage medium stores a computer program, which is executed by a processor to implement a synthetic impedance design method for a wired and wireless dual-load high-frequency system based on load impedance compression.

[0051] A computer device comprises a memory and a processor, wherein the memory stores a computer program, and when the processor executes the computer program, a synthetic impedance design method for a wired and wireless dual-load high-frequency system based on load impedance compression is implemented.

[0052] The present invention has the following beneficial effects:

[0053] Compared with the prior art, the present invention has the following advantages:

[0054] Based on the analysis of the limitations of traditional multi-load impedance trajectory design, the present invention makes the multi-load trajectory design clearer from the perspective of trajectory optimization, thereby facilitating the realization of load scalability. The present invention does not require additional costs, is simple to implement, and has broad application prospects in the high-frequency field. BRIEF DESCRIPTION OF THE DRAWINGS

[0055] In order to more clearly illustrate the specific implementation methods of the present invention or the technical solutions in the prior art, the drawings required for use in the specific implementation methods or the description of the prior art will be briefly introduced below. Obviously, the drawings described below are some implementation methods of the present invention. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying creative work.

[0056] Figure 1 This is a performance comparison chart of 100kHz and 6.78MHz rectifiers;

[0057] Figure 2 It is the impedance trajectory change diagram of the high-frequency system;

[0058] Figure 3 Based on Z pa Class E PA output performance graph with impedance variation;

[0059] Figure 4 This is the structural diagram of the intermediate constant voltage AC bus;

[0060] Figure 5 This is a conceptual diagram of load impedance compression;

[0061] Figure 6 This is the structure diagram of the subsequent circuit;

[0062] Figure 7 Based on Z rec_1 Coil efficiency analysis diagram of impedance trajectory;

[0063] Figure 8 It is the circuit equivalent analysis diagram;

[0064] Fig. 9 For different C r And the wired branch impedance change diagram under ICN value;

[0065] Fig.10 This is the impedance trajectory change diagram of the wireless / wired dual-load high-frequency system;

[0066] Fig.11 PA outputs constant voltage source u through IMN ab Schematic diagram;

[0067] Fig.12 Z pa Impedance trajectory design diagram;

[0068] Fig.13 This is a 6.78MHz wired / wireless dual-load high-frequency system diagram;

[0069] Fig.14 This is the experimental prototype diagram;

[0070] Fig.15This is a waveform diagram showing the power output of 33.87W when the mutual inductance is 1.796uH;

[0071] Fig.16 This is a waveform diagram showing the power output of 36.28W when the mutual inductance is 1.48uH;

[0072] Fig.17 This is a waveform diagram showing that when the mutual inductance is 1.157uH, the power output is 40.93W. DETAILED DESCRIPTION

[0073] The technical solution of the present invention will be described clearly and completely below in conjunction with the accompanying drawings. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0074] In the description of the present invention, it should be noted that the terms "center", "upper", "lower", "left", "right", "vertical", "horizontal", "inner", "outer", etc., indicating the orientation or positional relationship, are based on the orientation or positional relationship shown in the drawings, and are only for the convenience of describing the present invention and simplifying the description, rather than indicating or implying that the device or element referred to must have a specific orientation, be constructed and operated in a specific orientation, and therefore cannot be understood as limiting the present invention. In addition, the terms "first", "second", and "third" are used for descriptive purposes only, and cannot be understood as indicating or implying relative importance.

[0075] In the description of the present invention, it should be noted that, unless otherwise clearly specified and limited, the terms "installed", "connected", and "connected" should be understood in a broad sense, for example, it can be a fixed connection, a detachable connection, or an integral connection; it can be a mechanical connection or an electrical connection; it can be a direct connection, or it can be indirectly connected through an intermediate medium, or it can be the internal communication of two components. For ordinary technicians in this field, the specific meanings of the above terms in the present invention can be understood according to specific circumstances.

[0076] In addition, the technical features involved in the different embodiments of the present invention described below can be combined with each other as long as they do not conflict with each other.

[0077] The present invention is described in detail below in conjunction with specific embodiments. Specific embodiment one:

[0079] according to Figure 1-Figure 17 As shown, the specific optimization technical solution adopted by the present invention to solve the above technical problems is: the present invention relates to a wired and wireless dual-load high-frequency system and parameter design method based on load impedance compression.

[0080] The present invention provides a wired and wireless dual-load high-frequency system based on load impedance compression, the system comprising: a constant AC voltage source, a capacitor C tx 、Inductance L tx 、Inductance L rx , capacitor C rx , diode D1, diode D2, capacitor C r1 , capacitor C r2 、Inductance L 01 、Inductance L 02 , capacitor C0, resistor R0, capacitor C b1 、Inductance L b1 , capacitor C b2 、Inductance L b2 、Diode D F1 、Diode D F2 , capacitor C rF1 , capacitor C rF2 、Inductance L 0F1 、Inductance L 0F2 , capacitor C 0F and resistor R 0F ;

[0081] One end of the constant AC voltage source is connected to a capacitor C tx and capacitor C b1 One end of the capacitor C tx The other end of the inductor L tx One end of the inductor L tx and inductor L rx Mutual inductance; inductance L rx One end of the capacitor C rx One end of the capacitor C rx The other end is connected to diode D1 and capacitor C r1 and inductor L 01 One end of the inductor L 01 The other end of the inductor L 02 , capacitor C0 and one end of resistor R0, the other end of capacitor C0 is connected to the other end of resistor R0 and grounded; diode D1 and capacitor C r1 The other end is connected to diode D2 and capacitor C r2 One end of the diode D2 and capacitor C r2 The other end of the inductor L rx and inductor L 02 The other end of

[0082] The other end of the constant AC voltage source is connected to the inductor L tx and inductor L b1 The other end of the capacitor C b1 The other end of the inductor Lb1 and capacitor C b2 One end of the capacitor C b2 The other end of the inductor L b2 One end of the inductor L b2 The other end of the diode D F1 , capacitor C rF1 and inductor L 0F1 One end of the inductor L 0F1 The other end of the inductor L 0F2 , capacitor C 0F and resistor R 0F One end of the capacitor C 0F The other end of the resistor R 0F The other end of the diode D F1 and capacitor C rF1 The other end is connected to the diode D F2 and capacitor C rF2 One end of the diode D F2 and capacitor C rF2 The other end of the inductor L b1 and inductor L 0F2 the other end. Specific embodiment 2:

[0084] The difference between the second embodiment of the present invention and the first embodiment is that:

[0085] By setting the load branch reflected impedance Z ref_1 and Z ref_2 Connect in parallel to reduce the synthetic impedance Z IMN , add an impedance conversion network in the wired branch to adjust the reflected impedance Z ref_2 , to achieve fixed power transmission of wired branches. Specific embodiment three:

[0087] The difference between the third embodiment of the present invention and the second embodiment is that:

[0088] For wireless branches, compensate for the reactive components on the secondary side to improve coil efficiency;

[0089] By using the load-pull function, the efficient impedance range of the coil under different coupling coefficients k is depicted with the rectifier input impedance as the reference;

[0090] Select the minimum coupling coefficient for full compensation and design the rectifier parallel capacitor to achieve the maximum efficiency of the coil under this coupling coefficient;

[0091] The impedance trajectory is obtained by analyzing and calculating the impedance values ​​under different coupling coefficients;

[0092] Compare the impedance trajectory to see whether the impedance values ​​under different coupling coefficients are within the high efficiency range of the coil, and adjust the value of the rectifier parallel capacitor;

[0093] When the stable voltage is generated, the wired load power will remain constant, keeping the impedance value unchanged; when ignoring the internal resistance of the line, determine the synthetic impedance Z IMN . Specific embodiment four:

[0095] The difference between the fourth embodiment of the present invention and the third embodiment is that:

[0096] For the wireless branch, the reactive component on the secondary side is compensated to improve the coil efficiency, that is:

[0097]

[0098] When the output power of the wireless branch changes, its corresponding X rec_1 Traditional approach to fully compensated design:

[0099]

[0100] At this time, the reflected impedance Z ref_1 for:

[0101]

[0102] The coil efficiency is expressed as follows:

[0103]

[0104] In the traditional method, the coil efficiency is simplified to:

[0105]

[0106] The peak efficiency is derived as:

[0107]

[0108] The real impedance of the rectifier corresponding to the peak efficiency of the coil is:

[0109]

[0110] Using the load-pull function, the rectifier input impedance Z rec_1 As a benchmark, the efficient impedance range of the coil under different coupling coefficients k is depicted. Specific embodiment five:

[0112] The difference between the fifth embodiment of the present invention and the fourth embodiment is that:

[0113] For wired load branches, ICN is used to reduce the reflected impedance Zref_2 , according to Kirchhoff's voltage / current law:

[0114]

[0115] Among them, u ab is the sinusoidal voltage source generated by PA through IMN, and further derivation yields:

[0116] Specific embodiment six:

[0118] The difference between the sixth embodiment of the present invention and the fifth embodiment is that:

[0119] Let the resonant element satisfy the following equation:

[0120]

[0121] The simplified result is:

[0122]

[0123] When u ab After the stable voltage is generated, the wired load power P oF will remain constant, keeping Z rec_2 of unchanged. Specific embodiment seven:

[0125] The difference between the seventh embodiment of the present invention and the sixth embodiment is that:

[0126] The reflected impedance eliminates the reactive component and moves left and right on the zero imaginary impedance horizontal axis, with parameter flexibility to reduce the synthetic impedance Z IMN The range of change requires Z ref_2 Approaching the left on the horizontal axis. Specific embodiment eight:

[0128] The difference between the eighth embodiment of the present invention and the seventh embodiment is that:

[0129] When the internal resistance of the line is ignored, Z IMN equal:

[0130] Specific embodiment nine:

[0132] The difference between the ninth embodiment of the present invention and the eighth embodiment is that:

[0133] The present invention provides a computer-readable storage medium on which a computer program is stored. The program is executed by a processor to implement a synthetic impedance design method for a wired and wireless dual-load high-frequency system based on load impedance compression. Specific embodiment ten:

[0135] The difference between the tenth embodiment of the present invention and the ninth embodiment is that:

[0136] The present invention provides a computer device, comprising a memory and a processor, wherein the memory stores a computer program, and when the processor executes the computer program, a synthetic impedance design method for a wired and wireless dual-load high-frequency system based on load impedance compression is implemented. Specific embodiment eleven:

[0138] After entering the Industrial Scientific Medical Band (ISM), namely 6.78MHz, 13.56MHz and 27.12MHz, the rectifier input voltage and current no longer present zero phase angle characteristics, such as Figure 1 As shown. Due to the increase in the proportion of parasitic capacitance effect, the input current exceeds the input voltage, and the rectifier exhibits capacitive impedance characteristics, and produces nonlinear changes with power changes. Take the full-bridge rectifier as an example:

[0139]

[0140] Among them, Q j is the junction charge stored in the diode junction capacitance. rec_FBR and X rec_FBR R and V represent the real and imaginary input impedance of the full-bridge rectifier respectively. o Corresponding to the rectifier output load and voltage respectively. In low-frequency systems, the load-independent characteristics of LCC / S (or bilateral LCC) obtained by the classical fundamental wave analysis method will change in high-frequency systems. Therefore, the present invention will perform trajectory analysis from a more accurate impedance compression perspective.

[0141] Figure 2 The impedance trajectory changes of the single wired load system and the wired / wireless dual load system under traditional impedance compression design are demonstrated. Figure 3 Shows Figure 2 The output performance of the Class-E power amplifier (PA) used in the analysis. The yellow area is the high efficiency impedance range of the PA, with the PA output impedance Z pa Performing a load-pull sweep, as shown by the red line, the PA efficiency continues to increase as it moves toward the center. The purple curve is the PA power output contour, as shown by the green arrow, the power output gradually decreases. Then, for the single wire load case, the load resistance R L After the rectifier, the capacitive input impedance Z rec ,like Figure 2 (a). Obviously, at this time Z recIt is not within the efficient range of PA, so an impedance matching network (IMN) is needed to rec Convert to Z pa On the one hand, it can meet the needs of PA to work efficiently, and on the other hand, it can adjust the power level required by the system. However, when the number of loads begins to increase, such as Figure 2 As shown in (b), the impedance trajectory begins to become complicated. Similarly, the load R L1 , R L2 (purple triangle) becomes capacitive after passing through the rectifier.

[0142] The wireless branch needs to be able to move freely, so Z rec_1 (Cyan circle) The reflected impedance Z under offset is obtained through coil coupling ref_1 (red triangle). It can be seen that Z ref_1 The impedance variation range is relatively wide, which will lead to the synthetic impedance Z IMN (Green circle) has a larger range of variation. Under the same impedance compression design, Z pa It may not be possible to maintain PA high efficiency. Of course, some dynamic IMN can be used to re-optimize the impedance trajectory to ensure Z pa Falling into the high efficiency area, such as Figure 2 (c) shows. However, this will increase the complexity of the system. More importantly, the impedance trajectory of the system is still relatively chaotic, and the power distribution of each load cannot be accurately obtained. There is power coupling between the loads, and the independent characteristics of each load are not achieved. In this way, the requirements of load scalability cannot be further met.

[0143] Miniaturization is one of the main trends in the development of power supplies, and high frequency helps to improve the flexibility of the magnetic field space of wireless power transmission technology, making it more suitable for multi-load application scenarios. However, high frequency will also increase the sensitivity of the system and cause power coupling problems in multi-load designs. To this end, the present invention proposes a load impedance compression mechanism for wired / wireless dual-load applications, and adds an impedance conversion network to adjust the branch impedance value. From the perspective of impedance trajectory, the impact of changes in coupling coefficients is compressed, the complex and intertwined trajectory curves between multiple loads are decoupled, and the independence of load power is restored, providing a theoretical basis for further packaging and scalability design of multi-load modules.

[0144] Aiming at the problem of chaotic impedance trajectory of multiple loads under traditional impedance compression design, the present invention proposes a load impedance compression mechanism. The basic idea is to build an intermediate constant voltage AC bus and divide the complex impedance trajectory design into two parts to solve it, such as Figure 4 shown.

[0145] In this way, compared with the power that is difficult to determine when the PA directly outputs, the power obtained by the load through the secondary bus is actually easier to estimate. Therefore, in order to achieve the constant voltage effect of the PA output, it is necessary to greatly compress the synthetic impedance Z IMN , which is beneficial to Z PA After all, the impedance compression capability of static IMN is limited. The design idea of ​​load impedance compression came into being, such as Figure 5 As shown. By setting the load branch reflected impedance Z ref_1 and Z ref_2 Connect in parallel to reduce the synthetic impedance Z IMN The reflected impedance of the general wireless load branch is affected by the change of the coupling coefficient, resulting in Z ref_1 Therefore, adding an impedance conversion network (ICN) in the wired branch can not only adjust the reflected impedance Z ref_2 , while achieving fixed power transmission of the wired branch. Figure 6 The back-end circuit structure adopted by the present invention uses a full-wave rectifier, which is relatively common and can provide a good real impedance.

[0146] Next is the formula derivation, the rectifier input impedance is equal to:

[0147]

[0148] Among them, R rec and X rec Respectively represent the real and imaginary parts of the full-wave rectifier input impedance:

[0149]

[0150]

[0151]

[0152]

[0153] Here, D is the duty cycle of the diode, φ rec is the initial phase. The above formula can also be regarded as a nonlinear function of load, power and parallel capacitance.

[0154]

[0155] For the wireless branch, it is necessary to compensate the secondary side reactive component as much as possible to improve the coil efficiency, that is:

[0156]

[0157] However, as described in formula (8), when the output power of the wireless branch changes, its corresponding X rec_1 Therefore, the above formula cannot guarantee that the secondary side is fully compensated under all coupling coefficients. The primary side compensation has no effect, and the traditional method can be used for full compensation design.

[0158]

[0159] At this time, the reflected impedance Z ref_1 for:

[0160]

[0161] Then the coil efficiency can be expressed as:

[0162]

[0163] In traditional methods, only the efficiency analysis of the coil in the full resonance state is usually considered. At this time, the coil efficiency can be simplified to:

[0164]

[0165] Further, the peak efficiency is derived:

[0166]

[0167] At this time, the real impedance of the rectifier corresponding to the peak efficiency of the coil is:

[0168]

[0169] Formula (15) is usually used as a reference for the design of wireless branch rectifier parameters. At the maximum power point, let the real part of the rectifier input impedance equal to However, in the present invention, since the wireless branch cannot be fully compensated in all offset conditions, a new design method needs to be proposed from the perspective of impedance trajectory. Similarly, the load pull function is used. Here, the rectifier input impedance Z rec_1 As a benchmark, the efficient impedance range of the coil under different coupling coefficients (k) is depicted, such as Figure 7 shown.

[0170] From equations (12) to (14), it can be seen that the increase of mutual inductance (M) helps to improve the efficiency of the coil. Therefore, the minimum coupling coefficient is first selected for complete compensation, and the rectifier parallel capacitor is designed to achieve the maximum efficiency of the coil under this coupling coefficient. Then, Z under different coupling coefficients is analyzed and calculated. rec_1 Value, can be obtained Figure 7 Finally, compare Z rec_1 Impedance trace, observe Z under different coupling coefficientsrec_1 Whether they are all within the high efficiency range of the coil, so as to further adjust the rectifier parallel capacitor C r Therefore, for the wireless branch, C r The choice is relatively flexible.

[0171] For the wired load branch, the design idea is to reduce the reflected impedance Z through ICN. ref_2 . First, according to Kirchhoff's voltage / current law:

[0172]

[0173] In the formula, u ab is the sinusoidal voltage source generated by PA through IMN. Further deduction yields:

[0174]

[0175] At this time, let the resonant element satisfy the following equation:

[0176]

[0177] The simplified result is:

[0178]

[0179] When u ab After the stable voltage is generated, the wired load power P oF will remain constant, keeping Z rec_2 is unchanged [see formula (8)]. On this basis, formula (18) and formula (19) are guaranteed to be valid. This is actually a state interlocking process. Fig. 9 Different C r The input impedance Z of the wired branch rectifier under the resonance value of ICN rec_2 and the reflected impedance Z ref_2 It can be seen that the reflected impedance eliminates the reactive component and can move left and right on the zero imaginary impedance horizontal axis, which has strong parameter flexibility. IMN The range of change requires Z ref_2 On the horizontal axis, it approaches to the left. But on the other hand, Z ref_2 The value should not be too small, otherwise it will increase the branch current I t_2 , resulting in additional losses. When the internal resistance of the line is ignored, Z IMN equal:

[0180]

[0181] Fig.10The final impedance trajectory change and its corresponding impedance parameters of the wireless / wired dual-load high-frequency system are given. It can be seen that the wide range of reflected impedance Z caused by the coupling change ref_1 At the appropriate Z ref_2 Under the impedance compression, Z IMN The range of variation is greatly reduced. IMN The real part change is only 7.25Ω. The three impedance trajectories are basically maintained near the zero imaginary impedance horizontal axis, reducing the reactive loss of the system.

[0182] In addition, Z IMN The extreme compression will greatly reduce the difficulty of designing the impedance compression network, making it easier to pa Impedance trajectory planning. IMN is placed after the classic Class E amplifier to play an impedance compression role. Fig.13 , further derive the PA output impedance Z pa :

[0183]

[0184] At this time, it is necessary to adjust the three parameter degrees of freedom of IMN (X T1 -X T3 ), so that Z pa When the impedance trajectory falls into the high-efficiency range of the PA, its power transmission direction satisfies the relationship between constant voltage output and load, that is, as the load increases, the power obtained should decrease to keep the voltage constant.

[0185] Under such constraints, Fig.12 Given Z pa The trajectory of the real impedance changes little, while the imaginary part satisfies:

[0186]

[0187] Table 1 Circuit parameters of dual-load WPT system

[0188]

[0189] In order to verify the correctness of the proposed theory, an experimental prototype was built. Fig.13 The complete circuit of the proposed system is shown in Table 1. The parameter values ​​of each component are given in Table 1. Fig.14 For the physical prototype, two electronic loads are used to simulate the R o and R oF. For high-frequency systems, actual circuit production and debugging should be more cautious than low-frequency kHz systems. For example, when drawing a hardware circuit board, great care should be taken to avoid two different networks being directly opposite to each other to avoid generating additional parasitic capacitance. After all, the capacitance value commonly used in high-frequency systems is already in the pF level. At the same time, since most high-frequency systems use hollow coils to make resonant inductors, the inductors should be placed along the orthogonal magnetic field direction and leave enough distance to avoid unnecessary coupling between inductor components. For parameter determination, it is necessary to combine a high-precision, high-bandwidth impedance network analyzer to perform multiple measurements and take the average value to reduce device errors. Finally, it is necessary to re-correct each parameter according to different circuit boards, actual welding components and circuit simulation to obtain results close to theory.

[0190] Then from the experimental results, when the mutual inductance changes from 1.157μH to 1.796μH, the wired branch load always outputs 30W and the voltage remains at around 20V. Fig.15 (c)- Fig.17 (c) is shown. Although the rectified waveform has a certain degree of asymmetry, it is still within an acceptable range. From a certain perspective, this is also one of the unique working conditions of this wired / wireless dual-load high-frequency system. As for the wireless load branch, due to the secondary filtering effect of the coil matching, the rectified waveform of this part is not affected by harmonics and maintains good symmetry. As the mutual inductance decreases, its power output capacity gradually increases. Fig.15 Corresponding to the total system output of 33.87W, the total system efficiency is 85.8%. Fig.16 Corresponding to the total system output of 36.28W, the total system efficiency is 88.5%. Fig.17 The total system output is 40.93W, and the total system efficiency is 87.3%. This ideal efficiency is mainly due to the load compression principle, which reduces the synthetic impedance Z. IMN The numerical variation range is small, thus achieving high efficiency operation of the power amplifier.

[0191] like Fig.15 (a) to Fig.17 As shown in (a), it can be seen that under different coil coupling conditions, the PA maintains a good soft switching state. Before the driving waveform reaches the threshold voltage of the GaN switch, the drain-source voltage of the PA has almost dropped to 0 voltage. In addition, by observing the primary and secondary currents of the coil, it can be found that it exhibits a good sinusoidal characteristic, which proves that the sinusoidal nature of the intermediate bus voltage is good. The secondary current leads the primary current by nearly 90 degrees, indicating that the compensation network plays a corresponding role.

[0192] The above is only a preferred implementation of a wired and wireless dual-load high-frequency system and parameter design method based on load impedance compression. The protection scope of a wired and wireless dual-load high-frequency system and parameter design method based on load impedance compression is not limited to the above embodiments. All technical solutions under this idea belong to the protection scope of the present invention. It should be pointed out that for those skilled in the art, several improvements and changes without departing from the principle of the present invention should also be regarded as the protection scope of the present invention.

Claims

1. A wired and wireless dual-load high-frequency system based on load impedance compression, characterized by: The system includes: a constant AC voltage source, a capacitor C tx 、Inductance L tx 、Inductance L rx , capacitor C rx , diode D1, diode D2, capacitor C r1 , capacitor C r2 、Inductance L 01 、Inductance L 02 , capacitor C0, resistor R0, capacitor C b1 、Inductance L b1 , capacitor C b2 、Inductance L b2 、Diode D F1 、Diode D F2 , capacitor C rF1 , capacitor C rF2 、Inductance L 0F1 、Inductance L 0F2 , capacitor C 0F and resistor R 0F ; One end of the constant AC voltage source is connected to a capacitor C tx and capacitor C b1 One end of the capacitor C tx The other end of the inductor L tx One end of the inductor L tx and inductor L rx Mutual inductance; inductance L rx One end of the capacitor C rx One end of the capacitor C rx The other end is connected to diode D1 and capacitor C r1 and inductor L 01 One end of the inductor L 01 The other end of the inductor L 02 , capacitor C0 and one end of resistor R0, the other end of capacitor C0 is connected to the other end of resistor R0 and grounded; diode D1 and capacitor C r1 The other end is connected to diode D2 and capacitor C r2 One end of the diode D2 and capacitor C r2 The other end of the inductor L rx and inductor L 02 The other end of The other end of the constant AC voltage source is connected to the inductor L tx and inductor L b1 The other end of the capacitor C b1 The other end of the inductor L b1 and capacitor C b2 One end of the capacitor C b2 The other end of the inductor L b2 One end of the inductor L b2 The other end of the diode D F1 , capacitor C rF1 and inductor L 0F1 One end of the inductor L 0F1 The other end of the inductor L 0F2 , capacitor C 0F and resistor R 0F One end of the capacitor C 0F The other end of the resistor R 0F The other end of the diode D F1 and capacitor C rF1 The other end is connected to the diode D F2 and capacitor C rF2 One end of the diode D F2 and capacitor C rF2 The other end of the inductor L b1 and inductor L 0F2 the other end.

2. The system according to claim 1, characterized in that: By setting the load branch reflected impedance Z ref_1 and Z ref_2 Connect in parallel to reduce the synthetic impedance Z IMN , add an impedance conversion network in the wired branch to adjust the reflected impedance Z ref_2 , to achieve fixed power transmission of wired branches.

3. A synthetic impedance design method for a wired and wireless dual-load high-frequency system based on load impedance compression, characterized in that: For wireless branches, compensate for the reactive components on the secondary side to improve coil efficiency; By using the load-pull function, the efficient impedance range of the coil under different coupling coefficients k is depicted with the rectifier input impedance as the reference; Select the minimum coupling coefficient for full compensation and design the rectifier parallel capacitor to achieve the maximum efficiency of the coil under this coupling coefficient; The impedance trajectory is obtained by analyzing and calculating the impedance values ​​under different coupling coefficients; Compare the impedance trajectory to see whether the impedance values ​​under different coupling coefficients are within the high efficiency range of the coil, and adjust the value of the rectifier parallel capacitor; When the stable voltage is generated, the wired load power will remain constant, keeping the impedance value unchanged; when ignoring the internal resistance of the line, determine the synthetic impedance Z IMN .

4. The method according to claim 3, characterized in that: For the wireless branch, the reactive component on the secondary side is compensated to improve the coil efficiency, that is: When the output power of the wireless branch changes, its corresponding X rec_1 Traditional approach to fully compensated design: At this time, the reflected impedance Z ref_1 for: The coil efficiency is expressed as follows: In the traditional method, the coil efficiency is simplified to: The peak efficiency is derived as: The real impedance of the rectifier corresponding to the peak efficiency of the coil is: Using the load-pull function, the rectifier input impedance Z rec_1 As a benchmark, the efficient impedance range of the coil under different coupling coefficients k is depicted.

5. The method according to claim 4, characterized in that: For wired load branches, ICN is used to reduce the reflected impedance Z ref_2 , according to Kirchhoff's voltage / current law: Among them, u ab is the sinusoidal voltage source generated by PA through IMN, and further derivation yields:

6. The method according to claim 5, characterized in that: Let the resonant element satisfy the following equation: The simplified result is: When u ab After the stable voltage is generated, the wired load power P oF will remain constant, keeping Z rec_2 of unchanged.

7. The method according to claim 6, characterized in that: The reflected impedance eliminates the reactive component and moves left and right on the zero imaginary impedance horizontal axis, with parameter flexibility to reduce the synthetic impedance Z IMN The range of change requires Z ref_2 Approaching the left on the horizontal axis.

8. The method according to claim 7, characterized in that: When the internal resistance of the line is ignored, Z IMN equal:

9. A computer-readable storage medium having a computer program stored thereon, characterized in that: The program is executed by a processor to implement the method according to claims 3-8.

10. A computer device comprising a memory and a processor, wherein the memory stores a computer program, wherein: When the processor executes the computer program, the method of claims 3-8 is implemented.

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