Single sideband DFT-S-OFDM

Abstract

The invention relates to a signal transmission device, which is used for precoding based on discrete Fourier transform (DFT), and transmitting a modulation symbol by using orthogonal frequency division multiplexing (OFDM), and in particular relates to a signal transmission device, which is used for transmitting a modulation symbol by using orthogonal frequency division multiplexing (OFDM) based on discrete Fourier transform (DFT) precoding. A signal transmission apparatus for generating a DFT extended OFDM (DFT spread OFDM, DFT-s-OFDM) signal by receiving an input (x [m]) comprising M modulation symbols, m = 0, 1,..., M-1; the signal transmission device is configured to receive an input (x [m]), and phase-shift the input (x [m]), thereby generating a phase-shifted input # imgabs0 #, for precoding the phase-shifted input using DFT, thereby generating M Fourier coefficients (X [k]) and sorting the Fourier coefficients and selecting M / 2 Fourier coefficients. And the signal transmission device is used for generating the DFT-s-OFDM signal based on the M / 2 selected Fourier coefficients. The signal transmission apparatus improves a block error rate and a peak-to-average power ratio for modulation of low spectral efficiency.

Description

Technical Field

[0001] The present disclosure generally relates to the field of communication systems, and more particularly, to a signal transmission device and a method for the signal transmission device. Background Art

[0002] In general, the third-generation partnership project (3GPP) long term evolution (LTE) and new radio (NR) systems support quadrature phase shift keying (QPSK) modulation technology. QPSK is used for uplink and downlink and is applicable to all control channels and data channels. Due to its low spectral efficiency and high reliability, QPSK is preferred in communications with medium payloads (for example, control messages or small transport blocks). LTE and NR also support binary phase shift keying (BPSK) modulation technology for uplink control channels. In addition, NR also supports π / 2-BPSK modulation for uplink control channels and uplink data channels. BPSK and π / 2-BPSK modulation technologies carry one bit per symbol and are therefore used for low-rate communications. In addition, π / 2-BPSK modulation only supports discrete Fourier transform spread orthogonal frequency division multiplexing (DFT-s-OFDM), which is suitable for transmissions with limited coverage that require low transmit power back-off. DFT-s-OFDM is also used in optical transmission systems in combination with pulse amplitude modulation (PAM) (e.g., 4-PAM). The advantage of PAM (such as BPSK) is its low complexity, which is beneficial for the implementation of transmitters and receivers. On the other hand, the disadvantage is its low spectral efficiency, which can be improved by M-ary PAM (M-PAM) and the like. However, high-order PAM is not widely used in commercial cellular systems and requires a higher SNR compared to quadrature amplitude modulation (QAM). Therefore, QPSK modulation is commonly used, which has twice the spectral efficiency of BPSK.

[0003] The spectral efficiency of PAM (e.g., BPSK) can be improved by using single sideband (SSB) transmission. The SSB version of the signal occupies half the bandwidth of the original signal, but still carries the same amount of information. Therefore, the spectral efficiency is doubled. In addition, SSB transmission is used in photonics and optics (e.g., radio-over-fiber (RoF)) and is also combined with OFDM to provide high data rates in optical fibers. One problem is that SSB transmission will produce inter-symbol interference (ISI) of QAM (e.g., QPSK), so advanced signal processing and ISI elimination are required. However, for one-dimensional modulation, such as PAM, there is no ISI and SSB transmission can be utilized without ISI elimination. For DFT-s-OFDM, under the assumption of perfect channel equalization and time-invariant channel, the bit error rate (BER) of high-order M-PAM with SSB transmission is the same as 2M-QAM without SSB transmission. QPSK combined with DFT-s-OFDM has been widely used in cellular communications. The proposed scheme has shown that under certain conditions, BPSK with SSB transmission has the same BER but a higher peak-to-average-power ratio (PAPR) compared to QPSK. Moreover, in conventional DFT-s-OFDM, the modulation symbols are only time-multiplexed. In addition, SSB transmission in conventional DFT-s-OFDM is not very flexible and is subject to various limitations. Therefore, there are technical challenges to exploit the full potential of SSB transmission using DFT-s-OFDM.

[0004] Therefore, based on the above discussion, there is a need to overcome the above-mentioned disadvantages associated with using SSB transmission in combination with conventional DFT-s-OFDM. Summary of the invention

[0005] The present disclosure provides a signal transmission device and a method for the signal transmission device. The present disclosure provides a solution to solve the problem that the existing use of SSB transmission and conventional DFT-s-OFDM in combination is not flexible and is subject to multiple limitations. The purpose of the present disclosure is to provide a solution that at least partially overcomes the problems encountered in the prior art, and to provide an improved signal transmission device and an improved method for the signal transmission device to achieve higher reliability and spectrum efficiency.

[0006] One or more objects of the present disclosure are achieved by what is provided in the attached independent claims. Advantageous implementations of the present disclosure are further defined in the dependent claims.

[0007] In one aspect, the present disclosure provides a signal transmission device for transmitting modulation symbols using orthogonal frequency-division multiplexing (OFDM) based on discrete Fourier transform (DFT) precoding. The signal transmission device is also used to generate a DFT-s-OFDM signal by receiving an input (x[m]) including M modulation symbols (e.g., m=0, 1, ..., M-1), where M is an even number. The signal transmission device is also used to phase shift the input (x[m]) to generate a phase-shifted input The signal transmission device is also used to precode the phase shift input using DFT to generate M Fourier coefficients (X[k]) and sort the Fourier coefficients and select M / 2 Fourier coefficients. The signal transmission device is also used to generate the DFT-s-OFDM signal based on the selected M / 2 Fourier coefficients.

[0008] The disclosed signal transmission apparatus is capable of utilizing the full potential of SSB transmission with DFT-s-OFDM because in a DFT-s-OFDM signal with SSB transmission, modulation symbols are multiplexed in both the time and frequency domains by sorting and selecting Fourier coefficients, in contrast to conventional DFT-s-OFDM in which modulation symbols are multiplexed only in the time domain. This is advantageous in providing diversity gain in time-frequency selective channels. Furthermore, by sorting and selecting Fourier coefficients, the signal transmission apparatus exhibits improved block error rate (BLER) and peak-to-average power ratio (PAPR).

[0009] In one implementation, the generated DFT-s-OFDM signal also includes a cyclic prefix.

[0010] Through the cyclic prefix, the DFT-s-OFDM signal exhibits robustness.

[0011] In another implementation, the signal transmission device is characterized in that the time-discrete low-pass equivalent signal is generated by the following equation:

[0012]

[0013] Wherein, N represents the number of time samples, q[k] is a function for mapping Fourier coefficients to subcarriers, and g[k] is a function for selecting and sorting the Fourier coefficients.

[0014] By selecting and ordering the Fourier coefficients, the orthogonality condition is maintained.

[0015] In another implementation, the g[k] function is defined as:

[0016] g[k]=f[h[k]]

[0017] Among them, the function

[0018] f[i]=f 1 i+f 0 (modM)

[0019] And function

[0020]

[0021] Among them, M / P and P are integers, and the coefficient f 1 and f 0 is also an integer, where f 1 The greatest common divisor of and M is 1, and k = 0, 1, ..., M / 2-1, (modM) represents modulo M addition, Represents the floor function.

[0022] In another implementation, P is set to 1, f 1 Set to 2.

[0023] In another implementation, P is set to M, f 1 Set to the smallest integer greater than 1 such that f 1 The greatest common divisor of and M is 1.

[0024] In another implementation, the function f[i] is configured to generate other indices in addition to the corresponding h[k].

[0025] In another implementation, according to the signal transmission device according to any one of the above claims, the phase shift of the modulation symbol m is a complex exponential function is acquired, and the phase shift input Determined by the following equation:

[0026]

[0027] Among them, α, β and γ are real values.

[0028] In another implementation, the parameters of the function g[k] are determined to provide orthogonal signaling by satisfying the following equation:

[0029]

[0030] where C is a constant, δ[t] is the Kronecker delta function for integer t, Re{} is the real part operator, and * denotes the complex conjugate, where for m = 0, 1, …, M-1 and n = 0, 1, …, N-1, w[m,n] is defined as:

[0031]

[0032] In another implementation, the mapping q[k] is to a set of consecutive subcarriers.

[0033] In another implementation, the mapping q[k] is to a non-contiguous set of subcarriers.

[0034] In another implementation, the input symbol (x[m]) is a real-valued modulation symbol.

[0035] In another implementation, the input symbol (x[m]) is based on a π / 2 rotation pulse amplitude modulation (PAM) scheme, where a=0, β=π / 2, P=1, f 1 =1 and f 0 =(M+2) / 4.

[0036] The orthogonality condition is satisfied using the π / 2 rotation pulse amplitude modulation (eg π / 2 BPSK).

[0037] In another implementation, the input symbols (x[m]) are based on a Zadoff-Chu sequence.

[0038] The transmission of a chirp sequence can be achieved by using the Zadoff-Chu sequence.

[0039] On the other hand, the present disclosure provides a method for use in a signal transmission device, the signal transmission device being used to transmit modulation symbols based on discrete Fourier transform (DFT) precoding using orthogonal frequency-division multiplexing (OFDM). The method includes generating a DFT-s-OFDM signal by receiving an input (x[m]) including M modulation symbols, where M is an even number. The method also includes phase shifting the input (x[m]) to generate a phase shifted input (x[m]), and precoding the phase shifted input using DFT to generate M Fourier coefficients (X[k]). The method also includes sorting the Fourier coefficients and selecting M / 2 Fourier coefficients, and generating the DFT-s-OFDM signal based on the selected M / 2 Fourier coefficients.

[0040] The method achieves all the advantages and technical effects of the signal transmission device.

[0041] It should be appreciated that all of the above implementations may be combined.

[0042] It should be noted that all devices, elements, circuits, units and components described in this application can be implemented in software or hardware elements or any type of combination thereof. All steps performed by various entities described in this application and the described functions to be performed by various entities are intended to indicate that each entity is suitable for or used to perform corresponding steps and functions. Although in the description of the following specific embodiments, the specific functions or steps performed by external entities are not reflected in the description of the specific detailed elements of the entities performing the specific steps or functions, it should be clear to the technician that these methods and functions can be implemented by corresponding software or hardware elements or any combination thereof. It should be understood that the features of the present disclosure are easy to be combined in various combinations without departing from the scope of the present disclosure defined by the appended claims.

[0043] Other aspects, advantages, features and objects of the present disclosure will become apparent from the accompanying drawings and detailed description of illustrative implementations interpreted in conjunction with the following appended claims. BRIEF DESCRIPTION OF THE DRAWINGS

[0044] The above summary of the invention and the detailed description of the following illustrative embodiments can be better understood by reading in conjunction with the accompanying drawings. In order to illustrate the present disclosure, exemplary structures of the present disclosure are shown in the accompanying drawings. However, the present disclosure is not limited to the specific methods and tools disclosed herein. In addition, it should be understood by those skilled in the art that the accompanying drawings are not drawn to scale. Where possible, the same elements are represented by the same numbers.

[0045] Embodiments of the present disclosure are described below by way of example only with reference to the following drawings, in which:

[0046] Figure 1A is a block diagram illustrating various exemplary components of a signal transmission device according to an embodiment of the present disclosure;

[0047] Figure 1B is a network environment diagram illustrating communication between a transmitter and a receiver according to an embodiment of the present disclosure;

[0048] Figure 1C is a block diagram illustrating various exemplary components of a signal transmission device according to another embodiment of the present disclosure;

[0049] Figure 2 P and f according to an embodiment of the present disclosure are shown 1 A graphical representation of the block error rate (BLER) for different combinations of

[0050] Figure 3 A graphical representation showing the relationship between block error rate (BLER) and signal-to-noise ratio (SNR) according to an embodiment of the present disclosure;

[0051] Figure 4 A graphical representation showing the relationship between M modulation symbols and a signal-to-noise ratio (SNR) according to an embodiment of the present disclosure;

[0052] Figure 5 A graphical representation showing the relationship between peak-to-average-power-ratio (PAPR) and complementary cumulative distribution function (CCDF) according to an embodiment of the present disclosure;

[0053] Figure 6 A graphical representation showing the relationship between block error rate (BLER) and signal-to-noise ratio (SNR) according to an embodiment of the present disclosure;

[0054] Figure 7 is a flowchart of a method for a signal transmission device according to an embodiment of the present disclosure.

[0055] In the drawings, underlined numbers are used to indicate the item in which the underlined number is located or an item adjacent to the underlined number, and non-underlined numbers are items identified by a line connecting the non-underlined number to the item. When a number is not underlined and has an associated arrow, the non-underlined number is used to identify the general item to which the arrow points. DETAILED DESCRIPTION

[0056] The following detailed description describes embodiments of the present disclosure and ways in which these embodiments can be implemented. Although some modes of implementing the present disclosure have been disclosed, those skilled in the art will recognize that there may be other embodiments for implementing or practicing the present disclosure.

[0057] Figure 1A is a block diagram showing various exemplary components of a signal transmission device according to an embodiment of the present disclosure. Figure 1A , which shows a block diagram 100A of a signal transmission device 102 including a transmitter 104 and a receiver 106.

[0058] The signal transmission device 102 may include appropriate logic, circuits, interfaces and / or codes for modulating symbol communication. Examples of the signal transmission device 102 may include, but are not limited to, a transceiver, a base station, a user equipment, etc. The signal transmission device 102 may be used in applications such as ultra-reliable low latency communication (URLLC), vehicle-to-everything (V2X), etc.

[0059] The transmitter 104 may include suitable logic, circuitry, interfaces and / or code that may be operable to transmit modulation symbols to the receiver 106. Examples of the transmitter 104 may include, but are not limited to, a machine type communication (MTC) device, a computing device, a transmitting device, an evolved universal mobile telecommunications system (UMTS) terrestrial radio access (E-UTRAN) NR-dual connectivity (EN-DC) device, a server, custom hardware for wireless communication, or any other portable or non-portable electronic device, etc.

[0060] The receiver 106 may include suitable logic, circuitry, interfaces and / or code that may be operable to receive modulation symbols transmitted by the transmitter 104. Examples of the receiver 106 may include, but are not limited to, a server, a smartphone, custom hardware for wireless communication, a receiving device, or any other portable or non-portable electronic device.

[0061] Figure 1B is a network environment diagram illustrating communication between a transmitter and a receiver according to an embodiment of the present disclosure. Figure 1B Combination Figure 1A The elements of the reference are described. Figure 1B , a network environment diagram 100B is shown in which communication between a sender 104 and a receiver 106 is shown. A communication network 108 is also shown.

[0062] In one implementation, the transmitter 104 is configured to send the modulation symbols to the receiver 106 via a communication network 108 (e.g., a propagation channel). The communication network 108 includes a medium (e.g., a communication channel) through which the transmitter 104 may communicate with the receiver 106 of the signal transmission device 102. Examples of the communication network 108 may include, but are not limited to, a cellular network (e.g., a long-term evolution (LTE) 4G, 5G, or 5G NR network, such as a sub-6 GHz, centimeter wave, or millimeter wave communication network), a wireless sensor network (WSN), a cloud network, a local area network (LAN), a vehicle-to-everything (V2X) network, a metropolitan area network (MAN), and / or the Internet. The transmitter 104 in the network environment diagram 100B is configured to connect to the receiver 106 according to various wireless communication protocols. Examples of such wireless communication protocols, communication standards and techniques may include, but are not limited to, IEEE 802.11, 802.11p, 802.15, 802.16, 1609, Worldwide Interoperability for Microwave Access (Wi-MAX), Transmission Control Protocol and Internet Protocol (TCP / IP), User Datagram Protocol (UDP), Hypertext Transfer Protocol (HTTP), Long-term Evolution (LTE), File Transfer Protocol (FTP), Enhanced Data GSM Environment (EDGE), Voice over Internet Protocol (VoIP), email, instant messaging and / or Short Message Service (SMS) protocols, and / or other cellular or IoT communication protocols.

[0063] Figure 1C is a block diagram illustrating various exemplary components of a signal transmission device according to an embodiment of the present disclosure. Figure 1C Combination Figure 1A and Figure 1BThe elements of the reference are described. Figure 1C , which shows a block diagram 100C of a signal transmission device 102, which includes an antenna 110, a phase shifter 112, a precoder 114, a signal generator 116, a memory 118, and a processor 120. In one implementation, each of the antenna 110, the phase shifter 112, the precoder 114, the signal generator 116, the memory 118, and the processor 120 may be part of the transmitter 104. In another implementation, each of the antenna 110, the phase shifter 112, the precoder 114, the signal generator 116, the memory 118, and the processor 120 is an independent circuit or module (and may not be part of the transmitter 104).

[0064] The antenna 110 may include suitable logic, circuitry, interfaces and / or code for receiving an input (x[m]) comprising M modulation symbols. Examples of the antenna 110 may include, but are not limited to, a radio frequency transceiver, a network interface, a telematics unit, or any antenna suitable for use in a user device, a repeater, a base station, or other portable or non-portable communication device. The antenna 110 may communicate wirelessly using various wireless communication protocols.

[0065] The phase shifter 112 may comprise suitable logic, circuitry, interfaces and / or code that may be operable to phase shift an input (x[m]) to generate a phase shifted input

[0066] The precoder 114 may include suitable logic, circuitry, interfaces and / or code for performing a discrete Fourier transform (DFT) on the phase shifted input. Precoding is performed to generate M Fourier coefficients (X[k]). Examples of the precoder 114 may include, but are not limited to, a DFT precoder, a discrete cosine transform (DCT) precoder, and the like.

[0067] The signal generator 116 may comprise suitable logic, circuitry, interfaces and / or code that may be operable to generate a DFT-spread-orthogonal frequency division multiplexing (DFT-s-OFDM) signal based on the M / 2 selected Fourier coefficients.

[0068] The memory 118 may include suitable logic, circuitry, interfaces and / or code for storing machine code and / or instructions executable by the processor 120. The memory 118 may temporarily store one or more DFT-s-OFDM signals before the transmitter 104 of the signal transmission device 102 transmits the signals. Examples of implementations of the memory 118 may include, but are not limited to, an Electrically Erasable Programmable Read-Only Memory (EEPROM), a Random Access Memory (RAM), a Read Only Memory (ROM), a Hard Disk Drive (HDD), a flash memory, a Secure Digital (SD) card, a Solid-State Drive (SSD), a computer-readable storage medium and / or a CPU cache memory. The memory 118 may store an operating system and / or a computer program product for operating the signal transmission device 102. Computer-readable storage media used to provide non-transitory memory may include, but are not limited to, electronic storage devices, magnetic storage devices, optical storage devices, electromagnetic storage devices, semiconductor storage devices, or any suitable combination of the above.

[0069] The processor 120 may include appropriate logic, circuitry, interfaces, and / or code for executing instructions stored in the memory 118. Examples of the processor 120 may include, but are not limited to, an integrated circuit, a coprocessor, a microprocessor, a microcontroller, a complex instruction set computing (CISC) processor, an application-specific integrated circuit (ASIC) processor, a reduced instruction set (RISC) processor, a very long instruction word (VLIW) processor, a central processing unit (CPU), a state machine, a data processing unit, and other processors or circuits. In addition, the processor 120 may refer to one or more separate processors, processing devices, or processing units as part of a machine.

[0070] In operation, the signal transmission device 102 is used to transmit modulation symbols using orthogonal frequency-division multiplexing (OFDM) based on discrete Fourier transform (DFT) precoding. The signal transmission device 102 is also used to generate a DFT-s-OFDM signal by receiving an input (x[m]) including M modulation symbols, m=0,1,…,M-1, where M is an even number. The modulation symbols are multiplexed in both the time domain and the frequency domain, which provides diversity gain in the time-frequency selective channel, while in contrast, the modulation symbols in the conventional DFT-s-OFDM are only time-multiplexed. In addition, the modulation symbols are precoded using DFT precoding. The generated DFT-s-OFDM signal has improved reliability in terms of block error rate (BLER) and PAPR at low spectral efficiency. The antenna 110 of the signal transmission device 102 is used to receive the input (x[m]).

[0071] According to one embodiment, the input symbol (x[m]) is a real-valued modulation symbol. The input symbol (x[m]) is a real-valued modulation symbol, and M is an even number.

[0072] The signal transmission device 102 is further configured to generate a phase-shifted input by performing a phase shift on the input (x[m]). The DFT-s-OFDM signal is generated by using the phase shifter 112 to phase shift the input (x[m]) to generate the phase shifted input represented by equation (1)

[0073]

[0074] The modulation symbols x[m] are real-valued.

[0075] According to one embodiment, the phase shift of the modulation symbol m is derived from the complex exponential function and phase shift input Determined by the following equation:

[0076]

[0077] Among them, α, β and γ are real values.

[0078] Phase shift input According to equation (1), by considering the complex exponential function In addition, the parameters α, β and γ are all real-valued parameters.

[0079] According to one embodiment, the phase shift function includes α = 0. In one implementation, the complex exponential function for phase-shifting the input (x[m]) includes α = 0.

[0080] The signal transmission device 102 is further configured to generate a DFT-s-OFDM signal by precoding the phase-shifted input using the DFT, thereby generating M Fourier coefficients (X[k]). By using a precoder 114, for example, a DFT precoder precodes the phase-shifted input to generate M Fourier coefficients (X[k]). The generation of the M Fourier coefficients (X[k]) is represented by Equation (2):

[0081]

[0082] The signal transmission device 102 is further configured to generate a DFT-s-OFDM signal by sorting the Fourier coefficients and selecting M / 2 Fourier coefficients. Since only M / 2 Fourier coefficients are used, it is called SSB transmission. The generated M Fourier coefficients (X[k]) have been sorted, and M / 2 Fourier coefficients are selected therefrom.

[0083] According to one embodiment, the signal transmission device 102 is characterized in that the time-discrete low-pass equivalent signal is generated by the following equation:

[0084]

[0085] where N represents the number of time samples, q[k] is a function that maps the Fourier coefficients to subcarriers, and g[k] is a function for selecting and sorting the Fourier coefficients. The DFT-s-OFDM signal generated based on the selected M / 2 Fourier coefficients is represented by Equation (3). Since M is considered even and L = M / 2, this can maximize the spectral efficiency compared to the conventional DFT-s-OFDM with L = M. In addition, the selection of the Fourier coefficients is performed by the function g[k] and cannot be arbitrarily selected. The reason is that orthogonal signaling should be provided through the function g[k] and the phase modulation function. It can be mathematically proven that the specifically disclosed general function g[k] enables the selection of M / 2 Fourier coefficients, plus specific associated phase values α, β, and γ, thereby achieving orthogonal signaling. In the above implementation, the time-discrete signal s[n] is used. In another implementation, for 0 ≤ t < T, the low-pass equivalent time-continuous signal can be defined as where is the subcarrier frequency. For -T CP ≤ t < T, the low-pass equivalent time-continuous signal can be defined to include a cyclic prefix (CP) of length T CP .

[0086] The signal transmission device 102 is further configured to generate a DFT-s-OFDM signal based on the M / 2 selected Fourier coefficients. The DFT-s-OFDM signal is generated by using the signal generator 116 and based on the M / 2 selected Fourier coefficients. If there is CP, the sample n of the generated DFT-s-OFDM signal is -N CP ,-N CP The time discrete representation of +1,…,N-1 is expressed by equation (4):

[0087]

[0088] According to one embodiment, the generated DFT-s-OFDM signal also includes a cyclic prefix. The generated DFT-s-OFDM signal represented by equation (4) includes a cyclic prefix (CP), and the number of samples of the CP is N CP .

[0089] According to one embodiment, the mapping q[k] is a mapping to a continuous set of subcarriers. In one implementation, the mapping of 0 to (M / 2-1) Fourier coefficients can be a continuous set of N subcarriers. The one-to-one function q[k] maps the M / 2 Fourier coefficients (X[k]) to a subset of N subcarriers. Several types of mapping functions can be used, for example, continuous mapping (q[k]=k+d), comb mapping (q[k]=pk+d), where p and d are integers, or any other form of interleaved mapping or non-continuous mapping.

[0090] According to one embodiment, mapping q[k] is mapped to a non-contiguous set of subcarriers. In one implementation, the mapping of 0 to (M / 2-1) Fourier coefficients can be a non-contiguous set of N subcarriers. Non-contiguous mapping can bring higher frequency diversity than continuous mapping while achieving higher PAPR of the signal.

[0091] According to one embodiment, the g[k] function is defined as:

[0092] g[k]=f[h[k]] (5)

[0093] Among them, the function

[0094] f[i]=f 1 i+f 0 (mod M) (6)

[0095] And function

[0096]

[0097] Among them, M / P and P are integers, and the coefficient f1 and f 0 is also an integer, where f 1 The greatest common divisor of and M is 1, and k = 0, 1, ..., M / 2-1, (modM) represents modulo M addition, represents the floor function. The function according to equation (5) extracts M / 2 from the M Fourier coefficients in a block manner, where M / P (M / 2P) determines the number of consecutive indices of the block when P is an even (odd) integer. In other words, M / P should be an integer, and P is the number of blocks. According to equation (6), f 1 and f 0 is an integer, where f 0 Determine the starting index offset. Select f 1 The coefficients should be such that the index generated by f[i] is unique. 1 The greatest common divisor of and M is 1(gcd(f 1 ,M)=1), where gcd(A,B) is the greatest common divisor of A and B.

[0098] According to one embodiment, the function g[k] is used to provide consecutive indices from 0 to (M / 2-1). The function according to equation (5) extracts M / 2 from the M Fourier coefficients in a block manner, where M / P (M / 2P) determines the number of consecutive indices of the block when P is an even (odd) integer.

[0099] According to one embodiment, the g[k] function is used to provide non-contiguous indices. In one implementation, the g[k] function selects non-contiguous indices.

[0100] According to one embodiment, P is set to 1, f 1 Set to 2. In one implementation, P is set to 1, f 1 Set to 2, this will create the comb.

[0101] According to one embodiment, P is set to M, f 1 Set to the smallest integer greater than 1 such that f 1 The greatest common divisor of and M is 1. In one implementation, if P is set to M, then f 1 Set to the smallest integer greater than 1 such that f 1 The greatest common divisor of M is 1 or gcd(f 1 ,M)=1.

[0102] According to one embodiment, the function f[i] is configured to generate indices other than the corresponding h[k]. It should be noted that f[i] does not necessarily reorder the selected index h[k], but may generate indices other than its input. That is, for integer and A collection of possible

[0103] In an exemplary scenario, let h = {h[k]} and g = {g[k]} represent integer sequences, and consider the following example where M = 30. Since M = 2·3·5, P∈{1,2,3,5,6,10,15,30}, when P = 1, f 1 Can be 2, or, f 1 can be any integer whose prime factorization does not include P. The following calculation shows how to obtain the index.

[0104] Select the first M / 2 Fourier coefficients:

[0105] P=1、f 1 =1, f 0 =0

[0106] h={0,1,2,…,14}

[0107] g={0,1,2,…,14}

[0108] Select M / 2 consecutive Fourier coefficients:

[0109] P = 1, f 1 =1, f 0 ≠0

[0110] h={0,1,2,…,14}

[0111] g={f 0 ,f 0 +1,f 0 +2,…,f 0 +29}(mod 30)

[0112] Select other Fourier coefficients:

[0113] P=1、f 1 =2, f 0 ≠0

[0114] h={0,1,2,…,14}

[0115] g={f 0 ,f 0 +2,f 0 +4,…,f 0 +28}(mod 30)

[0116] P = 30, f 1 =1, f 0 ≠0

[0117] h={0,2,4,…,28}

[0118] g={f 0 ,f 0 +2,f 0 +4,…,f 0 +58}(mod 30)

[0119] Select based on a block of Fourier coefficients:

[0120] P=1、f 1 =7, f 0 =0

[0121] h={0,1,2,…,14}

[0122] g={0,7,14,21,28,5,12,19,26,3,10,17,24,1,8}

[0123] Selection based on multiple Fourier coefficient blocks:

[0124] P = 3, f 1 =1, f 0 =0

[0125] h={0,1,2,…,9,20,21,…,24}

[0126] The first block contains M / P=10 indices and the last block contains M / 2P=5 indices.

[0127] g={0,1,2,3,4,5,6,7,8,9,20,21,22,23,24}

[0128] P = 5, f 1 =1, f 0 =0

[0129] h={0,1,…,5,12,13,…,17,24,25,26}

[0130] The first two blocks contain M / P=6 indices and the last block contains M / 2P=3 indices.

[0131] g={0,1,2,3,4,5,12,13,14,15,16,17,24,25,26}

[0132] P = 3, f 1 =7, f 0 =0

[0133] h={0,1,2,…,9,20,21,…,24}

[0134] g={0,7,14,21,28,5,12,19,26,3,20,27,4,11,18}

[0135] P = 5, f 1 =7, f 0 =0

[0136] h={0,1,…,5,12,13,…,17,24,25,26}

[0137] g={0,7,14,21,28,5,24,1,8,15,22,29,18,25,2}

[0138] The advantage of using equations (5), (6) and (7) is that the function g[k] is obtained by 1 and f 0 The parameterization in provides a high degree of flexibility and the analytical tractability required to demonstrate orthogonal transmission. 1 and f 0 The introduction of also makes it possible to use complex-valued modulation symbols.

[0139] According to one embodiment, the parameters of the function g[k] are determined to provide orthogonal signaling by satisfying the following equation:

[0140]

[0141] where C is a constant, δ[t] is the Kronecker delta function for integer t, Re{} is the real part operator, and * denotes the complex conjugate, where for m = 0, 1, …, M-1 and n = 0, 1, …, N-1, w[m,n] is defined as:

[0142]

[0143] Specific values ​​of the parameters α, β and γ are generated together with the function of equation (5) to produce orthogonal signaling. To find such values, it is realized that the signal of equation (3) with N=M can be expressed in the form of equation (8):

[0144]

[0145] where w[m,n] is called the basis function. The key point is that since the input x[m] is a real-valued modulation symbol, if the complex-valued basis function is orthogonal in the real domain, that is, 0≤p≤M-1 and 0≤m≤M-1, the orthogonal signal is obtained according to equation (9):

[0146]

[0147] Where δ[mp] is the Kronecker delta function. If the basis functions are orthogonal in the real domain, a receiver (e.g., receiver 106) can detect the modulation symbols by correlating the received signal with the basis function set and extracting the corresponding real values. Equation (9) is equivalent to the following condition of equation (10):

[0148]

[0149] Orthogonality does not depend on the function q[k] or the parameter γ, and the present disclosure does not impose any restrictions on them. 1 and f 0 , Equation (10) has multiple solutions for (α, β). If α = 0, it is called linear phase; if α ≠ 0, it is called chirped phase. For example, the solution of equation (10) can be found by numerical search, and usually α and β are multiples of π / M.

[0150] It can be seen that for conventional DFT-s-OFDM, where all M DFT coefficients are used and there is no reordering of the DFT coefficients, the basis function becomes w[m,n] = δ[nm]. This means that the modulation symbol x[m] is transmitted over time samples n = m, i.e. the modulation symbols are time multiplexed. This is in contrast to (8), where the effect of the new basis function is that the modulation symbols become multiplexed in both the time and frequency domains.

[0151] In an exemplary scenario, if M=P=24, f 1 =7, f 0 =0, through searching, we can find that t 0 =0,1,…,48 and t 1 =1,…,23 satisfies t 0 and t 1 The solution of equation (10) for all combinations of .

[0152] In addition, closed-form solutions for some cases are derived and included in Table 1. However, these are not just 1 and f 0 The solution.

[0153] Table 1: Phase values ​​of associated parameters

[0154]

[0155] It can be demonstrated by numerical examples that the M Fourier coefficients X[k] are composed of M / 2 complex conjugate pairs, and the function g[k] of equation (5) selects one Fourier coefficient of each pair. Transmitting only one coefficient of a pair does not lose information, because the only difference between the complex conjugate pairs of Fourier coefficients is the sign of the complex valued part. Therefore, M modulation symbols x[m] can be signaled using only M / 2 Fourier coefficients. However, for the disclosed solution, the order relationship between the complex conjugate pairs of Fourier coefficients depends on the values ​​α, β, and P. For different values ​​of α, β, and P, the values ​​of the Fourier coefficients are also different.

[0156] According to one embodiment, the input symbol (x[m]) is based on a π / 2 rotation pulse amplitude modulation (PAM) scheme, where a=0, β=π / 2, P=1, f 1 =1 and f 0 =(M+2) / 4. Typically, only real-valued modulation symbols (e.g., BPSK) can be used for conventional SSB transmission, which excludes modulation formats such as π / 2-BPSK. One advantage of using function (5) is that if P, f 1 and f 0 This limitation is alleviated if the selection method enables orthogonal signaling when α=0 and β=π / 2. This means that α and β are predefined. Compared to conventional SSB transmission where the phase angle cannot be arbitrarily selected, the signal transmission device 102 is used to use a PAM (i.e., including BPSK) constellation that rotates by π / 2 radians between consecutive symbols. Due to the use of PAM with a π / 2 rotation, the Fourier coefficients have a specific symmetry when a phase shift of π / 2 radians is applied. Assuming k=0,1,…,M / 2, then for a PAM modulated signal with a phase rotation of π / 2, the Fourier coefficients are symmetric according to equations (11) and (12):

[0157]

[0158] In addition, assuming k = 1, 2, ..., M / 2-1, then:

[0159]

[0160] Using equations (11) and (12), we can see that if M is an even integer not divisible by 4, then there are M Fourier coefficients, each with a complex conjugate pair. This means that it is possible to select L = M / 2 Fourier coefficients, i.e., to select one from each complex conjugate pair. Similarly, if M is an even integer divisible by 4, then there are M-2 coefficients, each with a complex conjugate pair. This means that it is possible to select L = (M-2) / 2+2 = M / 2+1 Fourier coefficients, i.e., to select one from each complex conjugate pair and then select the remaining two Fourier coefficients.

[0161] Taking L = M / 2 as an example, by setting α = 0, β = π / 2, P = 1 and f 1 = 1 to achieve a π / 2 rotation of PAM. By evaluating equation (10), it can be confirmed that if f 0 =(M+2) / 4, to obtain the orthogonal signaling of this parameter combination. Since M is required to be an even integer that cannot be divided by 4, f 0 becomes an integer. In addition, it should be noted that symmetry can be obtained by further generalization, for example, if equation (1) is generalized using γ=π / 2 according to equation (13):

[0162]

[0163] but:

[0164] X[M / 2-k]=-X * [k] (14)

[0165] X[M / 2+k]=-X * [Mk] (15)

[0166] In addition, there are alternative signal representations (or equivalent matrix representations) for the DFT-s-OFDM signal represented by equations (1), (2) and (3). The equivalent matrix representation includes the following notation, where [.]′ and [.]H represent the transpose and the Hermitian transpose, respectively, and diag[·] is a diagonal matrix:

[0167] -DFT matrix: for k = 0, 1, ..., M-1 and l = 0, 1, ..., M-1, W M =[w kl ]and

[0168] - Phase matrix: for m=0,1,…,M-1,

[0169] -Selection matrix: for k = 0, 1, ..., M / 2-1 and l = 0, 1, ..., M-1, G = [g kl ] and g kl ∈{0,1}

[0170] - Mapping matrix: for k = 0, 1, ..., N-1 and l = 0, 1, ..., M / 2-1, Q = [q kl ] and q kl ∈{0,1}

[0171] -Symbol vector: for m=0,1,…,M-1,x=[x[m]]′

[0172] -Signal vector: for n=0,1,…,N-1, s=[s[n]]′

[0173] For the selection matrix, g kl = 1, and each row and column has M / 2 elements. kl = 1. The positions of the elements are determined by g[k] so that the kth row contains an element in column g[k].

[0174] For the mapping matrix, q kl = 1, and each row and column q kl = 1. The positions of the elements are determined by q[k] so that the kth row contains an element in column q[k].

[0175] Then, the transmitted signal (i.e., DFT-s-OFDM signal) can be written in the form of equation (16):

[0176]

[0177] The basis function defined by equation (8) is the matrix Column, so 2Re{w H w}=I, where I is the identity matrix. In addition, by considering the case of N=M, then by defining the precoding matrix according to equation (17):

[0178]

[0179] The equivalent operation is to perform precoding with P before the precoder 114, which can be seen from equation (18):

[0180]

[0181] Therefore, the transmitter 104 of the signal transmission device 102 can alternatively be implemented as a DFT-s-OFDM technique, where its input (ie, input (x[m])) is precoded according to equation (17).

[0182] According to one embodiment, the input symbol (x[m]) is based on the Zadoff-Chu sequence. In one implementation, the input symbol (x[m]) is based on the Zadoff-Chu sequence, and DFT-s-OFDM signal transmission can also be referred to as chirp sequence transmission. In one case, when the input symbol x[m]=1, the input to the precoder 114 (e.g., DFT precoder) is a chirp sequence The chirp sequence can also be referred to as a predefined reference or synchronization signal sequence. Alternatively, if there are different such sets of sequences, information can be transmitted by selecting the sequence (i.e., the sequence is a codeword), and the receiver (e.g., receiver 106) detects the selected sequence. In all cases, if the phase values are selected according to equation (10), orthogonal transmission is achieved. However, SSB transmission can also be applied to other phase values. The following properties represented by equations (19) and (20) can be proven mathematically and show the existence of symmetry in the Fourier coefficients under certain conditions.

[0183] Property 1. If there is a sequence y[m] that satisfies equation (19):

[0184] y[m]=y[M - m] (19)

[0185] then its Fourier coefficients satisfy:

[0186] Y[k]=Y[M - k] (20)

[0187] Consider the case where α = πu / M, β = 0, γ = 0. When gcd(u,M)=1, which is called the Zadoff-Chu sequence, it can be directly verified that the Zadoff-Chu sequence satisfies equation (19). The inverse u- 1 of u is defined according to equation (21):

[0188] u -1 u (mod M)≡1 (21)

[0189] It can be proven that if gcd(u,M)=1, then there exists a unique integer u -1 <M that satisfies equation (21). Thus, the DFT has the following property:

[0190]

[0191] Therefore, equations (21) and (22) imply that it is possible to reconstruct the sequence y[m] from M / 2 Fourier coefficients. The coefficients P, f 1 and f 0The requirement is that the sequence y[m] to be detected is complex-valued. However, from equation (20), it can be concluded that if P,f 1 and f 0 , so that the set If a is an integer (where ), then M / 2 Fourier coefficients are selected so that the sequence y[m] can be reconstructed. This condition ensures that only one Fourier coefficient is selected from each pair of symmetric coefficients according to equation (20).

[0192] In an exemplary scenario, consider P = 1, f 1 =1 and f 0 = 0. M / 2 Set to M / 2×M / 2 identity matrix, 0 M / 2 Let M / 2×M / 2 zero matrix be formed. According to equation (23), M / 2×M selection matrix is ​​formed:

[0193] G=(I M / 2 0 M / 2 ) (twenty three)

[0194] At a receiver (eg, receiver 106), the coefficients are replicated according to equations (20) and (22) by the following matrix as shown in equation (24):

[0195]

[0196] Among them, for According to equation (20), the matrix T M / 2 The Fourier coefficient k is repeated at frequency Mk, as shown in equation (25):

[0197]

[0198] Wherein, according to equation (22), the Fourier coefficient M / 2 is multiplied by the phase value, as shown in the following equation (26):

[0199]

[0200] Thus, by multiplying the matrix V in the receiver (ie, receiver 106) M / 2 , from the M / 2 selected and transmitted Fourier coefficients GW M The M Fourier coefficients of x are obtained from x, that is, the obtained parameters are defined by equation (27):

[0201] DGW M x=W M x (27)

[0202] It can be further simplified,

[0203]

[0204] Because it can be shown as:

[0205] T M / 2 =W M / 2 W M / 2 (29)

[0206] Therefore, the signal transmission device 102 is able to fully utilize the SSB transmission with DFT-s-OFDM because in the DFT-s-OFDM signal, the modulation symbols are multiplexed in both the time domain and the frequency domain. This is advantageous in providing diversity gain in time-frequency selective channels. In addition, by sorting and selecting Fourier coefficients, the signal transmission device 102 exhibits improved BLER and PAPR. Compared with conventional DFT-s-OFDM with SSB transmission, the signal transmission device 102 supports SSB transmission using a π / 2 rotated PAM constellation (i.e., π / 2-BPSK). The signal transmission device 102 is used to introduce a phase rotation (e.g., π / 2) into the BPSK modulation so that the orthogonality condition defined by equation (10) is satisfied. π / 2-BPSK is applicable to the physical uplink shared channel (PUSCH) channel of the LTE / NR system. In addition, the signal transmission device 102 is used to introduce a function g[k], which is used to select and sort Fourier coefficients commensurate with the phase shift according to the orthogonality condition. The signal transmission device 102 is used to introduce signaling that supports SSB transmission with DFT-s-OFDM, for example, UE capability signaling that notifies the network whether the UE supports the modulation format, and signaling sent by the network to the UE to use the modulation format. This can be an on / off signal or a more detailed scheme, for example, a new configurable modulation and coding scheme (MCS) table containing some entries with a new modulation format. In addition, the signal transmission device 102 can be applied to channels using DFT-s-OFDM, i.e., PUSCH in the uplink of LTE and new radio (NR). In LTE, the physical side-link control channel (PSCCH), physical side-link shared channel (PSSCH), physical side-link broadcast channel (PSBCH) and physical side-link discovery channel (PSDCH) also use DFT precoding. The above channels include the sidelink for LTE V2V. Among them, the narrowband Internet of Things (NB-IoT) system uses DFT-s-OFDM of narrowband PUSCH (NPUSCH). In addition, NR can be extended to operate in frequency bands higher than 71 GHz, which can be considered in Release 19 and later versions.Due to the special characteristics of this high frequency band (e.g., phase noise, low efficiency of power amplifier, etc.), OFDM may not be an appropriate waveform, and it is recommended to introduce DFT-s-OFDM in the downlink channel. Therefore, in the upcoming NR version, SSB transmission with DFT-s-OFDM may be applicable to the physical downlink shared channel (PDSCH) and / or the physical downlink control channel (PDCCH).

[0207] In addition, in one implementation, the receiver 106 can be implemented as a matched filter, which associates the received signal with a set of basis functions and extracts the real part. Alternatively, the receiver 106 can be used to perform the inverse operation of the transmitter 104 of the signal transmission device 102. For example, the transmitter 104 is used to perform DFT of M modulation symbols, and the receiver 106 is used to perform inverse DFT (inverse-DFT, IDFT) on the received M modulation symbols. In the special case of selecting other Fourier coefficients (i.e., P = 1 and f 1 =2 or P = M and f 1 = 1), the disclosed scheme enables efficient implementation of the transmitter 104 and the receiver 106. Then, it is sufficient to compute a DFT of size M / 2 (e.g., by a fast Fourier transform (FFT) algorithm) instead of size M. Assuming f 1 =2, then g[k] = 2k, and the following equation holds:

[0208]

[0209] Similarly, it can be proved that:

[0210]

[0211] Therefore, the modulation symbols are first preprocessed according to equations (30) and (31), and then a DFT of size M / 2 is performed. Therefore, there is no need to calculate a DFT of size M and then select M / 2 Fourier coefficients.

[0212] Figure 2 P and f according to an embodiment of the present disclosure are shown 1 A graphical representation of the block error rate (BLER) for different combinations of . Figure 2 Combination Figure 1A , Figure 1B and Figure 1C The elements of the reference are described. Figure 2, wherein a graphical representation 200 is shown, which includes: coefficients f ranging from 1 to 16 are shown 1 The x-axis 202 shows the range from 5x10 -3 Up to 11x10 -3 The Y-axis 204 shows the BLER values.

[0213] The BLER is evaluated on a time-frequency selective channel, where the power-delay curve follows the International Telecommunication Union (ITU) vehicle A model and the speed is set to 0 km / h or 500 km / h, as shown in Table 2. When the subcarrier spacing is 15 kHz and the carrier frequency is 6 GHz, 500 km / h corresponds to a maximum Doppler shift of 2.78 kHz, i.e., 19% of the subcarrier spacing. The number of symbols M is a multiple of 12, i.e., the number of subcarriers per resource block in LTE and NR, e.g., for DFT-s-OFDM with SSB transmission, M = 24 symbols are transmitted on one resource block with L = 24 / 2 = 12 subcarriers. The Fourier coefficients are continuously mapped to the subcarriers, i.e., q[k] = k.

[0214] Table 2: Evaluation parameters for BLER simulation

[0215] parameter set up Number of symbols M＝24、48、72、96、120 Channel Vehicle A, 0km / h, 500km / h Subcarrier spacing 15kHz Carrier frequency 6GHz Receiver MMSE Channel Code 3GPP polar code, code rate 1 / 3, 1 / 2, 2 / 3, 3 / 4 Channel Estimation ideal

[0216] Referring to graphical representation 200, first line 206, second line 208, third line 210, fourth line 212, fifth line 214, sixth line 216, and seventh line 218 collectively illustrate the relationship between P and f. 1 206 shows the BLER for different combinations of M=30 modulation symbols at a signal-to-noise ratio (SNR) of 12 dB. For example, the first line 206 shows the BLER for P=1, the second line 208 shows the BLER for P=3, and the third line 210 shows the BLER for different combinations of f 1 Similarly, fourth line 212 shows the BLER at P=6, fifth line 214 shows the BLER at P=10, sixth line 216 shows the BLER at P=15, and seventh line 218 shows the BLER at P=30 at f 1 In one implementation, P, f 1 and f 0 to minimize BLER. In the following, f 0 = 0, because its value has little effect on BLER. In addition, Figure 2The BLER at a fixed SNR value (e.g., 12 dB) when M=30 is shown, indicating the relationship between BLER and parameters P and f 1 Therefore, in the evaluation, P = M and f 1 Set to the smallest integer greater than 1 such that f 1 and M is 1 (i.e., gcd(f 1 ,M)=1) to provide gain for all M modulation symbols.

[0217] Figure 3 A graphical representation of the relationship between block error rate (BLER) and signal-to-noise ratio (SNR) according to an embodiment of the present disclosure is shown. Figure 3 Combination Figure 1A , Figure 1B and Figure 1C The elements of the reference are described. Figure 3 , wherein a graphical representation 300 is shown, which includes: an X-axis 302 showing a SNR ranging from 4dB to 20dB, an X-axis 303 showing a SNR ranging from 10 -5 to 10 0 The Y-axis 304 shows the BLER values.

[0218] The graphical representation 300 shows a conventional QPSK (eg, using M / 2 modulation symbols), a BPSK rotation scheme (ie, P=1 and f 1 = 1) and BPSK using the disclosed method (where P = 24 and f 1 =7), these schemes transmit the same number of bits and transmit on M / 2 subcarriers, and therefore have the same spectral efficiency and coding gain. Referring to the graphical representation 300, the first line 306, the second line 308, and the third line 310 collectively show the relationship between BLER and SNR on the vehicle A channel at 500 km / h. For example, the first line 306 shows the relationship between BLER and SNR for conventional QPSK. Similarly, the second line 308 shows the relationship between BLER and SNR for the conventional BPSK rotation scheme (where P = 1, f 1 =1). In addition, according to one embodiment, the third line 310 shows the relationship between BLER and SNR with P = 24 (ie, P = M), f 1 =7. In the graphical representation 300, M=24 modulation symbols are considered. This has the advantage that it is comparable to conventional QPSK and BPSK with P=1, f 1 = 1, the disclosed BPSK rotation scheme with P = 24 (ie, P = M), f 1=7 wins out with a BLER below 10%. Similar results are obtained for other values ​​of M modulation symbols (eg, for P=M), such as for M=48, 72, 96, 120.

[0219] Figure 4 A graphical representation of the relationship between M modulation symbols and signal-to-noise ratio (SNR) according to an embodiment of the present disclosure is shown. Figure 4 Combination Figure 1A , Figure 1B and Figure 1C The elements of the reference are described. Figure 4 , wherein a graphical representation 400 is shown, which includes: an X-axis 402 showing M modulation symbols ranging from 20 to 120, and a Y-axis 404 showing SNR values ​​ranging from 12dB to 18dB.

[0220] Referring to the graphical representation 400, the first line 406, the second line 408, and the third line 410 show the BLER of 10 on the vehicle A channel at 500 km / h with M modulation symbols. -3 For example, the first line 406 shows the relationship between M modulation symbols and the required SNR for conventional QPSK. Similarly, the second line 408 shows the relationship between M modulation symbols and conventional BPSK (where P = 1, f 1 =1). In addition, according to one embodiment, the third line 410 shows the relationship between M modulation symbols and (where P = M, f 1 =7). Referring to graphical representation 400, third line 410 shows that the gain is substantial and is on the order of about 0.5-2.5 dB, and is greater if the M modulation symbols are fewer (eg, for M=48).

[0221] Figure 5 A graphical representation of the relationship between peak-to-average-power-ratio (PAPR) and complementary cumulative distribution function (CCDF) according to an embodiment of the present disclosure is shown. Figure 5 Combination Figure 1A , Figure 1B and Figure 1C The elements of the reference are described. Figure 5 , wherein a graphical representation 500 is shown, which includes: an X-axis 502 showing a PAPR ranging from 0 to 8, an X-axis 503 showing a PAPR ranging from 10 -5 to 100 The CCDF values ​​are on the Y-axis 504 .

[0222] Referring to the graphical representation 500, a first line 506, a second line 508, and a third line 510 illustrate the relationship between the PAPR and CCDF of different modulation schemes by using an upsampling factor of 10. For example, the first line 506 shows the relationship between the PAPR and CCDF of conventional QPSK. Similarly, the second line 508 shows the relationship between the PAPR and CCDF of conventional BPSK (where P = 1, f 1 =1). In addition, according to one embodiment, the third line 510 shows the relationship between PAPR and CCDF with P = M = 24, f 1 =7. In addition, graphical representation 500 shows the relationship between PAPR and CCDF for BPSK with P=M=24, f 1 =7, the PAPR of the disclosed BPSK is the same as the PAPR of conventional QPSK. For a modulation scheme with an upsampling factor of 10, similar results are obtained when M = 120 modulation symbols are considered. However, in another implementation scenario, π / 2-BPSK with SSB transmission is compared with conventional QPSK. In this scenario, the PAPR of π / 2-BPSK with SSB transmission is about 1 dB lower than that of conventional QPSK.

[0223] Figure 6 A graphical representation of the relationship between block error rate (BLER) and signal-to-noise ratio (SNR) according to an embodiment of the present disclosure is shown. Figure 6 Combination Figure 1A , Figure 1B and Figure 1C The elements of the reference are described. Figure 6 , wherein a graphical representation 600 is shown, which includes: an X-axis 602 showing a SNR ranging from 4dB to 20dB, an X-axis 603 showing a SNR ranging from 10 -5 to 10 0 The Y-axis 604 shows the BLER values.

[0224] The graphical representation 600 shows a graph with P=M=48, f 1=7 BPSK and conventional QPSK. Referring to graphical representation 600, first line 606A, second line 606B, third line 606C, fourth line 606D, fifth line 608A, sixth line 608B, seventh line 608C, and eighth line 610D collectively show the relationship between BLER and SNR for different code rates, such as code rates of 1 / 3, 1 / 2, 2 / 3, and 3 / 4, on vehicle A channel at 500 km / h. For example, each of first line 606A, second line 606B, third line 606C, and fourth line 606D shows the relationship between BLER and SNR for conventional QPSK. In addition, according to one embodiment, each of fifth line 608A, sixth line 608B, seventh line 608C, and eighth line 608D shows the relationship between BLER and SNR for conventional QPSK with P=48 (i.e., P=M), f 1 =7 BPSK BLER and SNR relationship. In addition, from Figure 6 It can be seen that at BLER = 10 -3 , SSB transmission with code rate 3 / 4 (i.e., for the first line 606A and the fifth line 608A) achieves the same BLER as conventional QPSK with code rate 2 / 3, which corresponds to a throughput gain of 12.5%, as shown in FIG. Figure 6 In practice, control channels such as the physical uplink control channel (PUCCH) or the physical downlink control channel (PDCCH) are usually transmitted in 10 -2 The BLER of the physical uplink shared channel (PUSCH) and physical downlink shared channel (PDSCH) of URLLC is defined as 10 -3 Up to 10 -8 Range, V2X application is 10 -2 Up to 10 -5 .

[0225] Based on these evaluations, the signal transmission device 102 improves the reliability of transmitting small and medium payloads on channels with Doppler frequency shift. The PAPR can be reduced (e.g., by 1 dB). The lower BLER of SSB transmission means that a smaller SNR (e.g., about 0.5-2.5 dB) is required to support the same reliability as QPSK. Alternatively, SSB transmission can use a higher code rate, thereby having higher spectral efficiency than QPSK while providing the same BLER. The disclosed SSB transmission scheme is expected to be used to replace QPSK:

[0226] When the Doppler frequency is not negligible compared to the subcarrier spacing.

[0227] oThe Doppler frequency depends on the UE's speed and the carrier frequency.

[0228] For small or medium transport block sizes.

[0229] o Since the coding gain is smaller and the effect of Doppler frequency is larger, the gain is expected to be larger for small blocks.

[0230] For smaller BLER, for example, below 0.1%.

[0231] oFor smaller BLERs, the required SNR is large enough so that the effect of the Doppler frequency dominates, while transmission is expected to be noise limited at low SNRs.

[0232] For scenarios with limited coverage

[0233] o Conventional QPSK has a larger PAPR than SSB transmission using π / 2-BPSK.

[0234] Figure 7 is a flowchart of a method for a signal transmission device according to an embodiment of the present disclosure. Figure 7 Combination Figure 1A , Figure 1B and Figure 1C The elements of the reference are described. Figure 7 , which shows the Figure 1A The method 700 of the signal transmission device 102 of FIG. 700 includes steps 702 to 710. The signal transmission device 102 is used to perform the method 700.

[0235] A method 700 is provided for use in a signal transmission device 102, which is used to transmit modulation symbols using orthogonal frequency-division multiplexing (OFDM) based on discrete Fourier transform (DFT) precoding. The modulation symbols are multiplexed in both the time domain and the frequency domain, which provides diversity gain in a time-frequency selective channel. In addition, the modulation symbols are precoded using DFT precoding. The method 700 is applicable to transmissions with small payloads, large reliability requirements, and high rates defined for URLLC and V2X.

[0236] At step 702, method 700 includes generating a DFT-s-OFDM signal by receiving an input (x[m]) including M modulation symbols. The input (x[m]) includes M modulation symbols, where m=0, 1, ..., M-1. The input symbol (x[m]) is a real-valued modulation symbol, and M is an even number.

[0237] At step 704, the method 700 further includes generating a phase-shifted input ( ) to generate a DFT-s-OFDM signal. By using ( Figure 1C The phase shifter 112 performs phase shift on the input (x[m]) to generate a phase shifted input ( ), for example, already in Figure 1C Described in detail.

[0238] At step 706, method 700 further includes generating a DFT-s-OFDM signal by precoding the phase shifted input using DFT to generate M Fourier coefficients (X[k]). Precoding is performed to generate M Fourier coefficients (X[k]), for example, Figure 1C Described in detail.

[0239] At step 708, the method 700 further includes generating a DFT-s-OFDM signal by sorting the Fourier coefficients and selecting M / 2 Fourier coefficients. The generated M Fourier coefficients (X[k]) are sorted and M / 2 Fourier coefficients are selected therefrom. For example, Figure 1C Described in detail.

[0240] At step 710, the method 700 further includes generating a DFT-s-OFDM signal based on the M / 2 selected Fourier coefficients. The DFT-s-OFDM signal is generated by using ( Figure 1C The signal generator 116 generates, for example, Figure 1C Described in detail.

[0241] Steps 702 to 710 are merely illustrative, and other alternatives may be provided in which one or more steps are added, one or more steps are deleted, or one or more steps are provided in a different order without departing from the scope of the claims herein.

[0242] In one aspect, a computer program product is provided, including program instructions for executing method 700 when executed by one or more processors (e.g., processor 120) in a signal transmission device 102. In another aspect, a computer system is provided, including one or more processors (e.g., processor 120) and one or more memories (e.g., memory 118), the one or more memories (i.e., memory 118) storing program instructions, which, when executed by the one or more processors (i.e., processor 120), cause the one or more processors (i.e., processor 120) to execute method 700. In yet another aspect, the present disclosure provides a non-transitory computer-readable medium storing computer-implemented instructions, which, when executed by a computer, cause the computer to perform the operations of method 700.

[0243] Without departing from the scope of the present disclosure as defined by the appended claims, the embodiments of the present disclosure described above may be modified. "Including", "comprising", "incorporating", "having", "being / being", etc., used to describe and claim the present disclosure, are intended to be interpreted in a non-exclusive manner, i.e., items, components or elements not explicitly described may also exist. References to the singular should also be interpreted as involving the plural. The word "exemplary" used herein means "as an example, instance or illustration". Any embodiment described as "exemplary" is not necessarily interpreted as being more preferred or more advantageous than other embodiments, and / or excluding the combination of features of other embodiments. The word "optionally" used herein means "provided in some embodiments and not provided in other embodiments". It should be understood that certain features of the present disclosure described in the context of a single embodiment for the sake of clarity may also be provided in a single embodiment by combination. Conversely, the various features of the present disclosure described in the context of a single embodiment for the sake of brevity may also be provided individually or by any appropriate combination or in an appropriate form in any other described embodiment of the present disclosure.

Claims

1. A signal transmission device (102) for transmitting modulation symbols using orthogonal frequency division multiplexing (OFDM) based on discrete Fourier transform (DFT) precoding, wherein the signal transmission device (102) is further used to generate a DFT-spread OFDM (DFT-s-OFDM) signal by: Receiving an input (x[m]) comprising M modulation symbols, m=0, 1, ..., M-1, where M is an even number; The input (x[m]) is phase shifted to generate a phase shifted input Precoding the phase-shifted input using DFT to generate M Fourier coefficients (X[k]); Sorting the Fourier coefficients and selecting M / 2 Fourier coefficients; The DFT-s-OFDM signal is generated based on the M / 2 selected Fourier coefficients.

2. The signal transmission device (102) according to claim 1, in, The generated DFT-s-OFDM signal also includes a cyclic prefix.

3. The signal transmission device (102) according to claim 1 or 2, It is characterized in that The time-discrete low-pass equivalent signal is generated by the following equation: Wherein, N represents the number of time samples, q[k] is a function for mapping Fourier coefficients to subcarriers, and g[k] is a function for selecting and sorting the Fourier coefficients.

4. The signal transmission device (102) according to claim 3, in, The g[k] function is used to provide consecutive indices from 0 to (M / 2-1).

5. The signal transmission device (102) according to claim 3, in, The g[k] function is used to provide non-contiguous indexes.

6. The signal transmission device (102) according to claim 5, in, The g[k] function is defined as: g[k]=f[h[k]] Among them, the function f[i]=f 1 i+f 0 (mod M) And function Among them, M / P and P are integers, and the coefficient f 1 and f 0 is also an integer, where f 1 The greatest common divisor of and M is 1, and k = 0, 1, ..., M / 2-1, (mod M) represents modulo M addition, Represents the floor function.

7. The signal transmission device (102) according to claim 6, in, P is set to 1, f 1 Set to 2.

8. The signal transmission device (102) according to claim 6, in, P is set to M, f 1 Set to the smallest integer greater than 1 such that f 1 The greatest common divisor of and M is 1.

9. The signal transmission device (102) according to any one of claims 6 to 8, in, The function f[i] is used to generate other indexes except the corresponding h[k].

10. The signal transmission device (102) according to any one of the preceding claims, in, The phase shift of the modulation symbol m is derived from the complex exponential function is acquired, and the phase shift input Determined by the following equation: Among them, α, β and γ are real values.

11. The phase shift function according to claim 10, in, α＝0。 12. The signal transmission device (102) according to claim 10 or 11, in, The parameters of the function g[k] are determined to provide orthogonal signaling by satisfying the following equation: where C is a constant, δ[t] is the Kronecker delta function for integer t, Re{} is the real part operator, and * denotes the complex conjugate, where for m = 0, 1, …, M-1 and n = 0, 1, …, N-1, w[m,n] is defined as:

13. The signal transmission device (102) according to any one of the preceding claims, in, The mapping q[k] is a mapping to a set of consecutive subcarriers.

14. The signal transmission device (102) according to any one of claims 1 to 12, in, The mapping q[k] is a mapping to a non-contiguous set of subcarriers.

15. The signal transmission device (102) according to any one of the preceding claims, in, The input symbols (x[m]) are real-valued modulation symbols.

16. The signal transmission device (102) according to any one of claims 1 to 14, in, The input symbol (x[m]) is based on a π / 2 rotation pulse amplitude modulation (PAM) scheme, where a=0, β=π / 2, P=1, f 1 =1 and f 0 =(M+2) / 4.

17. The signal transmission device (102) according to any one of claims 1 to 14, in, The input symbols (x[m]) are based on a Zadoff-Chu sequence.

18. A method (700) for a signal transmission device (102), in, The signal transmission device (102) is used to transmit modulation symbols using orthogonal frequency division multiplexing (OFDM) based on discrete Fourier transform (DFT) precoding, and the method (700) includes generating a DFT-spread OFDM (DFT-s-OFDM) signal by the following operations: Receiving an input (x[m]) comprising M modulation symbols, where M is an even number; The input (x[m]) is phase shifted to generate a phase shifted input Precoding the phase-shifted input using DFT to generate M Fourier coefficients (X[k]); Sorting the Fourier coefficients and selecting M / 2 Fourier coefficients; The DFT-s-OFDM signal is generated based on the selected M / 2 Fourier coefficients.

19. A computer program product, in, The computer program product comprises program instructions for performing the method (700) according to claim 18 when executed by one or more processors in a signal transmission device (102).

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