A self-resonant wireless charging system with low voltage and current stress

Abstract

This invention discloses a low-voltage-current stress autonomous resonant wireless charging system, comprising a transmitter and a receiver. The transmitter includes an adaptive resonant circuit and a buck topology. The adaptive resonant circuit includes a DC-AC inverter structure, a transmitter resonant structure, and an energy negative feedback structure, used to achieve secondary injection of reactive power, converting reactive power into active power and realizing detuning correction. The buck topology is used to reduce voltage. The receiver includes a resonant structure and a semi-active rectification structure. The resonant structure is used to cancel the equivalent reactance of the receiver at the transmitter, ensuring the system is always in a resonant state. The semi-active rectification structure is used to maintain a constant equivalent load of the receiver at the transmitter, ensuring decoupling of the output current from the coupling coefficient and achieving constant current output. The system proposed in this invention achieves autonomous resonance under detuning conditions, with a fast response speed, and can meet various system detuning requirements caused by complex environments.

Description

Technical Field

[0001] This invention belongs to the field of wireless power transmission technology, and particularly relates to a low-voltage and current stress autonomous resonant wireless charging system. Background Technology

[0002] Wireless Power Transfer (WPT) technology can transfer electricity from the primary side to the secondary side without physical contact, eliminating the need for traditional plugs and sockets and improving convenience and safety. It has attracted widespread attention in fields such as smart homes, electric vehicles, and medical implants. However, under harsh operating conditions, the reactance parameters of WPT systems are prone to random fluctuations due to variations in coupling strength and resonant component parameters, leading to system detuning and even the risk of runaway.

[0003] Existing reactance control schemes use a component matrix to adjust the number of inductors and capacitors involved in operation in real time. While the principle is simple, smooth control is difficult because the values ​​of inductors and capacitors are fixed, and the system is large and costly. Cascaded DC-DC converters can achieve equivalent load control; however, this method lacks reactance control capability and can only weaken, not eliminate, detuning. Strategies using variable capacitors and inductors utilize switching devices to control the operating time of individual capacitors and inductors, achieving equivalent control of resonant parameters. These strategies offer advantages in cost and power density, but the high voltage stress required for the switching devices in series with the resonant network limits power density.

[0004] Therefore, the universal WPT detuning correction technology with high power density, low cost, and low voltage stress is of great significance for ensuring the safety and robustness of the system. Summary of the Invention

[0005] To address the aforementioned technical problems, this invention proposes a low-voltage and current stress autonomous resonant wireless charging system. This system can utilize simpler control logic to achieve rapid adaptive detuning correction to cope with various impedance interference factors. At the same time, it can reduce the voltage and current stress on the system structure, ensuring the safety and stability of the system. In addition, the system also achieves constant current output with decoupling of output current and coupling strength, thereby solving the problems existing in the prior art.

[0006] To achieve the above objectives, the present invention provides a low-voltage-current stress autonomous resonant wireless charging system, comprising a transmitter and a receiver.

[0007] The transmitter includes an adaptive resonant circuit and a buck topology;

[0008] The adaptive resonant circuit includes a DC-AC inverter structure, a transmitter resonant structure, and an energy negative feedback structure, which are used to realize the secondary injection of reactive power, converting reactive power into active power, and realizing detuning correction; the step-down topology is used to realize voltage reduction.

[0009] The receiving end includes a resonant structure and a semi-active rectification structure;

[0010] The resonant structure is used to cancel the equivalent reactance of the receiver at the transmitter, so that the system is always in a resonant state; the semi-active rectification structure is used to keep the equivalent load of the receiver at the transmitter constant, ensuring that the output current is decoupled from the coupling coefficient and realizing constant current output.

[0011] Optionally, the DC-AC inverter structure is a half-bridge inverter circuit composed of a first switch and a second switch, used to convert the input DC voltage into AC voltage.

[0012] The transmitter resonant structure includes a first resonant capacitor and a transmitter coil; one end of the first resonant capacitor is connected to the transmitter coil, and the other end is connected to a first energy storage capacitor and a first capacitor respectively; the other end of the transmitter coil is connected to the source of a first switching transistor and the drain of a second switching transistor respectively.

[0013] The energy negative feedback structure includes a third switch, a fourth switch, and a second energy storage capacitor; one end of the second energy storage capacitor is connected to the drain of the third switch, and the other end is connected to the source of the fourth switch.

[0014] Optionally, the energy negative feedback structure is used to control the phase difference between the drive signals on the third and fourth switches and the first and second switches in the DC-AC inverter structure, thereby achieving detuning correction.

[0015] Optionally, the buck topology includes a first energy storage capacitor, a first capacitor, a second capacitor, and an inductor;

[0016] One end of the inductor is connected to the first capacitor and the second capacitor respectively, and the other end is connected to the source of the third switch and the drain of the fourth switch respectively; the other end of the first capacitor is connected to the first energy storage capacitor and the transmitting coil respectively; the other end of the second capacitor is connected to the source of the fourth switch; one end of the first energy storage capacitor is connected to the first capacitor and the transmitting coil respectively, and the other end is connected to the source of the fourth switch.

[0017] The first capacitor, the second capacitor, and the inductor form a T-type buck network, which is used to transfer the voltage on the third and fourth switching transistors to the first energy storage capacitor, thereby reducing the voltage.

[0018] Optionally, the resonant structure includes a receiving coil and a second resonant capacitor; one end of the receiving coil is connected to the second resonant capacitor, and the other end is connected to the drain of the sixth switch and the cathode of the second diode, respectively; the other end of the second resonant capacitor is connected to the drain of the fifth switch and the cathode of the first switch.

[0019] The semi-active rectification structure includes a fifth switch, a sixth switch, a first diode, and a second diode; it also includes an output filter capacitor and an output resistor, which are connected in parallel across the two ends of the semi-active rectification structure. One end of the output filter capacitor and the output resistor are connected to the cathodes of the first diode and the second diode, respectively, and the other end is connected to the sources of the fifth switch and the sixth switch, respectively.

[0020] This invention also provides a voltage and current stress optimization method for a low-voltage and current stress autonomous resonant wireless charging system, comprising the following steps:

[0021] Based on a low-voltage-current stress autonomous resonant wireless charging system, the functional relationship between the feedback terminal voltage and the conversion voltage is obtained.

[0022] Based on the functional relationship between the feedback voltage and the switching voltage, the reactance fluctuation range for maintaining low voltage stress is obtained;

[0023] Based on a low-voltage and current stress autonomous resonant wireless charging system, the functional relationship of the feedback current is obtained.

[0024] Based on the functional relationship of the feedback current, the reactance fluctuation range for maintaining low current stress is obtained.

[0025] Optionally, before obtaining the functional relationship between the feedback voltage and the conversion voltage, it is necessary to obtain the functional relationship between the inverter voltage at the transmitter and the feedback voltage.

[0026] The functional relationship between the inverter voltage at the transmitting end and the feedback voltage is shown in the following equation:

[0027]

[0028] Among them, V fa1 With V fa2 These are the inverter voltage v1 and the feedback voltage v, respectively. i2 The amplitude, X1 is the total reactance of the coil branch, R eq This is the equivalent resistance of the receiver at the transmitter.

[0029] Optionally, the functional relationship between the feedback terminal voltage and the conversion voltage is as follows:

[0030]

[0031] Where Z3 is capacitor C T1 The impedance, Z4 is the capacitance C T2 The impedance, Z5 is the inductance L T1 The impedance, Z v2 V represents the equivalent impedance of the reactive power storage circuit. i2 It is the feedback terminal voltage, V C2 It is a voltage conversion.

[0032] Optionally, the condition that needs to be met to maintain the reactance fluctuation range under low voltage stress is shown in the following formula:

[0033]

[0034] Where ω is the system angular frequency.

[0035] Optionally, the functional relationship of the feedback current is as follows:

[0036] i LT1 =i LT11 +i LT12 ,

[0037] Among them, i LT1 For L T1 The total current on, i LT12 It is v i2 In L T1 The current on, i LT11 Is v1 in L T1 The current on Z in1 Is it only considering V? i1 The equivalent impedance of the system, Z in2 Is it only considering V? i2 The equivalent impedance of the system at time, v fa1 The fundamental frequency of the inverter voltage v1, v fa2 For the feedback terminal voltage v i2 The fundamental frequency, Z3 is the capacitance C T1 The impedance, Z4 is the capacitance C T2 The impedance, Z5 is the inductance L T1 Z2 is the impedance of capacitor C2, Z RL1 It is capacitor C T2 and inductor L T1 The parallel equivalent impedance.

[0038] Compared with the prior art, the present invention has the following advantages and technical effects:

[0039] This invention proposes a low-voltage and current stress autonomous resonant wireless charging system. This system achieves autonomous resonance under detuned conditions and has a fast response speed, which can meet various system detuning conditions caused by complex environments.

[0040] The autonomous resonant wireless charging system with low voltage and current stress proposed in this invention can obtain the functional relationship between the feedback voltage and the conversion voltage, as well as the functional relationship between the feedback current. Based on the functional relationship between the feedback voltage and the conversion voltage, the reactance fluctuation range for maintaining low voltage stress can be obtained; based on the functional relationship between the feedback current, the reactance fluctuation range for maintaining low current stress can be obtained. By adopting the above scheme, the problem of high voltage and current stress on the feedback side of the autonomous resonant structure is solved, and the voltage and current stress on the feedback side can be reduced to less than one time that of the input side voltage, greatly improving the safety performance of the system. Attached Figure Description

[0041] The accompanying drawings, which form part of this application, are used to provide a further understanding of this application. The illustrative embodiments and descriptions of this application are used to explain this application and do not constitute an undue limitation of this application. In the drawings:

[0042] Figure 1 This is a schematic diagram of the structure of a self-resonant wireless power supply system with low voltage and current stress according to an embodiment of the present invention.

[0043] Figure 2 This is a schematic diagram of the initial structure of the adaptive resonant circuit according to an embodiment of the present invention;

[0044] Figure 3 This is a schematic diagram illustrating the active power conversion principle of a present invention embodiment;

[0045] Figure 4 This is a voltage and current stress curve diagram of an embodiment of the present invention, wherein (a) is the voltage stress as a function of C T2 (a) is a curve showing the change of X1, and (b) is a curve showing the change of current stress with X1. Detailed Implementation

[0046] It should be noted that, unless otherwise specified, the embodiments and features described in this application can be combined with each other. This application will now be described in detail with reference to the accompanying drawings and embodiments.

[0047] It should be noted that the steps shown in the flowchart in the accompanying drawings can be executed in a computer system such as a set of computer-executable instructions, and although a logical order is shown in the flowchart, in some cases the steps shown or described may be executed in a different order than that shown here.

[0048] With technological advancements, some scholars have proposed constructing energy negative feedback loops in WPT systems to study real-time detuning correction mechanisms suitable for complex environments. Instead of eliminating reactive power, this involves introducing negative feedback control logic that transforms reactive power from a detuned state into reactive power to correct the detuning state, thus reusing the reactive power accumulated within the detuned WPT system. However, while introducing this spontaneous energy negative feedback mechanism, when the system's detuning degree is too high, the voltage stress at the feedback terminal will increase sharply due to excessive energy pickup, bearing several times the voltage and current stress at the transmitting end. This may pose potential safety and stability risks to the equipment.

[0049] To address the aforementioned issues, this embodiment proposes an autonomous resonant wireless power supply system with low voltage and current stress.

[0050] like Figure 1 As shown, the transmitter of this system mainly consists of two parts: an adaptive resonant circuit and a buck topology. The adaptive resonant circuit includes a DC-AC inverter structure, a transmitter resonant structure, and an energy negative feedback structure. The buck topology is a buck network composed of three capacitors and one inductor. The receiver mainly consists of a resonant structure and a semi-active rectification structure.

[0051] The DC-AC inverter structure is a half-bridge inverter circuit composed of two MOSFET switches S1 and S2, which converts the input DC voltage V... i1 Convert to AC voltage V1.

[0052] The transmitting resonant structure consists of a first resonant capacitor C1 and a transmitting coil L1 in the main circuit. One end of the transmitting coil L1 is connected to the first resonant capacitor C1, and the other end is connected to the first energy storage capacitor C2 and the first capacitor C... T1 Connected. One end of the first resonant capacitor C1 is connected to the source of MOSFET switch S1 and the drain of MOSFET S2, and the other end is connected to the transmitting coil L1; wherein, MOSFET switch S1 is the first switch and MOSFET switch S2 is the second switch.

[0053] The energy negative feedback structure consists of two MOSFET switches S3 and S4 and a second energy storage capacitor C. i2 Composition. Second energy storage capacitor C i2 One end is connected to the drain of MOSFET S3, and the other end is connected to the source of MOSFET S4. i2 It is the second energy storage capacitor C i2 The terminal voltage between the two ends. The terminal voltage between the source of MOSFET switch S3 and the drain of S4 is V. i2This is also the feedback terminal voltage mentioned in this invention. Secondary reactive power injection is achieved by controlling the phase difference between the drive signals on the two switches S3 and S4 at the energy negative feedback terminal and the two MOSFET switches S1 and S2 in the DC-AC inverter structure, converting reactive power into active power and realizing detuning correction. MOSFET switch S3 is the third switch, and MOSFET switch S4 is the fourth switch.

[0054] The step-down topology consists of a first energy storage capacitor C2 and a first capacitor C. T1 Second capacitor C T2 and inductor L T1 Composition, wherein the first capacitor C T1 Second capacitor C T2 and inductor L T1 The three components form a T-type buck network, which transmits the high voltage V that the S3 and S4 switches need to withstand. i2 The voltage v transferred to the first energy storage capacitor C2 c2 This achieves the voltage reduction function, lowering the feedback terminal voltage. C2 is the first energy storage capacitor, V c2 It is the voltage across its two ends. Inductance L T1 One end and the first capacitor C T1 Second capacitor C T2 One end is connected to the source of MOSFET switch S3 and the drain of MOSFET S4. The second capacitor C... T2 One end and inductor L T1 First capacitor C T1 One end is connected to the first terminal, and the other end is connected to the source of the MOSFET switch S4. The first capacitor C... T1 One end is connected to the second capacitor C T2 Inductor L T1 One end is connected to the first energy storage capacitor C2, and the other end is connected to the first energy storage capacitor C2 and the transmitting coil L1. One end of the first energy storage capacitor C2 is connected to the first capacitor C... T1 One end is connected to the transmitting coil L1, and the other end is connected to the source of the MOSFET switch S4.

[0055] The receiver resonant structure includes a receiving coil L s Second resonant capacitor C s Receiving coil L s One end and the second resonant capacitor C s One end is connected to the drain of the MOSFET switch S6 and the cathode of the second diode D2. The second resonant capacitor C... s One end and receiving coil L sOne end is connected to the drain of MOSFET switch S5 and the cathode of the first diode D1; wherein MOSFET switch S5 is the fifth switch and MOSFET switch S6 is the sixth switch.

[0056] The receiver's semi-active rectification structure includes two MOSFET switches S5 and S6 and two diodes D1 and D2. o This is the output filter capacitor, R is the output resistor, and C is the output filter capacitor. o Both R and R are connected in parallel across the semi-active rectifier. One end is connected to the cathode of diodes D2 and D1, and the other end is connected to the source of MOSFET switches S5 and S6.

[0057] As an optional implementation, depending on the function of each part, the transmitter topology can be divided into an adaptive resonant circuit and a buck topology. A primary topology based on adaptive resonant technology with energy negative feedback is as follows: Figure 2 As shown, v1 and v i2 The relationship between the amplitude and phase α is solved by formulas (1)-(3). Where, v fa With v fa2 They are square waves v1 and v i2 The fundamental frequency, V fa With V fa2 X1 is the amplitude, and X2 is the total reactance of the coil branch. Its operating logic is as follows:

[0058] 1) When the system is detuned, i z1 The phase difference with v1 introduces reactive power, resulting in a decrease in power transmission energy.

[0059] 2) such as Figure 3 As shown, if the total current i z1 Decomposed into active current i in phase with v1 a And the reactive current i that is ±90° out of phase with v1 b And then, in conjunction with a certain method, i b By extracting energy from the system and injecting it a second time at an appropriate time, reactive power can be converted into active power, thus achieving detuning correction.

[0060] 3) In practice, it can be composed of S3, S4, and C. i2 The circuit formed achieves the above logic: changing the phase of S3 and S4 can affect the flow into the second energy storage capacitor C. i2 The energy is filtered, and if the phase is set properly, C can be ensured. i2 While storing and delaying the release of reactive power, the initial active current i is not stored. a It has an impact.

[0061] 4) Depending on the nature of the total input reactance X1, there are two ways to set the phase of S3 and S4: if X1 is inductive, then v i2 Detuning correction can be achieved by laging v1 by approximately 90° without any additional operation; conversely, v... i2 It is about 90° ahead of V1.

[0062]

[0063] X1=j(ωL1-1 / ωC1)+X eq

[0064]

[0065] In the previous description, it can be seen from formula (2) that V i2 With V i1 The ratio is positively correlated with the proportion of reactance to resistance in the total input impedance. If X1 is too large, Figure 2 S3 and S4 in the middle will withstand several times the force of V. i1 The voltage stress caused by this leads to safety hazards. Therefore, a [measure / method] was introduced. Figure 1 C T1 C T2 and L T1 This is a T-type step-down circuit. The voltage across C2 is v. C2 Will replace Figure 2 Chinese v i2 The detuning correction function of capacitors, compared with switching devices, is smaller in size and has a higher voltage rating, making them more suitable for withstanding high voltage stress.

[0066] Assuming the reactance of each branch in the T-type step-down topology is as shown in formula (4), then v C2 With V i2 The amplitude relationship is shown in formula (5). Where Z v2 This represents the equivalent impedance of the Reactive power storage circuit.

[0067]

[0068] As shown in (6), for a T-type resonant network, certain conditions can be met to ensure that the input and output voltage ratio is constant, which is suitable for reducing the voltage stress of S3 and S4.

[0069] The current stress on S3 and S4 also needs to be considered, i. zv2 It can be obtained using the superposition principle. i2 In L T1 Current i on LT12 The calculation formula is as follows:

[0070]

[0071] v1 in L T1 Current i on LT11 The solution can then be obtained based on the principle of current shunting, as shown in the following formula.

[0072]

[0073] L T1 The total current i on LT1 for:

[0074] i LT1 =i LT11 +i LT12 (9)

[0075] The feedback voltage and current stress results are as follows Figure 4 As shown, by setting appropriate parameters, the voltage and current stress at the feedback end can be guaranteed to be within one time the voltage and current output from the DC power supply at the transmitting end.

[0076] The equivalent load of the semi-active rectifier at the receiving end is an impedance, not a typical pure resistor. When a rechargeable battery or supercapacitor acts as a load, the changes in output voltage and current are a slow process, so the load can be equivalent to a resistor R. Studies have shown that a semi-active rectifier and a resistive load can be expressed using the equivalent fundamental impedance as:

[0077] Zs eq =Rs eq +jXs eq (10)

[0078] In the formula, the equivalent reactance and equivalent resistance are respectively:

[0079]

[0080] Where θ is the conduction angle of the semi-active rectifier, it can be seen that with a constant load R, the equivalent reactance and equivalent resistance are only related to the conduction angle θ of the active rectifier. The total impedance at the receiving end is:

[0081]

[0082] According to the mutual inductance model, the total impedance Z at the receiving end r The equivalent impedance Z at the transmitting end eq for:

[0083]

[0084] in:

[0085]

[0086] Because the system of this invention has autonomous resonance characteristics, the reactance X of the receiver can be equivalent to that of the transmitter. eq This cancels out the resistance, ensuring the system remains in a resonant state at all times, with only the equivalent resistance R at the receiver being equal to that at the transmitter. eq This will affect the system output. By adjusting the conduction angle of the semi-active rectifier at the receiver, the equivalent load R at the transmitter can be ensured. eq The constant current ensures that the output current is decoupled from the coupling coefficient, thus achieving constant current output.

[0087] In summary, the topology proposed in this invention enables autonomous resonant wireless charging with low voltage and current stress, and the output has constant current characteristics.

[0088] The system proposed in this invention achieves autonomous resonance under detuned conditions with fast response speed, capable of handling various system detunings caused by complex environments. By adopting the above scheme, the problem of high voltage and current stress on the feedback side of the autonomous resonant structure is solved, reducing the voltage and current stress on the feedback side to less than half of the input side voltage, thus greatly improving the system's safety performance. The system ensures that the voltage and current stress remain below half the input voltage even with a coupling strength between 0.1 and 0.3. At an output power of 50W, the efficiency remains stable above 93%, and the output exhibits constant current characteristics; the current remains constant when the coil moves within a certain range, ensuring the system's stability.

[0089] The main difference between the research approach of this invention and most current research approaches is that it does not eliminate reactive power, but rather reuses the reactive power accumulated in the detuned WPT system. It constructs a negative feedback control logic that follows the process of "increased detuning degree → increased reusable reactive power → enhanced detuning correction capability → system eventually returns to resonance state". This allows for rapid adaptive detuning correction that can cope with various impedance interference factors without changing the resonant circuit structure and operating frequency.

[0090] This invention introduces a spontaneous energy negative feedback mechanism that "corrects the detuning state by introducing reactive power into reactive power." However, when the system detuning degree is too high, the voltage and current stress at the feedback end will increase sharply due to excessive energy pickup, and will be subjected to voltage and current stresses several times that at the transmitting end. This may bring hidden dangers to the safety and stability of the equipment. To address this, this invention introduces a step-down circuit structure to reduce the voltage and current stress at the energy negative feedback end, reducing the stress to less than one time that at the transmitting end, thus greatly improving the safety and performance of the energy negative feedback side.

[0091] The system proposed in this invention exhibits constant current output characteristics. By detecting the output current and changing the drive signal of the MOSFET in the semi-active rectifier structure, the equivalent impedance of the semi-active rectifier and output resistor at the receiving end can be controlled, thereby controlling the overall equivalent impedance of the receiving end at the transmitting end. Since the system is autonomously resonant, the equivalent reactance of the receiving end at the transmitting end is canceled out, leaving only the equivalent resistance. By keeping the equivalent resistance constant, the output current can be kept constant, unaffected by the coupling coefficient, thus achieving constant current output.

[0092] This invention also provides a voltage and current stress optimization method for a low-voltage and current stress autonomous resonant wireless charging system, comprising the following steps:

[0093] A specific topology that can reduce voltage and current stress is constructed based on the principle of energy negative feedback.

[0094] The relationship between the feedback terminal voltage and the conversion voltage is calculated based on the topology to obtain the reactance fluctuation range that can maintain low voltage stress.

[0095] The feedback current is calculated based on the topology and system circuit model to obtain the reactance fluctuation range that can maintain low current stress.

[0096] Furthermore, the specific topology that can reduce voltage and current stress is as follows: Figure 1 Medium capacitor C T1 C T2 and inductor L T1 The step-down circuit consists of C2 with its terminal voltage V. C2 Will replace Figure 2 Chinese v i2 The mistuning correction function.

[0097] Furthermore, the relationship between the inverter voltage at the transmitting end and the feedback voltage is as follows:

[0098]

[0099] Among them, V fa1 With V fa2 These are the inverter voltage v1 and the feedback voltage v, respectively. i2 The amplitude, X1 is the total reactance of the coil branch, R eq This is the equivalent resistance of the receiver at the transmitter.

[0100] Furthermore, the relationship between the feedback terminal voltage and the conversion voltage is as follows:

[0101]

[0102] Where Z3 is capacitor C T1 The impedance, Z4 is C T2The impedance, Z5 is the inductance L T1 impedance, Z v2 V represents the equivalent impedance of the Reactive Power Storage circuit. i2 It is the feedback terminal voltage, V C2 It is a voltage conversion.

[0103] Furthermore, the following conditions need to be met for blood pressure reduction:

[0104]

[0105] Where ω is the system angular frequency, and by selecting appropriate reactance values ​​of Z3, Z4, and Z5 after satisfying the above conditions, low voltage stress can be achieved.

[0106] Furthermore, the current relationship is as follows:

[0107]

[0108] Among them, i LT1 For L T1 The total current on, i LT12 It is v i2 In L T1 The current on, i LT11 Is v1 in L T1 The current on Z in1 Is it only considering V? i1 The equivalent impedance of the system, Z in2 Is it only considering V? i2 By determining the equivalent impedance of the system and selecting appropriate reactance values ​​for Z3, Z4, and Z5, low current stress can be achieved.

[0109] The above are merely preferred embodiments of this application, but the scope of protection of this application is not limited thereto. Any variations or substitutions that can be easily conceived by those skilled in the art within the scope of the technology disclosed in this application should be included within the scope of protection of this application. Therefore, the scope of protection of this application should be determined by the scope of the claims.

Claims

1. An autonomous resonant wireless charging system with low voltage current stress, comprising a transmitting end and a receiving end, characterized in that, the transmitting end comprises an adaptive resonant circuit and a step-down topology; wherein the adaptive resonant circuit comprises a DC-AC inverter structure, a transmitting end resonant structure and an energy negative feedback structure; the adaptive resonant circuit is used to realize the secondary injection of reactive power, convert the reactive power into active power, and correct the detuning; the step-down topology is used to realize the voltage reduction; the receiving end comprises a resonant structure and a semi-active rectification structure; wherein the resonant structure is used to offset the equivalent reactance of the receiving end at the transmitting end, so that the system is always in a resonant state; the semi-active rectification structure is used to constantly equivalent load of the receiving end at the transmitting end, to ensure the output current and the coupling coefficient decoupling, and to realize constant current output; the step-down topology comprises a first energy storage capacitor, a first capacitor, a second capacitor and an inductor; one end of the inductor is connected with the first capacitor and the second capacitor respectively, and the other end is connected with the source of the third switch tube and the drain of the fourth switch tube respectively; the other end of the first capacitor is connected with the first energy storage capacitor and the transmitting coil respectively; the other end of the second capacitor is connected with the source of the fourth switch tube; one end of the first energy storage capacitor is connected with the first capacitor and the transmitting coil respectively, and the other end is connected with the source of the fourth switch tube; wherein the first capacitor, the second capacitor and the inductor constitute a T-type step-down network, which is used to transfer the voltage on the third switch tube and the fourth switch tube to the first energy storage capacitor, to realize the voltage reduction.

2. The system according to claim 1, characterized in that, the DC-AC inverter structure is a half-bridge inverter circuit composed of a first switch tube and a second switch tube, which is used to convert the input DC voltage into AC voltage; the transmitting end resonant structure comprises a first resonant capacitor and a transmitting coil; one end of the transmitting coil is connected with the first resonant capacitor, and the other end is connected with the first energy storage capacitor and the first capacitor respectively; the other end of the first resonant capacitor is connected with the source of the first switch tube and the drain of the second switch tube respectively; the energy negative feedback structure comprises a third switch tube, a fourth switch tube and a second energy storage capacitor; one end of the second energy storage capacitor is connected with the drain of the third switch tube, and the other end is connected with the source of the fourth switch tube.

3. The system according to claim 2, characterized in that, the energy negative feedback structure is used to control the phase difference between the driving signals on the third switch tube and the fourth switch tube and the first switch tube and the second switch tube in the DC-AC inverter structure, to realize the detuning correction.

4. The system according to claim 1, characterized in that, the resonant structure comprises a receiving coil and a second resonant capacitor; one end of the receiving coil is connected with the second resonant capacitor, and the other end is connected with the drain of the sixth switch tube and the cathode of the second diode respectively; the other end of the second resonant capacitor is connected with the drain of the fifth switch tube and the cathode of the first switch tube. The semi-active rectifier structure comprises a fifth switch tube, a sixth switch tube, a first diode and a second diode; further comprising an output filter capacitor and an output resistor, which are connected in parallel at both ends of the semi-active rectifier structure, one end of which is connected with the cathode of the first diode and the second diode respectively, and the other end is connected with the source of the fifth switch tube and the sixth switch tube respectively.

5. A method of voltage and current stress optimization, the method comprising: The low-voltage current stress autonomous resonant wireless charging system based on any one of claims 1-4 comprises the following steps: The low-voltage current stress autonomous resonant wireless charging system obtains the functional relationship between the terminal voltage of the second energy storage capacitor and the terminal voltage of the first energy storage capacitor; wherein the terminal voltage of the second energy storage capacitor is taken as the feedback terminal voltage, and the terminal voltage of the first energy storage capacitor is taken as the conversion voltage. Based on the functional relationship between the terminal voltage of the second energy storage capacitor and the terminal voltage of the first energy storage capacitor, the range of reactance fluctuation for maintaining low-voltage stress is obtained. The low-voltage current stress autonomous resonant wireless charging system obtains the functional relationship of the current on the inductor in the step-down topology; wherein the current on the inductor in the step-down topology is taken as the feedback terminal current. Based on the functional relationship of the current on the inductor in the step-down topology, the range of reactance fluctuation for maintaining low-current stress is obtained.

6. The method of claim 5, wherein, Before obtaining the functional relationship between the feedback terminal voltage and the conversion voltage, the functional relationship between the inverter voltage of the transmitting end and the feedback terminal voltage needs to be obtained; The functional relationship between the inverter voltage of the transmitting end and the feedback terminal voltage is as follows: ， wherein V fa1 and V fa2 are the amplitudes of the inverse voltage v1 and the feedback voltage v i2 , respectively, and X1 is the total reactance of the coil branch, is the equivalent resistance of the receiving end at the transmitting end.

7. The method of claim 6, wherein, The functional relationship between the feedback terminal voltage and the conversion voltage is as follows: ， where Z3 is the impedance of the capacitor C T1 , Z4 is the impedance of the capacitor C T2 , Z5 is the impedance of the inductor L T1 , Z v2 represents the equivalent impedance of the reactive power storage circuit, V i2 is the voltage at the feedback terminal, and V C2 is the converted voltage.

8. The method of claim 7, wherein, The conditions that need to be met for obtaining the range of reactance fluctuation for maintaining low-voltage stress are as follows: ， Wherein, ω is the system angular frequency.

9. The method of claim 5, wherein, The functional relationship of the feedback terminal current is as follows: ， where i LT1 is the total current through L T1 , i LT12 is the current through v i2 , i T1 is the current through L LT11 , i T1 is the current through v1, Z in1 is the equivalent impedance of the system considering only V i1 , Z in2 is the equivalent impedance of the system considering only V i2 , v fa1 is the fundamental of the inverter voltage v1, v fa2 is the fundamental of the feedback voltage v i2 , Z3 is the impedance of the capacitor C T1 , Z4 is the impedance of the capacitor C T2 , Z5 is the impedance of the inductor L T1 , Z2 is the impedance of the capacitor C2, Z RL1 is the parallel equivalent impedance of the capacitor C T2 and the inductor L T1 .

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