Bilateral variable duty ratio phase shift modulation method for dual-active bridge converter

Abstract

The invention discloses a bilateral variable duty ratio phase shift modulation method for a dual-active bridge converter, and belongs to the technical field of converter phase shift modulation. The invention aims to solve the problem that an existing asymmetric duty ratio phase shift modulation strategy is difficult to consider current performance and soft switching range optimization at the same time. Comprising the steps that variable duty ratio modulation is adopted for all switching tubes of the dual-active bridge converter, and trigger signals of the switching tubes on the same bridge arm are complementary; according to the unequal relation among D0, D1 and D2, selecting a mode 1 to a mode 3 with the duty ratio of Q1, Q3, Q5 and Q7 being less than 50% for modulation; determining a per-unit value expression of a current peak-to-peak value of the inductor L based on the D0, the D1 and the D2, and optimizing the current peak-to-peak value by taking the per-unit value of the current peak-to-peak value as an optimization target and combining the per-unit value of the transmission power and constraint conditions of the D0, the D1 and the D2; and obtaining optimal solutions of D0, D1 and D2, and carrying out phase shift modulation on the dual-active bridge converter based on the optimal solutions. The method is used for phase shift modulation of the dual-active bridge converter.

Description

Technical Field

[0001] The invention relates to a double-side variable duty ratio phase shift modulation method for a dual active bridge converter, belonging to the technical field of converter phase shift modulation. Background Art

[0002] Bidirectional DC-DC converter is one of the core equipment of DC microgrid system, which plays an important role in bidirectional energy flow and maintaining system power balance. Among them, Dual Active Bridge (DAB) converter has become a commonly used topology in isolated bidirectional DC-DC converter due to its advantages such as wide voltage regulation range, bidirectional isolation, high power density and easy soft switching. It also has good application prospects in photovoltaic, electric vehicles and aerospace.

[0003] Phase-shift modulation is the mainstream modulation method of DAB converters, which realizes transmission power control by changing the phase difference of the driving signal between different bridge arms. For the widely used single-phase-shift modulation, extended-phase-shift modulation and triple-phase-shift modulation, the duty cycle of all switches is 50%, and the working waveform within half a switching cycle is symmetrical, so they can also be called symmetrical duty cycle phase-shift modulation (SDM) strategy. However, researchers have found that the symmetrical inductor current waveform in SDM control will lead to serious circulating power loss. Most strategies optimize the RMS current or reactive power of the inductor, and it is difficult to achieve ZVS of all switches within the full load range, especially in the low power range or when the voltage is mismatched.

[0004] Therefore, an asymmetric duty cycle phase shift modulation (ADM) strategy with a duty cycle of not 50% was proposed. However, the existing ADM strategy usually cannot take into account the optimization of current performance and soft switching range, resulting in limited improvement in light load efficiency and failure to maintain high efficiency under heavy load. Therefore, modulation strategies with better performance and higher efficiency need to be studied. Summary of the invention

[0005] Aiming at the problem that the existing asymmetric duty cycle phase shift modulation strategy is difficult to take into account both current performance and soft switching range optimization, the present invention provides a double-side variable duty cycle phase shift modulation method for a dual active bridge converter.

[0006] A double-side variable duty cycle phase shift modulation method for a dual active bridge converter of the present invention comprises:

[0007] One arm of the primary full-bridge of the dual-active-bridge converter includes switch Q1 and switch Q2, and the other arm includes switch Q3 and switch Q4. The input voltage is U1, and the input filter capacitor is C1; one arm of the secondary full-bridge includes switch Q5 and switch Q6, and the other arm includes switch Q7 and switch Q8. The output voltage is U2, and the output filter capacitor is C2; the output voltage at the midpoint of the primary full-bridge is U ab , and the output voltage at the midpoint of the secondary full-bridge is U cd ; the primary full-bridge and the secondary full-bridge are connected by a high-frequency transformer with a turns ratio of n, and the in-phase terminal of the primary side of the high-frequency transformer is connected to inductor L;

[0008] Variable duty cycle modulation is adopted for all switches, and the trigger signals of the switches on the same arm are complementary; the phase shift ratio between switch Q1 and switch Q4 is defined as the internal phase shift ratio D1, and the phase shift ratio between switch Q1 and switch Q5 is defined as the external phase shift ratio D2; the phase shift ratio between switch Q1 and switch Q3 is defined as D0; then the duty cycles of switches Q1, Q3, Q5, and Q7 are expressed as D0 + D1;

[0009] According to the inequality relationship among D0, D1, and D2, select modes 1 to 3 with the duty cycles of Q1, Q3, Q5, and Q7 less than 50% for modulation:

[0010] Mode 1: D1 ≤ D2 ≤ D0 + D1 ≤ 0.5, and D0 + D1 + D2 ≤ 1 - D0;

[0011] Mode 2: 0 ≤ D1 ≤ D2, and D0 + D1 ≤ 0.5;

[0012] Mode 3: D1 ≤ D2 ≤ D0 + D1 ≤ 0.5, and 1 - D0 ≤ D0 + D1 + D2;

[0013] When the per-unit value of the transmission power is in the interval [(2k - 2) / k 2 , 1], select mode 3 for operation; k represents the voltage transmission ratio;

[0014] When the per-unit value of the transmission power is in [0, (2k - 2) / k 2 , select to operate in mode 1 or mode 2;

[0015] The modulation methods for the three modes include:

[0016] Based on D0, D1, and D2, determine the per-unit value expression of the peak-to-peak current of inductor L. Taking the per-unit value of the peak-to-peak current as the optimization objective, optimize the peak-to-peak current in combination with the per-unit value of the transmission power and the constraint conditions of D0, D1, and D2; obtain the optimal solutions of D0, D1, and D2, and perform phase-shift modulation on the dual-active-bridge converter based on the optimal solutions.

[0017] According to the bilateral variable duty cycle phase - shift modulation method of the dual - active - bridge converter of the present invention, in mode 1, the calculation method of the per - unit value of the transmitted power is as follows:

[0018] Within a switching period, the inductor current i L (t) is:

[0019]

[0020] where t is time, t0 = 0, t1 = D1T s , t2 = D2T s , t3 = (D0 + D1)T s , t4 = (D0 + D1 + D2)T s , t5 = (1 - D0)T s , t6 = (1 - D0 - D1 + D2)T s , t7 = T s ; where T s is the switching period;

[0021] According to the integral of the inductor current within a switching period being 0, we get:

[0022]

[0023] Define the voltage transfer ratio k = U1 / nU2, then the current base value i N and the transmitted - power base value P N are:

[0024]

[0025] where f s is the switching frequency;

[0026] Then the per - unit value of the inductor current is calculated as:

[0027]

[0028] Then the per - unit value p * of the transmitted power is:

[0029]

[0030] According to the bilateral variable duty cycle phase - shift modulation method of the dual - active - bridge converter of the present invention, the per - unit value i Lpp * of the peak - to - peak value of the current of the inductor L is:

[0031] i Lpp * = 8[k(D0)+(-D0 - D1 + 2D2)].

[0032] According to the bilateral variable duty cycle phase - shift modulation method of the dual - active - bridge converter of the present invention, the conditions for all switching tubes to achieve soft - switching in Mode 1 are as follows:

[0033]

[0034] According to the bilateral variable duty cycle phase - shift modulation method of the dual - active - bridge converter of the present invention, the following optimization problem is established:

[0035]

[0036] According to the bilateral variable duty cycle phase - shift modulation method of the dual - active - bridge converter of the present invention, the Lagrangian extreme equation and optimization conditions are constructed:

[0037]

[0038]

[0039] In the formula, E represents the Lagrangian extreme equation, and λ is the Lagrange multiplier.

[0040] According to the bilateral variable duty cycle phase - shift modulation method of the dual - active - bridge converter of the present invention, the Lagrangian extreme equation is solved to obtain the optimal solutions of D0, D1, and D2:

[0041]

[0042] According to the bilateral variable duty cycle phase - shift modulation method of the dual - active - bridge converter of the present invention, the per - unit value p of the transmitted power in Mode 2 * is:

[0043] p * = 8(-D0D1 + 2D0D2);

[0044] The per - unit value p of the transmitted power in Mode 3 * is:

[0045] p * = 4(-1 + 4D0 - 4D0 2 + 2D1 - 6D0D1 - 4D1 2 + 2D2 + 4D1D2 - 4D2 2 );

[0046] The per - unit value i of the peak - to - peak current of the inductor L in Mode 2 Lpp * is:

[0047]

[0048] The per - unit value i of the peak - to - peak current of the inductor L in Mode 3 Lpp * is:

[0049] i Lpp * = 8[k(D0)+(-D0 - D1 + 2D2)].

[0050] According to the dual - active - bridge converter's dual - side variable duty - cycle phase - shift modulation method of the present invention, in mode 3, the conditions for all switching devices to achieve soft switching are:

[0051]

[0052] According to the dual - active - bridge converter's dual - side variable duty - cycle phase - shift modulation method of the present invention, in mode 3, the optimal solutions of D0, D1, and D2 are:

[0053]

[0054] Advantages of the present invention: The present invention proposes a dual - side variable duty - cycle phase - shift modulation (Dual - Side Variable Duty Cycle Phase - Shift Modulation, DVDM) method that combines extended phase - shift control (EPS control) and the ADM strategy. This method has three degrees of freedom involving three duty cycles. Different from the existing ADM schemes, in the method of the present invention, the primary and secondary full - bridges are driven by the same variable duty - cycle PWM signal. Therefore, the DVDM scheme actually controls four variables on the basis of having three degrees of freedom, including the duty cycles of the primary and secondary switching devices and the internal and external phase - shift ratios. This helps to improve the control performance without increasing the control complexity. In addition to the external phase - shift angle between the two full - bridges, there is also an internal phase - shift angle in the primary - side full - bridge. When the duty cycle of all switches is 50%, the EPS control can be regarded as a special case of this control strategy. By optimizing the peak - to - peak value of the inductor current, the RMS current can also be greatly reduced. In the high - power range, all power switches have soft - switching capabilities, and at the same time, the ZVS range under light load is extended, and both the conduction loss and the switching loss are effectively reduced, thus significantly improving the DAB efficiency. Description of the Drawings

[0055] Figure 1 is a schematic diagram of the topology structure of the dual - active - bridge converter described in the present invention; in the figure, T represents the high - frequency transformer,

[0056] Figure 2 is a typical driving - signal waveform diagram of the dual - side variable duty - cycle phase - shift modulation using the method of the present invention; in the figure, d1 represents the duty cycle of Q1, Q3, Q5, and Q7, and d2 represents the duty cycle of Q2, Q4, Q6, and Q8;

[0057] Figure 3 is a schematic diagram of six modulation modes of the method of the present invention;

[0058] Figure 4 It is a schematic diagram of the working waveform of Mode 1;

[0059] Figure 5 It is a schematic diagram of the working waveform of Mode 2;

[0060] Figure 6 It is a schematic diagram of the working waveform of Mode 3;

[0061] Figure 7 It is the experimental waveform diagram of voltage and current when k = 2 and the per-unit value of transmission power p * = 0.3; In the figure, ZVS indicates that soft switching can be achieved, No ZVS indicates that soft switching cannot be achieved, and time is time;

[0062] Figure 8 It is the experimental waveform diagram of voltage and current when k = 2 and the per-unit value of transmission power p * = 0.6;

[0063] Figure 9 It is the experimental waveform diagram of voltage and current when k = 2 and the per-unit value of transmission power p * = 0.8;

[0064] Figure 10 It is the experimental waveform diagram of the voltage and drive signal of switch Q1 when k = 2 and the per-unit value of transmission power p * = 0.3; In the figure, u ds1 is the voltage of switch Q1, Duty is the duty cycle of switch Q1, and u gs1 is the drive signal of switch Q1;

[0065] Figure 11 It is the experimental waveform diagram of the voltage and drive signal of switch Q4 when k = 2 and the per-unit value of transmission power p * = 0.3;

[0066] Figure 12 It is the experimental waveform diagram of the voltage and drive signal of switch Q5 when k = 2 and the per-unit value of transmission power p * = 0.3;

[0067] Figure 13 It is the experimental waveform diagram of the voltage and drive signal of switch Q8 when k = 2 and the per-unit value of transmission power p * = 0.3;

[0068] Figure 14 It is the experimental waveform diagram modulated by the method of the present invention when k = 2 and the per-unit value of transmission power p * = 0.2; In the figure, i Lrms represents the RMS current value of inductor L; In the figure, i Lpp represents the actual value of the peak-to-peak current of inductor L;

[0069] Figure 15 It is the experimental waveform diagram using FDM modulation when k = 2 and the per-unit value of transmission power p * = 0.2;

[0070] Figure 16 It is the experimental waveform diagram using FDM modulation when k = 2 and the per-unit value of transmission power p * = 0.2;

[0071] Figure 17 It is the experimental waveform diagram using the modulation method of the present invention when k = 2 and the per-unit value of transmission power p * = 0.7;

[0072] Figure 18 It is the experimental waveform diagram using FDM modulation when k = 2 and the per-unit value of transmission power p * = 0.7;

[0073] Figure 19 It is the experimental waveform diagram using OADM modulation when k = 2 and the per-unit value of transmission power p * = 0.7;

[0074] Figure 20 It is the comparison diagram of the efficiency curves of the modulation method of the present invention, FDM and OADM when k = 2. Detailed implementation manners

[0075] Next, the technical solutions in the embodiments of the present invention will be clearly and completely described in conjunction with the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all of the embodiments. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts shall fall within the protection scope of the present invention.

[0076] It should be noted that, without conflict, the embodiments in the present invention and the features in the embodiments may be combined with each other.

[0077] Next, the present invention will be further described in conjunction with the accompanying drawings, but it is not a limitation of the present invention.

[0078] In combination with Figures 1 to 6 as shown, the present invention provides a method for bilateral variable duty cycle phase shift modulation of a dual-active bridge converter, which is characterized by including

[0079] One arm of the primary full-bridge of the dual-active-bridge converter includes switch Q1 and switch Q2, and the other arm includes switch Q3 and switch Q4. The input voltage is U1, and the input filter capacitor is C1; one arm of the secondary full-bridge includes switch Q5 and switch Q6, and the other arm includes switch Q7 and switch Q8. The output voltage is U2, and the output filter capacitor is C2; the midpoint output voltage of the primary full-bridge is U ab , and the midpoint output voltage of the secondary full-bridge is U cd ; the primary full-bridge and the secondary full-bridge are connected by a high-frequency transformer with a turns ratio of n, and the in-phase end of the primary side of the high-frequency transformer is connected to inductor L;

[0080] Variable duty cycle modulation is adopted for all switches, and the trigger signals of the switches on the same arm are complementary; the phase shift ratio between switch Q1 and switch Q4 is defined as the internal phase shift ratio D1, and the phase shift ratio between switch Q1 and switch Q5 is defined as the external phase shift ratio D2; the phase shift ratio between switch Q1 and switch Q3 is defined as D0; then the duty cycles of switches Q1, Q3, Q5, and Q7 are expressed as D0 + D1;

[0081] According to the inequality relationship among D0, D1, and D2, select modes 1 to 3 in which the duty cycles of Q1, Q3, Q5, and Q7 are less than 50% for modulation:

[0082] When the per-unit value of the transmission power is in the interval [(2k - 2) / k 2 , 1], select mode 3 for operation; k represents the voltage transfer ratio;

[0083] When the per-unit value of the transmission power is in [0, (2k - 2) / k 2 , select to operate in mode 1 or mode 2;

[0084] Mode 1: D1 ≤ D2 ≤ D0 + D1 ≤ 0.5, and D0 + D1 + D2 ≤ 1 - D0;

[0085] Mode 2: 0 ≤ D1 ≤ D2, and D0 + D1 ≤ 0.5;

[0086] Mode 3: D1 ≤ D2 ≤ D0 + D1 ≤ 0.5, and 1 - D0 ≤ D0 + D1 + D2;

[0087] The modulation methods for the three modes include:

[0088] Based on D0, D1, and D2, determine the per-unit value expression of the peak-to-peak current of inductor L. Taking the per-unit value of the peak-to-peak current as the optimization target, optimize the peak-to-peak current in combination with the per-unit value of the transmission power and the constraint conditions of D0, D1, and D2; obtain the optimal solutions of D0, D1, and D2, and perform phase-shift modulation on the dual-active-bridge converter based on the optimal solutions.

[0089] The topology of the DAB converter is as shown in Figure 1 where L is the sum of the leakage inductance of the transformer and the series auxiliary inductance.

[0090] To better improve the efficiency of DAB and expand its ZVS range, especially under light load conditions, this embodiment proposes a bilateral variable duty cycle phase-shift modulation (Dual-Side Variable Duty Cycle Phase-Shift Modulation, DVDM) strategy that combines EPS control with three degrees of freedom and asymmetric phase-shift modulation. The main working principle and mode classification of the DVDM strategy are as follows:

[0091] The switching tubes on both sides of the transformer adopt variable duty cycle modulation, and the trigger signals of the switching tubes on the same bridge arm are complementary. The typical driving signal waveforms of the DVDM strategy are as shown in Figure 2 where the phase shift ratio between the primary-side switching tubes Q1 (Q2) and Q4 (Q3) is defined as the inner phase shift ratio D1, and the outer phase shift ratio D2 is the phase shift ratio between the switching tubes Q1 and Q5.

[0092] According to the inequality relationship between D0, D1, and D2, the six modes of DVDM are as shown in Figure 3 and their boundary conditions are listed respectively. These six modes consider the cases of D2>D0+D1, D2>1-D0, and D1>1-D0-D1+D2, because in this case the product of U ab and U cd will be less than 0 for a long time, resulting in an increase in reactive power and transformer losses. This embodiment adopts modes 1 to 3 with D0+D1≤0.5, and modes 4 to 6 are not considered because their power transmission capabilities are weak. The working waveforms of modes 1 to 3 are as shown in Figure 4 where.

[0093] Furthermore, taking mode 1 as an example, the calculation method of the per-unit value of the transmitted power in mode 1 is as follows:

[0094] Calculate its inductor current and transmitted power. In one switching period, the inductor current i L (t) is:

[0095]

[0096] where t is time, t0 = 0, t1 = D1T s , t2 = D2T s , t3 = (D0+D1)T s , t4 = (D0+D1+D2)T s , t5 = (1-D0)T s , t6 = (1-D0-D1+D2)Ts , t7 = T s ; where T s is the switching period;

[0097] Based on the fact that the integral of the inductor current over one switching period is 0, we get:

[0098]

[0099] Define the voltage transfer ratio k = U1 / nU2, then the current base value i N and the transfer power base value P N are:

[0100]

[0101] where f s is the switching frequency;

[0102] Then the per-unit value of the inductor current is calculated as:

[0103]

[0104] Then the per-unit value of the transfer power p * is:

[0105]

[0106] The per-unit value of the transfer power in Mode 2 and Mode 3 can be calculated similarly.

[0107] Optimized DVDM modulation strategy based on the peak-to-peak value of the inductor current:

[0108] When considering the losses of DAB, the conduction loss and winding loss are proportional to the square of the Root Mean Square (RMS) current. Therefore, the RMS current is often selected as the optimization target. However, DVDM has three degrees of freedom, and its RMS current calculation and expression are very complex, making it difficult to optimize. The peak-to-peak value of the inductor current i Lpp * not only has a simple expression but also can reflect the variation law of the RMS current. Therefore, the peak-to-peak value of the inductor current is selected as the optimization target.

[0109] The minimum value of the inductor current in Mode 1 appears at time t0, and the maximum value appears at time t3, as shown in the previous equation.

[0110] The per-unit value of the peak-to-peak value of the current of inductor L, i Lpp * is:

[0111] i Lpp * = 8[k(D0) + (-D0 - D1 + 2D2)].

[0112] To reduce the loss of the switching device and improve the transmission efficiency of the DAB, it is necessary to make the switching device achieve soft switching, that is, to reduce the voltage across its two ends to zero before the switching device is turned on. For Mode 1, the conditions for all switching devices to achieve soft switching are i L (t0) ≤ 0(Q1), i L (t1) ≤ 0(Q4), i L (t3) ≥ 0(Q2), i L (t5) ≥ 0(Q3), i L (t2) ≥ 0(Q 5,8 ), i L (t4) ≤ 0(Q6), i L (t6) ≤ 0(Q7). Therefore, it can be summarized as follows.

[0113] It is determined that the conditions for all switching devices to achieve soft switching in Mode 1 are:

[0114]

[0115] When dealing with optimization problems containing equality and inequality constraints, the Lagrange multiplier method (LMM) and the Karush-Kuhn-Tucker (KKT) conditions are widely used methods. The KKT conditions describe some necessary conditions that the optimal solution should satisfy. Taking the transmission power model as the equality constraint, the above soft switching conditions and mode boundary conditions as the inequality constraints, i Lpp * as the optimization objective.

[0116] Furthermore, the optimization problem is established as follows:

[0117]

[0118] Construct the Lagrangian extreme equation and the optimization conditions (KKT conditions):

[0119]

[0120] In the formula, E represents the Lagrangian extreme equation, and λ is the Lagrange multiplier.

[0121] Solve the Lagrangian extreme equation to obtain the optimal solutions of D0, D1, and D2 in Mode 1:

[0122]

[0123] Under the optimal phase shift ratio combination, for k and p * the minimum i Lpp * expression can be calculated as:

[0124]

[0125] Then, according to the boundary conditions of Mode 1, calculate the transmission power range operating in this mode:

[0126]

[0127] When DAB works in Mode 2, there are different expressions for the peak-to-peak value of the inductor current according to the inequality relationship of the phase shift ratio, and its analysis is rather complicated. However, according to the expression of the optimal solution in Mode 1, it is found that the optimal phase shift ratio of Mode 1 satisfies D1 = D2, which is the division boundary between Mode 1 and Mode 2. Therefore, the optimal phase shift ratio of Mode 2 is also located at this boundary and is the same as the optimal solution of Mode 1.

[0128] Furthermore, the per-unit value of the transmission power p under Mode 2 * is:

[0129] p * = 8(-D0D1 + 2D0D2);

[0130] The per-unit value of the transmission power p under Mode 3 * is:

[0131] p * = 4(-1 + 4D0 - 4D0 2 + 2D1 - 6D0D1 - 4D1 2 + 2D2 + 4D1D2 - 4D2 2 );

[0132] The per-unit value of the peak-to-peak current of the inductor L under Mode 2, i Lpp * is:

[0133]

[0134] The per-unit value of the peak-to-peak current of the inductor L under Mode 3, i Lpp * is:

[0135] i Lpp * = 8[k(D0) + (-D0 - D1 + 2D2)].

[0136] The same as Mode 1.

[0137] Under Mode 3, the conditions for all switching tubes to achieve soft switching are:

[0138]

[0139] Since the optimal control analysis of Mode 3 is similar to that of Mode 1, it is deduced that under Mode 3, the optimal solutions of D0, D1, and D2 are:

[0140]

[0141] In mode 3, i Lppmin * is expressed as:

[0142]

[0143] The transmission power in mode 3 must satisfy the condition:

[0144]

[0145] From the optimal solution of mode 3, it can be seen that the optimal phase shift ratio satisfies the relationship D1 + D2 = 0.5, which indicates that the duty cycle of all switches remains constant at 0.5. The optimized control in mode 3 is actually EPS control.

[0146] To sum up, in the high power range p * ∈[(2k - 2) / k 2 , 1], DAB operates in mode 3, while in the low power range p * ∈[0, (2k - 2) / k 2 , DAB operates in mode 1 (mode 2). The optimal phase shift ratio combinations of the DAB converter in modes 1 to 3, i Lppmin * , the corresponding transmission power ranges, and the duty cycles of all switch tubes are as follows:

[0147] Modes 1 and 2:

[0148]

[0149]

[0150]

[0151] Duty cycles of Q1, Q3, Q5, Q7:

[0152] Duty cycles of Q2, Q4, Q6, Q8:

[0153] Mode 3:

[0154]

[0155] Duty cycles of Q1, Q3, Q5, Q7:

[0156] Duty cycles of Q2, Q4, Q6, Q8:

[0157] Verification experiment:

[0158] To verify the above theoretical analysis, a DAB experimental prototype with 100 kHz, 50V / 25V, 10A was built. The experimental results of DAB under the DVDM, FDM, and OADM strategies at k = 2 were given, and the efficiency curves of the above three strategies were compared.

[0159] Figures 7 to 9 The steady-state working waveforms at different transmission powers under the DVDM strategy at k = 2 were given. Figures 10 to 13 Then it shows at k = 2, p * = 0.3, the voltage waveforms and drive signals of the switching transistors under the DVDM strategy. It can be seen from the figure that when p * = 0.3, DAB operates at the same internal and external phase shift ratios in the low-power section, and Q4 and Q5 conduct simultaneously. The duty cycles of all switches are not 50%. Except for switching transistors Q3 and Q4, other switching transistors achieve soft switching. And when p * = 0.6 and p * = 0.8, DAB operates in the high-power section, the duty cycles of all switching transistors are 50%, and all achieve soft switching. Therefore, their operating states and ZVS performances are consistent with the theoretical analysis.

[0160] Figures 14 to 19 It shows the working waveforms under different modulation strategies at k = 2, p * = 0.2 and p * = 0.7. In addition, their peak-to-peak inductor currents and RMS currents are listed separately. The i Lpp and i Lrms under DVDM are both the lowest, and the RMS current value at p * = 0.7 is relatively close to that under the FDM strategy. When p * = 0.2, all switching transistors under the FDM strategy achieve ZVS. Only Q3 and Q4 under the DVDM strategy do not achieve zero-voltage conduction, and only Q1, Q2, and Q4 under the OADM strategy have soft-switching performance. When p * = 0.7, all switching transistors under the three strategies achieve soft switching. Therefore, in line with the previous theoretical analysis, DVDM has an advantage in current optimization and is superior to the OADM strategy in terms of ZVS characteristics.

[0161] To better reflect the advantages of the DVDM strategy, Figure 20 The DAB efficiencies of the three modulation strategies over the full load range at k = 2 were compared. It can be seen that when k = 2, the transmission efficiencies of DVDM, FDM, and OADM at p *When it is 0.6, they reach the maximum values of 95.48%, 95.24% and 94.07% respectively. Compared with OADM and FDM, the efficiency of DVDM has generally improved. Under light load conditions, the improvement of DVDM compared to FDM is the most obvious. In addition, the efficiency of DVDM is at most 2.12% higher than that of FDM and at most 1.76% higher than that of OADM. In summary, under different voltage transfer ratios, the DVDM strategy can achieve higher efficiency in the entire power range, and the improvement is more obvious under light load.

[0162] In summary, the bilateral variable duty cycle phase shift modulation strategy with three degrees of freedom proposed by the present invention can meet the expected goals. According to the optimized modulation strategy, within the full load range, the peak-to-peak value and RMS current of the inductor current of the DAB converter are effectively reduced, and the ZVS range is broadened. In practical applications, the performance improvement of DAB under light load is the most obvious, the efficiency improvement is the largest, and the peak efficiency of the DVDM strategy can reach 95.48% when the voltage transfer ratio is 2.

[0163] Although the present invention has been described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the present invention. Therefore, it should be understood that many modifications can be made to the exemplary embodiments, and other arrangements can be designed, as long as they do not deviate from the spirit and scope of the present invention as defined by the appended claims. It should be understood that the different dependent claims and the features described herein can be combined in a manner different from that described in the original claims. It should also be understood that the features described in connection with a single embodiment can be used in other described embodiments.

Claims

1. A double-side variable duty cycle phase shift modulation method for a dual active bridge converter, characterized in that include, One arm of the primary full bridge of the dual active bridge converter includes switch tubes Q1 and Q2, and the other arm includes switch tubes Q3 and Q4. The input voltage is U1, and the input filter capacitor is C1. One arm of the secondary full bridge includes switch tubes Q5 and Q6, and the other arm includes switch tubes Q7 and Q8. The output voltage is U2, and the output filter capacitor is C2. The output voltage at the midpoint of the primary full bridge is U ab , the secondary full-bridge midpoint output voltage is U cd The primary full bridge and the secondary full bridge are connected through a high-frequency transformer with a turns ratio of n, and the primary in-phase end of the high-frequency transformer is connected to the inductor L; All switches are modulated with variable duty cycles, and the trigger signals of the switches on the same bridge arm are complementary. The phase shift ratio between switch Q1 and switch Q4 is defined as the inner shift ratio D1, and the phase shift ratio between switch Q1 and switch Q5 is defined as the outer shift ratio D2. The phase shift ratio between switch Q1 and switch Q3 is defined as D0. The duty cycles of switches Q1, Q3, Q5 and Q7 are expressed as D0+D1. According to the unequal relationship between D0, D1 and D2, select mode 1 to mode 3 with duty cycle less than 50% of Q1, Q3, Q5 and Q7 for modulation: Mode 1: D1≤D2≤D0+D1≤0.5, and D0+D1+D2≤1-D0; Mode 2: 0≤D1≤D2, and D0+D1≤0.5; Mode 3: D1≤D2≤D0+D1≤0.5, and 1-D0≤D0+D1+D2; When the transmission power per unit value is [(2k-2) / k 2 ,1] interval, select mode 3; k represents the voltage transfer ratio; When the transmission power per unit value is in [0, (2k-2) / k 2 ], choose to work in mode 1 or mode 2; The modulation methods of the three modes include: The per-unit value expression of the current peak-to-peak value of the inductor L is determined based on D0, D1 and D2, the per-unit value of the current peak-to-peak value is taken as the optimization target, and the peak-to-peak value of the current is optimized in combination with the per-unit value of the transmission power and the constraints of D0, D1 and D2; the optimal solution of D0, D1 and D2 is obtained, and the dual active bridge converter is phase-shifted and modulated based on the optimal solution.

2. The double-sided duty cycle phase shift modulation method of the dual active bridge converter according to claim 1, characterized in that: In mode 1, the calculation method of the transmission power per unit is: In one switching cycle, the inductor current i L (t) is: Where t is time, t0 = 0, t1 = D1T s , t2=D2T s , t3=(D0+D1)T s , t4=(D0+D1+D2)T s , t5=(1-D0)T s , t6=(1-D0-D1+D2)T s , t7 = T s Where T s is the switching cycle; According to the fact that the integral of the inductor current in one switching cycle is 0, we can obtain: Define the voltage transfer ratio k = U1 / nU2, then the current base value i N And the transmission power base value P N for: Where f s is the switching frequency; Then the per-unit value of the inductor current is calculated as: The transmission power per unit value p * for:

3. The double-sided duty cycle phase shift modulation method of the dual active bridge converter according to claim 2, characterized in that: The peak-to-peak value of the current of the inductor L is i Lpp * for: i Lpp * =8[k(D0)+(-D0-D1+2D2)]。 4. The double-sided duty cycle phase shift modulation method of the dual active bridge converter according to claim 3 is characterized in that: Under mode 1, the conditions for all switches to achieve soft switching are:

5. The double-sided duty cycle phase shift modulation method of the dual active bridge converter according to claim 4, characterized in that: The optimization problem is established as follows:

6. The double-sided duty cycle phase shift modulation method of the dual active bridge converter according to claim 5, characterized in that: Construct the Lagrange polar equations and optimization conditions: Where E represents the Lagrange polar equation and λ is the Lagrange multiplier.

7. The double-sided duty cycle phase shift modulation method of the dual active bridge converter according to claim 6, characterized in that: Solve the Lagrange polar equation and get the optimal solution for D0, D1 and D2:

8. The double-sided duty cycle phase shift modulation method of the dual active bridge converter according to claim 7, characterized in that: Transmission power per unit value p in mode 2 * for: p * =8(-D0D1+2D0D2); Transmission power per unit value p in mode 3 * for: p * =4(-1+4D0-4D0 2 +2D1-6D0D1-4D1 2 +2D2+4D1D2-4D2 2 ); The peak-to-peak value of the current of the inductor L in mode 2 is i Lpp * for: The peak-to-peak value of the current of the inductor L in mode 3 is i Lpp * for: i Lpp * =8[k(D0)+(-D0-D1+2D2)]。 9. The double-sided duty cycle phase shift modulation method of the dual active bridge converter according to claim 8, characterized in that: In mode 3, the conditions for all switches to achieve soft switching are:

10. The double-sided duty cycle phase shift modulation method of the dual active bridge converter according to claim 9, characterized in that: In mode 3, the optimal solutions of D0, D1 and D2 are:

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