Charging method of new energy automobile bus pulsating type high-frequency isolation integrated charging system

Abstract

The invention discloses a new energy automobile bus pulsating type high-frequency isolation integrated charging system charging method, based on a new energy automobile bus pulsating type high-frequency isolation integrated charging system, in a charging mode, a post-stage TPFL converter adopts a finite set model prediction control method, good sinusoidal property of a power grid charging current is ensured, and the charging efficiency is improved. The power factor is approximately 1, and the power grid current and the power grid voltage are consistent in phase; a direct current bus side is only connected in parallel with a buffer circuit consisting of a single MOSFET (Metal-Oxide-Semiconductor Field Effect Transistor) and a 10 [mu] F thin-film capacitor, high-frequency pulsation of intermediate bus voltage is realized through cooperation of the buffer circuit and a preceding-stage DAB, and zero-voltage switching of most power devices of a system is realized by utilizing a zero-voltage region of the intermediate bus voltage pulsation. The efficiency, the reliability and the power density of the system are improved.

Description

Technical Field

[0001] The present invention belongs to the technical field of power generation, transformation or distribution, involving electrical engineering, electrical machines, and power electronics technologies, and particularly relates to a charging method for a bus pulsating high-frequency isolation integrated charging system of a new energy vehicle. Background Art

[0002] In recent years, with the consumption of fossil energy and the aggravation of environmental problems, new energy vehicles have received increasing attention in the transportation field. In fact, the motor drive technology and power electronic conversion technology applied to new energy vehicles have become important innovations in the corresponding fields, and the charging technology has also become one of the key technologies for electric vehicles. According to different powers, electric vehicle charging devices can be divided into three types: less than 1.92 kW, greater than 1.92 kW and less than 19.2 kW, and greater than 19.2 kW. The first two charging devices can be installed inside new energy vehicles to become on-board chargers (OBCs). Because no specific charging piles and charging stations are required, on-board chargers have become a research hotspot in new energy vehicle charging technology.

[0003] In order to further improve the power density of on-board chargers, reduce the volume, weight, and cost of chargers, some scholars have proposed an integrated system that includes a motor drive system and an on-board charging system. However, previous research on integrated charging systems mostly based on traditional control methods that require large-value capacitors to be connected in parallel to the DC bus. However, large-value bus capacitors are large in volume and heavy in weight, and have become an important factor restricting the power density and reliability of integrated charging systems. Regarding how to reduce the capacitance value of the bus capacitor or even remove the capacitor on the DC bus, there have been some studies, but there is still a lack of research on the bus pulsating high-efficiency charging method for a high-frequency isolation integrated charging system with a small bus capacitor. Summary of the Invention

[0004] The purpose of the present invention is to provide a charging method for a bus pulsating high-frequency isolation integrated charging system of a new energy vehicle. This charging method is based on the bus pulsating high-frequency isolation integrated charging system of a new energy vehicle. In the charging mode, the latter-stage TPFL converter adopts a finite set model predictive control method, which ensures good sinusoidality of the grid charging current, and the power factor is approximately 1, and the grid current and grid voltage are in phase. Only a buffer circuit composed of a single MOSFET and a 10 μF thin-film capacitor is connected in parallel on the DC bus side. Through the cooperation of the buffer circuit and the front-stage DAB, high-frequency pulsation of the intermediate bus voltage is achieved, and the zero-voltage region of the intermediate bus voltage pulsation is utilized to achieve zero-voltage switching of most of the power devices in the system. Finally, the efficiency, reliability, and power density of the system are improved.

[0005] To achieve the above purpose, the solution of the present invention is:

[0006] A charging method for a bus pulsating high-frequency isolation integrated charging system of a new energy vehicle. The integrated charging system includes a front stage connected to a battery, a buffer circuit, and a rear stage connected to a motor / grid. Among them, the front stage is a dual-active full-bridge converter, and the rear stage is a three-phase four-leg structure; the dual-active full-bridge converter includes a transformer and the first to eighth switching tubes. The first switching tube Q1 and the second switching tube Q2 are sequentially connected to form a first branch, the third switching tube Q3 and the fourth switching tube Q4 are sequentially connected to form a second branch, the fifth switching tube Q5 and the sixth switching tube Q6 are sequentially connected to form a third branch, and the seventh switching tube Q7 and the eighth switching tube Q8 are sequentially connected to form a fourth branch. One end of the primary side of the transformer is connected between the first switching tube and the second switching tube, and the other end of the primary side of the transformer is connected between the third switching tube and the fourth switching tube. One end of the secondary side of the transformer is connected between the fifth switching tube and the sixth switching tube, and the other end of the secondary side of the transformer is connected between the seventh switching tube and the eighth switching tube; after the first branch and the second branch are connected in parallel, they are respectively connected to both ends of the battery; after the third branch and the fourth branch are connected in parallel, they are respectively connected to the buffer circuit. It includes:

[0007] In a single cycle, the first half cycle and the second half cycle are symmetric. Taking half a cycle as an example;

[0008] First, turn off the ninth switching tube in the buffer circuit;

[0009] When the intermediate bus voltage drops to the set threshold, turn on the second switching tube;

[0010] When the intermediate bus voltage drops to 0 and after at least one switching time delay, turn off the fifth switching tube and the eighth switching tube;

[0011] After completely turning off the fifth switching tube and the eighth switching tube, turn on the second switching tube again;

[0012] Before the current flowing through the transformer crosses zero and reverses, turn on the sixth switching tube and the seventh switching tube;

[0013] After the transient commutation process is completed, turn on the ninth switching tube again.

[0014] In the above charging method, the modal analysis of the front stage includes,

[0015] State 0: [Before t0]

[0016] Assume that a capacitor C is connected in parallel on the DC bus of the integrated charging P , and the parasitic capacitance of the IGBT is equivalent to an ideal model;

[0017] State 1: [t0, t1]

[0018] At time t0, the ninth switch tube is turned off, and the DC bus voltage starts to drop. During this process, the battery voltage remains stable. Therefore, the negative voltage across the transformer gradually increases, and the current gradually decreases;

[0019] State 2: [t1, t2]

[0020] At time t1, the second switch tube conducts, and the current flows through the second switch tube, and the DC bus voltage continues to decrease;

[0021] State 3: [t2, t3]

[0022] At time t2, all the energy in the capacitor C p is completely released, and the DC bus voltage also drops to 0, and the DC bus voltage is clamped close to 0V;

[0023] State 4: [t3, t4]

[0024] At time t3, the fifth switch tube and the eighth switch tube are turned off, and the DC bus voltage rises;

[0025] State 5: [t4, t5]

[0026] At time t4, the second switch tube is turned off, and the leakage inductance current charges the battery;

[0027] State 6: [t5, t6]

[0028] At time t5, the sixth switch tube and the seventh switch tube conduct. After decaying to zero, the current continues to flow through the sixth switch tube and the seventh switch tube and increases in the reverse direction;

[0029] State 7: [t6, t7]

[0030] At time t6, the current commutates from the freewheeling diodes of the first switch tube and the fourth switch tube to the freewheeling diodes of the second switch tube and the third switch tube;

[0031] State 8: [t7, t8]

[0032] At time t7, the ninth switch tube conducts. When the current is equal to the load current, the DC bus voltage reaches the maximum value, and then the current continues to increase and the DC bus voltage drops.

[0033] Among them, it also includes the control of the latter stage, including the following steps.

[0034] First, establish a mathematical model of the TPFL rectifier and represent the switching state of the rectifier with binary numbers;

[0035] Then, further derive the prediction model and value function of each phase current;

[0036] The instantaneous value of the grid voltage is sampled, and the frequency and amplitude of the grid voltage are calculated through a second-order phase-locked loop;

[0037] The current flowing through the three-phase winding is sampled;

[0038] The proportionality coefficient is calculated according to the given charging power and the measured peak value of the grid voltage, and the reference value of each phase current is calculated according to the proportionality coefficient and the instantaneous grid voltage;

[0039] According to the motor parameters and the measured three-phase current, the predicted values of the currents of each phase in the next cycle corresponding to all switching modes are obtained;

[0040] The error values corresponding to different switching states are calculated by using the reference values and predicted values of the currents of each phase, and the switching state with the smallest error value is selected as the switching state of the next stage in the next cycle;

[0041] The switching state of the fourth bridge arm of the latter stage is obtained according to the measured instantaneous grid voltage.

[0042] The switching state of the latter-stage circuit is switched at the moment when the DC bus voltage is close to zero voltage, and the switching devices in the latter-stage circuit all meet the requirements of zero-voltage switching.

[0043] In the above-mentioned front-stage circuit, the devices in the middle-bus-side H-bridge all achieve zero-voltage switching, while the third and fourth switching tubes in the battery-side H-bridge are always turned off, and only the first and second switching tubes in the battery-side H-bridge and the ninth switching tube in the buffer circuit do not achieve zero-voltage switching.

[0044] The above charging method further includes,

[0045] Firstly, consider the zero-voltage requirement of the middle-bus voltage, the zero-voltage switching requirement of the front-stage circuit and the current peak limit of the front-stage circuit, then analyze the active, passive devices and parasitic parameters in the circuit, calculate the current and voltage after the action of each switching device, and finally obtain the control strategy of the front-stage circuit by designing the coordinated cooperation between the action of the switching device and the freewheeling of the diode;

[0046] The fifth to eighth switching tubes of the H-bridge on the intermediate bus side conduct complementarily, thereby generating a high-frequency alternating voltage, which is transmitted to the H-bridge on the battery side through a high-frequency transformer; then, the first to fourth switching tubes rectify the voltage; the third and fourth switching tubes in the battery-side bridge are always turned off, and during the entire charging process, the current flows through the freewheeling diodes of the third and fourth switching tubes, reducing the switching losses of the third and fourth switching tubes; the first and second switching tubes conduct briefly alternately according to the polarity of the voltage. When the first switching tube or the second switching tube conducts, all the energy in the parasitic capacitance of the power device is released into the leakage inductance of the transformer, causing the intermediate bus voltage to drop to zero; finally, a zero-voltage region is generated in the intermediate bus voltage, creating conditions for the subsequent stage circuit to achieve zero-voltage switching; before the intermediate bus voltage rises, the first switching tube or the second switching tube is turned off to ensure the smooth commutation of the previous stage circuit.

[0047] After adopting the above scheme, the present invention has the following advantages compared with the prior art:

[0048] (1) The present invention realizes the integration of the new energy vehicle drive system, V2G system and battery charging system, reduces the volume and mass of the system, and improves the power density of the system. The large capacitor connected in parallel on the DC bus side in the traditional method is removed, further improving the power density and reliability of the system.

[0049] (2) In the charging mode, through the coordinated cooperation between the buffer circuit and the previous stage DAB converter, the present invention generates a high-frequency pulsating DC bus voltage, and designs a special control strategy to achieve zero-voltage switching of most of the power devices in the DAB converter, improving the efficiency of the system.

[0050] (3) In the charging mode, the present invention uses the finite set model predictive method to control the subsequent stage TPFL rectifier, making the grid-connected charging current have a high sinusoidality, small harmonic components, and a power factor approximately equal to 1. And by utilizing the characteristic of the finite set model predictive method to fix the switching moment, and cooperating with the high-frequency pulsating bus voltage generated by the DAB converter control strategy, the switching action of the TPFL converter is carried out in the zero-voltage region of the intermediate bus voltage, realizing zero-voltage switching of all the devices in the rectifier and improving the efficiency of the system. Description of the Drawings

[0051] Figure 1 is the topology structure of the new energy vehicle bus pulsating high-frequency isolation integrated charging system;

[0052] Among them, 1.1 is the front-stage dual-active full-bridge DC-DC converter (DAB), 1.2 is the buffer circuit, 1.3 is the subsequent stage three-phase four-leg (TPFL) converter, 1.4 is the stator winding of the motor, and 1.5 is the 10 μF film capacitor in the buffer circuit;

[0053] Figure 2 It is a schematic diagram of the driving signals and voltage pulse waveforms of the DAB converter and the buffer circuit in the charging method of the present invention;

[0054] Among them, 2.1 and 2.2 are the driving signals of Q2 and Q1 respectively, 2.3 is the driving signal of Q3 and Q4, 2.4 is the driving signal of Q6 and Q7, 2.5 is the driving signal of Q5 and Q8, 2.6 is the driving signal of Q s , 2.7 is the voltage between points A and B, and 2.8 is the intermediate bus voltage;

[0055] Figure 3 It is the control flow chart of the DAB converter in the charging method of the present invention;

[0056] Among them, 3.1 is the state of the current in the DAB, 3.2 is the state of the intermediate bus voltage, and 3.3 is the action state of the switching device in the DAB;

[0057] Figure 4 It is some key waveforms of the integrated charger within half a cycle;

[0058] Among them, 4.1 and 4.2 are the driving signals of Q2 and Q1 respectively, 4.3 is the driving signal of Q3 and Q4, 4.4 is the driving signal of Q s , 4.5 is the driving signal of Q5 and Q8, 4.6 is the driving signal of Q6 and Q7, 4.7 is the switching signal of the device Q9 in the subsequent stage TPFL, 4.8 is the current i r flowing through the leakage inductance L of the transformer in the DAB tr , and 4.9 is the intermediate bus voltage v PN ;

[0059] Figure 5 is the current path modal diagram in the charging mode of the charging method of the present invention within half a cycle;

[0060] Figure 6 It is the control block diagram of the subsequent stage TPFL converter in the charging method of the present invention;

[0061] Among them, 6.1 is a second-order phase-locked loop, 6.2 is a proportional coefficient calculator, 6.3 is a model predictive calculation module based on motor parameters and inverter topology, 6.4 is an error function module, and 6.5 is a TPFL converter;

[0062] Figure 7 It is the steady-state experimental result diagram of the charging method of the bus pulsating type high-frequency isolation integrated charging system for new energy vehicles;

[0063] Among them, 7.1 is the input-side grid voltage waveform, 7.2 is the input-side grid-connected charging current waveform, 7.3 is the FFT analysis result of the grid-connected current, 7.4 is the battery voltage, and 7.5 is the battery charging current;

[0064] Figure 8 It is a diagram of the experimental results of the intermediate bus voltage and the driving waveform;

[0065] Among them, 8.1 is the driving voltage waveform of the IGBT device Q2 in the battery-side H-bridge, 8.2 is the driving voltage waveform of the IGBT device Q6 in the DC bus-side H-bridge, 8.3 is the driving voltage waveform of the IGBT device Q5 in the DC bus-side H-bridge, and 8.4 is the waveform of the intermediate bus voltage v PN ;

[0066] Figure 9 It is a diagram of the experimental results of the dynamic performance;

[0067] Among them, 9.1 is the grid-connected charging current waveform before the charging power reference value is adjusted, and 9.2 is the grid-connected charging current waveform after the charging power reference value is adjusted;

[0068] Figure 10 It is the experimental result of the zero-voltage switching of the IGBT power device in the integrated charging system;

[0069] Among them, 10.1 is the DC bus voltage waveform, 10.2 is the driving waveform of Q9 in the TPFL. 10.3 is the waveform of the voltage across Q5 in the DAB, and 10.4 is the driving voltage waveform of Q5;

[0070] Figure 11 It is the waveform of the DAB current i tr ;

[0071] Among them, 11.1 is the waveform of the DAB converter current i tr ; 11.2 is the partial enlarged view of the waveform of the DAB converter current i tr ;

[0072] Figure 12 It is a diagram of the experimental results when the battery voltage is 300V;

[0073] Among them, 12.1 is the waveform of the DAB current i tr ; 12.2 is the intermediate bus voltage with high-frequency pulsation, 12.3 is the grid voltage, and 12.4 is the grid current. Specific implementation manners

[0074] The following further clarifies the inventive concept of the present invention in conjunction with the accompanying drawings and specific embodiments. It should be understood that these embodiments are only used to illustrate the present invention and not to limit the scope of the present invention. After reading the present invention, various equivalent forms of modifications made by those skilled in the art to the present invention all fall within the scope defined by the appended claims of this application.

[0075] The present invention provides a charging method for a bus pulsation type high-frequency isolation integrated charging system of a new energy vehicle, including:

[0076] In a single cycle, the first half cycle and the second half cycle are symmetric. Taking half a cycle as an example. First, turn off Q s , when Q s is turned off, at this time the buffer circuit is disconnected from the intermediate bus. Since the original discharge current of the buffer circuit will be transferred to the equivalent parasitic capacitance loop, the capacitor discharges rapidly, and the equivalent parasitic capacitance C p and the leakage inductance L r generate resonance, and the intermediate bus voltage drops. At this time, turn on Q2 at the lowest point of the intermediate bus voltage. In actual operation, in order to prevent the bus voltage from rising, some redundancy is reserved, and Q2 is turned on before the lowest point.

[0077] At this time, the DC bus voltage is zero, and the current in the DAB does not charge the battery. In order to improve the efficiency, the circulating current time should be minimized as much as possible, and at the same time, the switching time of the devices should be considered to ensure the normal operation of the DAB.

[0078] Therefore, it is necessary to turn off Q5 and Q8 after the intermediate bus voltage has completely dropped to 0 and at least after an interval of the switching time of all the devices used, and at the same time, all the device switches in the subsequent TPFL are carried out simultaneously with Q5 and Q8.

[0079] In order to minimize the circulating current time as much as possible, Q2 can be turned on again after Q5 and Q8 are completely turned off. At this time, the current starts to decrease rapidly. The interval between the actions of the front and rear switching devices is the turn-off time of one device.

[0080] At this time, the current flows through the freewheeling diodes of Q6 and Q7. In order to ensure the normal commutation of the current and the zero-voltage switching of Q6 and Q7, it is necessary to turn on Q6 and Q7 before the current flowing through the transformer in the DAB crosses zero and reverses.

[0081] Finally, turn on Q s again after the transient commutation process is completed. The turn-on moment of Q s needs to meet the requirement of turning on before the current in the buffer circuit decreases from charging to zero and then discharges reversely, so as not to affect the current flow in the buffer circuit.

[0082] As Figure 1 shown, the present invention provides a charging method for a bus pulsating type high-frequency isolation integrated charging system for new energy vehicles. The system includes:

[0083] The front stage of the integrated charging system is a dual-active full-bridge converter (DAB) 1.1, which is connected to the battery;

[0084] The rear stage of the integrated charging system is a three-phase four-leg (TPFL) structure 1.3, which is connected to the motor and the power grid.

[0085] There is no large-capacitance capacitor on the DC bus of the integrated charging system. Instead, a buffer circuit 1.4 is connected in parallel. The buffer circuit consists of a MOSFET high-speed switching device and a 10 μF thin-film capacitor 1.5.

[0086] The integrated charging system is connected to the single-phase power grid using a plug GS. When in the battery charging mode, the neutral point of the three-phase stator winding of the motor is connected to one end of the single-phase power grid, and the midpoint of the fourth bridge arm of the TPFL converter is connected to the other end of the single-phase power grid to achieve grid-connected charging. Theoretically, in the battery charging mode, the currents flowing through the three-phase stator windings are equal in magnitude and the same in direction, and the magnetic fluxes generated by them in the air gap cancel each other out, and the electromagnetic torque is 0. Therefore, the motor remains stationary during the charging process. In the battery charging mode of this integrated charging system, the stator winding of the motor is reused as the filter inductor on the grid side to achieve zero-voltage switching of the power switching device. The present invention mainly focuses on the bus voltage pulsation type charging method of the bus small-capacitance high-frequency isolation type integrated charging system for the invention content.

[0087] In the charging mode, power is transferred from the grid side to the battery side. In order to achieve zero-voltage switching (ZVS) operation while improving efficiency, a charging control strategy for the DAB converter based on high-frequency voltage pulses on the intermediate bus is proposed. At this time, the driving waveform of the DAB converter is as Figure 2 shown, and the control flow chart of the strategy is as Figure 3 shown. First, consider the zero-voltage requirement of the intermediate bus voltage, the ZVS requirement of the DAB converter devices, and the current peak limit of the DAB converter. Then, further analyze the active, passive components and parasitic parameters in the circuit, and calculate the currents and voltages after the operation of each switching device. Finally, by designing the coordination between the operation of the switching devices and the freewheeling of the diodes, the control strategy of the DAB is obtained. The switching tubes (Q5 / Q6 and Q7 / Q8) of the H-bridge on the intermediate bus side conduct complementarily, thereby generating a high-frequency alternating voltage, which is transmitted to the H-bridge on the battery side through the high-frequency transformer. Then, Q1~Q4 rectify the voltage. Q3 and Q4 in the battery-side bridge are always turned off, and the current flows through the freewheeling diodes of Q3 and Q4 during the entire charging process, reducing the switching losses of Q3 and Q4. Q1 and Q2 conduct briefly alternately according to the polarity of the voltage. When Q1 or Q2 conducts, all the energy in the parasitic capacitance of the power device is released into the leakage inductance of the transformer, causing the intermediate bus voltage to drop to zero. Finally, a zero-voltage region is generated in the intermediate bus voltage, creating conditions for the TPFL rectifier to achieve ZVS. Before the intermediate bus voltage rises, turn off Q1 or Q2 to ensure the smooth commutation of the DAB current.

[0088] To further elaborate on the control method of the DAB, the modal analysis method is used to analyze each state during the operation of the DAB. At this time, the driving waveforms and voltage and current waveforms of the power devices are asFigure 4 As shown, the circuit diagrams of each state are shown in Figure 5. The control method of the front-stage DAB circuit in the charging method of the bus pulsating high-frequency isolation integrated charging system for new energy vehicles includes the following steps:

[0089] In a switching period, the first half period and the second half period are symmetric and have the same mode. Therefore, only the mode of the first half period is analyzed here. To simplify the analysis, the following assumptions are made:

[0090] · The parasitic capacitances of all IGBTs are the same and much smaller than the capacitance of the active buffer circuit;

[0091] · The conduction voltage drop and switching process of the IGBT are ignored;

[0092] · Considering that the inductance value of the motor winding is large and the switching period is short, it is assumed that the load current remains unchanged within a switching period, and the current on each motor winding can be regarded as a constant current source;

[0093] · The winding of the transformer is set to 1:1, and no winding reduction is required;

[0094] State 0: [Before t0]

[0095] When performing modal analysis, an inductor C is connected in parallel to the DC bus of the proposed integrated system P , which does not exist in the actual circuit. This is to equivalently model the parasitic capacitance of the IGBT as an ideal model for easier analysis. Through this ideal model, the pulsation of the bus voltage and the commutation of the leakage inductance current can be analyzed more clearly

[0096] State 1: [t0, t1]

[0097] At time t0, Q s is turned off. Since the current i r flowing through the transformer leakage inductance L tr and the load current I load cannot change suddenly, the current i s originally existing in the buffer circuit is transferred to C p . The discharge current of C p reaches the sum of i s and i s before Q p is turned off. Since C p is very small and difficult to clamp the DC bus voltage, the DC bus voltage begins to drop. During this process, the battery voltage remains stable. Therefore, the negative voltage across L r gradually increases, and the current i tr gradually decreases.

[0098] State 2: [t1, t2]

[0099] At time t1, Q2 conducts, and at this time, L r resonates with the parasitic capacitances of Q1 and Q2, ultimately causing the voltage v ab to drop rapidly. Then the freewheeling diode of Q1 turns off, and the current flows through Q2. At this time, the voltage across L r is approximately equal to the DC bus voltage, so the current i tr increases. Since I load remains unchanged, the current i p increases as i tr increases, the discharging speed accelerates, and the DC bus voltage continues to decrease.

[0100] State 3: [t2, t3]

[0101] At time t2, all the energy in the capacitor C p is completely released, and the DC bus voltage also drops to 0. At this time, the value of the leakage inductance current i tr is greater than the load current, and a part of the leakage inductance current circulates in the middle bus side H-bridge through the freewheeling diodes of the two switching tubes Q6 and Q7. Therefore, the DC bus voltage v PN is clamped at nearly 0V. Since the voltage across L r is very small, it can be considered that i tr remains unchanged during this period

[0102] State 4: [t3, t4]

[0103] At time t3, Q5 and Q8 are turned off. At this time, since the DC bus voltage v PN is clamped at 0, both Q5 and Q8 are turned off with zero voltage. After Q5 and Q8 are turned off, the current i tr can only conduct through the freewheeling diodes of Q6 and Q7. In addition, the switching tubes in the three-phase four-leg converter also act at time t3 to achieve zero-voltage switching, and the load current I load also switches to I load2 due to the action of the three-phase four-leg converter. Then the current i tr and the load current I load2 charge the capacitor C p simultaneously, so the DC bus voltage rises rapidly. In addition, since the voltage across L r is equal to the negative value of the DC bus voltage, the current i tr starts to decrease.

[0104] State 5: [t4, t5]

[0105] At time t4, Q2 is turned off, and the leakage inductance current flows through the freewheeling diode of Q1 to the battery to charge the battery. At this time, L rThe negative pressure at both ends is the sum of the DC bus voltage and the battery voltage, and the leakage inductance current decays rapidly. When the DC bus voltage v PN rises above the voltage across the buffer circuit capacitor C s , the freewheeling diode of Q s conducts. Then the currents i tr and I load will charge the capacitors C p and C s simultaneously. Considering that this period is very short and the capacitance value of C s is much larger than that of C p , it can be considered that the DC bus voltage is clamped.

[0106] State 6: [t5, t6]

[0107] At time t5, Q6 and Q7 conduct. Q6 and Q7 need to conduct before the leakage inductance current i tr decays to 0 to achieve zero-voltage switching. After decaying to zero, the current continues to flow through Q6 and Q7 and increases in the reverse direction. However, due to the reverse recovery effect of the diodes, the current in the battery-side H-bridge still conducts through the freewheeling diodes of Q1 and Q4.

[0108] State 7: [t6, t7]

[0109] At time t6, the current i tr commutates from the freewheeling diodes of Q1 and Q4 to the freewheeling diodes of Q2 and Q3. At this time, the voltage across the leakage inductance L r changes from (v PN / k + V BAT ) to (v PN / k - V BAT ), where V BAT is the voltage across the battery, v PN is the DC bus voltage, and k is the turns ratio of the high-frequency transformer, which is 1 here. The change in the voltage across the leakage inductance L r results in a change in the rate of change of the current i tr . During this period, the value of the current i tr is always less than the load current I load , so C p and C s continue to charge, the DC bus voltage is slightly higher than the battery voltage, and i tr continues to increase.

[0110] State 8: [t7, t8]

[0111] At time t7, Q s conducts. Q s needs to conduct when the value of the current i tr rises to the same as the load current Iload2 Conduction before equality. When the current i tr is equal to the load current I load , at this time the DC bus voltage reaches the maximum value, and then the current i tr continues to increase, and the DC bus voltage v PN decreases. It should be noted that due to the clamping effect of the capacitor C s , with respect to the base value of 350V of the DC bus, the voltage change of v PN in states 6, 7, and 8 is very small.

[0112] In order to make full use of the zero-voltage region of the pulsating bus voltage to achieve zero-voltage switching of the devices in the TPFL rectifier, a control method for the TPFL rectifier is designed based on the idea of model predictive control, which not only realizes the zero-voltage switching of the switching devices but also effectively controls the charging power and charging current on the grid side.

[0113] As Figure 6 shown, the model predictive control method adopted by the control module of the subsequent-stage TPFL includes the following steps:

[0114] 1) First, establish the mathematical model of the TPFL rectifier and represent the switching state of the rectifier with binary numbers. For a two-level voltage source inverter, the TPFL rectifier has a total of 2 to the 4th power of possible states, that is, 16 kinds. Use S i to represent the voltage state of the i-th bridge arm: S i =1 means that the upper half of the bridge arm of this bridge arm is conducting, and at this time the output voltage of this bridge arm is positive; S i =0 means that the upper and lower halves of the bridge arm of this bridge arm are conducting, and at this time the output voltage of this bridge arm is negative. Therefore, the output voltage state of the TPFL rectifier can be represented by (1). The voltage between the neutral point and the negative end of the intermediate link can be calculated by (2). Then, the dynamic equations of each phase current are shown in (3). R s and L ls are the resistance and leakage inductance of the motor stator winding.

[0115] [v u ,v v ,v w ,v N =[S1,S2,S3,S4]·V pulse (1)

[0116] v n =v N +v g =S4V pulse +v g (2)

[0117]

[0118] 2) Then, further derivation is carried out to obtain the prediction models and value functions of the phase currents of each phase. When the currents flowing through the three-phase windings are controlled to be equal, the torques generated by the three-phase windings will cancel each other out in the air gap. Therefore, the rotor of the motor can remain stationary. Substituting Equation (2) into Equation (1), the expressions of the voltages of each phase in Equation (4) are obtained. To achieve digital control, Equation (3) is discretized into the form in Equation (5). i u (k), i v (k), i w (k) and v g are the sampled values of the three-phase stator currents and gate voltages at time k, respectively. Among them, i u (k + 1), i v (k + 1), i w (k + 1) are the predicted values of the three-phase currents at time k + 1. In the charging mode, the goal of the TPFL rectifier is to obtain a single-phase sinusoidal charging current synchronized with the grid voltage. In addition, to ensure smooth charging and the stationary state of the motor, it is also necessary to consider the uniform sharing of the charging currents of the three-phase bridge arms of the TPFL on the grid side. Therefore, the value function is configured in (6), where λ1 and λ2 are the weight factors of the total grid current and the phase currents, respectively. As a multi-objective optimization method, the weight factor is often used to adjust the weights of different objectives in the value function. The value of the weight factor can be adjusted according to the needs of current control and current sharing. i g *, i u *, i v * and i w * are the reference values of the total grid current and the phase currents. i u *, i v * and i w * are used to control the phase currents in the rectifier, and their values are all one-third of i g *.

[0119] Therefore, λ1 and λ2 can be designed according to the current amplitude of the control objective.

[0120]

[0121] 3) The voltage sampling module samples the instantaneous value of the grid voltage and calculates the frequency and amplitude of the grid voltage through a second-order phase-locked loop. The input of the phase-locked loop is the grid instantaneous voltage v g and the rated frequency w g , and the output is the fundamental component v1 and the quadrature component v2 of the grid voltage. By obtaining the fundamental component v1 and the quadrature component v2 of the grid voltage, the peak value and instantaneous phase of the grid voltage can be obtained according to Equation (7);

[0122]

[0123] 4) The current sampling module samples the currents \(i_a\), \(i_b\), and \(i_c\) flowing through the three-phase windings; u , \(i_b\), v and \(i_c\); w ;

[0124] 5) The proportional factor calculation module calculates the proportional coefficient \(k\) according to the given charging power \(P\) g * and the measured peak value \(V\) of the grid voltage. According to the proportional coefficient \(k\) g and the instantaneous grid voltage \(v\), the reference values \(i_a^*\), \(i_b^*\), \(i_c^*\) and \(i_d^*\) of each phase current can be calculated, and the calculation formula is as shown in Equation (8). g and the instantaneous grid voltage \(v\), the reference values \(i_a^*\), g \(i_b^*\), g \(i_c^*\) and \(i_d^*\) of each phase current can be calculated, and the calculation formula is as shown in Equation (8). u \(i_b^*\), v \(i_c^*\), w and \(i_d^*\) of each phase current can be calculated, and the calculation formula is as shown in Equation (8). g * and \(i_d^*\) of each phase current can be calculated, and the calculation formula is as shown in Equation (8).

[0125]

[0126] 6) The prediction model established by the finite set model prediction method calculates the predicted values of the currents of each phase in the next cycle corresponding to all 16 switching modes according to the motor parameters and the measured three-phase currents.

[0127] 7) In the error function module, through the set error function, as shown in Equation (6), the error values corresponding to different switching states are calculated using the reference values and predicted values of the currents of each phase. Finally, the switching state with the smallest error value is selected as the switching state of the TPFL converter in the next cycle;

[0128] 8) The switching state of the fourth leg of the TPFL is obtained according to the measured instantaneous grid voltage.

[0129] In order to realize the zero-voltage switching of the switching devices in the subsequent TPFL converter, the TPFL converter can only switch the switching state at a moment close to zero voltage of the DC bus. The finite set model prediction method adopted by the control module of the subsequent TPFL in the charging method of the new energy vehicle bus pulsating high-frequency isolation integrated charging system will result in a variable switching frequency, but this is due to the fact that the switching state of its devices is determined by the cost function. The possible action moment of each device is fixed, that is, at the beginning stage of each sampling period, at this time the value of the intermediate bus voltage is close to zero, so the switching devices in the subsequent TPFL converter all meet the requirements of zero-voltage switching.

[0130] In the front-stage DAB converter, the devices in the H-bridge on the intermediate bus side all achieve zero-voltage switching, while the devices Q3 and Q4 in the H-bridge on the battery side are always off. Therefore, only the devices Q1 and Q2 in the H-bridge on the battery side, and the switching device Q sZero voltage is not achieved. Q s When it operates, the voltage difference across its two ends is small, and the switching loss is small. Therefore, the overall efficiency of the system is effectively improved.

[0131] The steady-state performance experimental results of the charging method of the bus pulsating type high-frequency isolation integrated charging system for new energy vehicles are as Figure 7 and Figure 8 shown. The grid-connected charging current (7.2) has a high sinusoidality, a small total harmonic component, and the THD value is only 3.16% (7.3). And the grid current and the grid voltage (7.1) are in the same phase, and the power factor is approximately 1. The DC bus voltage pulsates at a high frequency, and the pulsation frequency is 40 kHz.

[0132] The dynamic performance experimental results of the charging method of the bus pulsating type high-frequency isolation integrated charging system for new energy vehicles are as Figure 9 shown. When the charging power reference value changes, the grid-connected charging current smoothly and quickly changes to a new steady-state value, demonstrating the good dynamic performance of this method.

[0133] The experimental results of zero-voltage switching of the charging method of the bus pulsating type high-frequency isolation integrated charging system for new energy vehicles are as Figure 10 shown, where 10.1 is the DC bus voltage waveform, 10.2 is the driving waveform of Q9 in the TPFL, and Q9 represents the latter-stage TPFL converter. 10.3 is the waveform of the voltage across Q5 in the DAB, and 10.4 is the driving voltage waveform of Q5, and Q5 represents the former-stage DAB converter. Analysis Figure 10 shows that these power devices have all achieved soft switching.

[0134] The current waveform of the DAB converter in the charging method of the bus pulsating type high-frequency isolation integrated charging system for new energy vehicles is as Figure 11 shown. Analysis shows that the experimentally obtained current i tr and Figure 4 are in full agreement with the analysis in, and the proposed DAB control method has been effectively verified.

[0135] The above experimental results are all based on a battery voltage of 350V. In order to further prove that the charging method disclosed in the present invention does not depend on a certain specific battery voltage, but can be applicable to different battery voltages, experiments were carried out under the condition of a battery voltage of 300V, and the obtained experimental results are as Figure 12 shown. Analysis shows that under different battery voltages, the charging method of the bus pulsating type high-frequency isolation integrated charging system for new energy vehicles can still operate normally.

[0136] The present invention uses a buffer circuit and a small capacitor to replace the large-value electrolytic capacitor connected in parallel on the DC bus side in the traditional method, further improving the power density and reliability of the system. For the charging mode, the present invention designs a special DAB control method. By utilizing the parameter characteristics of the small capacitor of the buffer circuit, the parasitic capacitance of the IGBT, and the leakage inductance of the transformer, and combining the coordinated operation of the switching devices in the DAB and the buffer circuit, a high-frequency pulsating bus voltage is generated on the intermediate bus to facilitate the zero-voltage switching of the three-phase four-leg rectifier. At the same time, zero-voltage switching of most of the devices in the DAB is also achieved. The present invention uses the method of finite set model prediction to control the TPFL converter. By utilizing the characteristic of fixed switching times of model predictive control and combining with the high-frequency pulsating bus voltage, zero-voltage switching of all the devices in the TPFL converter is achieved. Meanwhile, it is ensured that the grid-connected charging current is in phase with the grid voltage, the power factor is basically 1, the current waveform has a high sinusoidality, small harmonic components, and good power supply power quality. The present invention effectively improves the system efficiency while removing the large electrolytic capacitor on the intermediate bus side. Therefore, the charging method of the bus pulsating type high-frequency isolation integrated charging system for new energy vehicles disclosed in the present invention helps to reduce the volume and mass of the electric vehicle system, reduce the system loss, and improve its power density and reliability.

[0137] Although the preferred embodiments of the present invention have been described, additional changes and modifications can be made by those skilled in the art once they learn the basic creative concept. Therefore, the appended claims are intended to be construed to include the preferred embodiments as well as all changes and modifications falling within the scope of the present invention.

[0138] Obviously, those skilled in the art can make various changes and modifications to the present invention without departing from the spirit and scope of the present invention. Thus, if these modifications and variations of the present invention fall within the scope of the claims of the present invention and their equivalent technologies, the present invention is also intended to include these modifications and variations.

Claims

1. A charging method for a bus pulse type high frequency isolated integrated charging system for a new energy vehicle, the integrated charging system comprising a front stage connected to a battery, a buffer circuit and a rear stage connected to a motor / grid, wherein: The front stage is a dual-active full-bridge converter, and the rear stage is a three-phase four-bridge arm structure; the characteristics include: The first half cycle and the second half cycle in a single cycle are symmetrical, taking half a cycle as an example; First, turn off the ninth switch tube in the buffer circuit; When the intermediate bus voltage drops to a set threshold, the second switch tube is turned on; The intermediate bus voltage drops to 0, and the fifth switch tube and the eighth switch tube are turned off after a delay of at least one switching time; After completely turning off the fifth switch tube and the eighth switch tube, turning on the second switch tube again; Turning on the sixth switch tube and the seventh switch tube before the current flowing through the transformer crosses zero and reverses; After the transient commutation process is completed, the ninth switch tube is turned on again.

2. The method according to claim 1, characterized in that: In the charging method, the modal analysis of the front stage includes: State 0: [before t0] Assume that a capacitor C is connected in parallel to the DC bus of the integrated charger. P , the parasitic capacitance of IGBT is equivalent to an ideal model; State 1: [t0, t1] At time t0, the ninth switch is turned off, and the DC bus voltage begins to drop. During this process, the battery voltage remains stable, so the negative pressure at both ends of the transformer gradually increases and the current gradually decreases. State 2: [t1, t2] At time t1, the second switch tube is turned on, the current flows through the second switch tube, and the DC bus voltage continues to decrease; State 3: [t2, t3] At time t2, the capacitor C p The energy in the DC bus is completely released, and the DC bus voltage drops to 0, and the DC bus voltage is clamped at close to 0V; State 4: [t3, t4] At time t3, the fifth switch tube and the eighth switch tube are turned off, and the DC bus voltage recovers; State 5: [t4, t5] At time t4, the second switch is turned off, and the leakage inductor current charges the battery; State 6: [t5, t6] At time t5, the sixth switch tube and the seventh switch tube are turned on, and after decaying to zero, the current continues to flow through the sixth switch tube and the seventh switch tube and increases in the reverse direction; State 7: [t6, t7] At time t6, the current is commutated from the freewheeling diodes of the first switch tube and the fourth switch tube to the freewheeling diodes of the second switch tube and the third switch tube; State 8: [t7, t8] At time t7, the ninth switch tube is turned on, and when the current is equal to the load current, the DC bus voltage reaches the maximum value, and then the current continues to increase and the DC bus voltage decreases.

3. The method according to claim 1, characterized in that: It also includes the control of the subsequent stage, including the following steps: Firstly, the mathematical model of TPFL rectifier is established, and the switching state of the rectifier is represented by binary numbers; Then, the prediction model and value function of each phase current are further derived; The instantaneous value of the grid voltage is obtained by sampling, and the frequency and amplitude of the grid voltage are calculated through a second-order phase-locked loop; The current flowing through the three-phase windings is sampled; A proportionality coefficient is calculated according to a given charging power and a measured grid voltage peak value, and a reference value of each phase current is calculated according to the proportionality coefficient and the grid instantaneous voltage; According to the motor parameters and the measured three-phase currents, the predicted values ​​of the currents of each phase in the next cycle corresponding to all switching modes are obtained; The error values ​​corresponding to different switch states are calculated using the reference value and predicted value of each phase current, and the switch state with the smallest error value is selected as the switch state of the next cycle of the subsequent stage; The switch state of the fourth bridge arm of the rear stage is obtained according to the measured instantaneous voltage of the power grid.

4. The method according to claim 3, characterized in that: The latter circuit switches the switch state when the DC bus is close to zero voltage, and the switch devices in the latter circuit all meet the requirements of zero voltage switching.

5. The method according to claim 1, characterized in that: In the front-stage circuit, the devices in the H-bridge on the intermediate bus side all achieve zero voltage switching, while the third switch tube and the fourth switch tube of the devices in the H-bridge on the battery side are always turned off. Only the first switch tube and the second switch tube of the devices in the H-bridge on the battery side and the ninth switch tube in the buffer circuit do not achieve zero voltage.

6. The method according to claim 1, characterized in that: The charging method further comprises: Firstly, the zero voltage requirement of the intermediate bus voltage, the zero voltage switch requirement of the front-stage circuit and the current peak limit of the front-stage circuit are considered. Then, the active and passive devices and parasitic parameters in the circuit are analyzed, and the current and voltage after each switch device is actuated are calculated. Finally, the control strategy of the front-stage circuit is obtained by designing the coordination between the switch device action and the diode freewheeling. The fifth to eighth switch tubes of the H-bridge on the intermediate bus side are complementary turned on, thereby generating a high-frequency AC voltage, which is transmitted to the H-bridge on the battery side through a high-frequency transformer; then, the first to fourth switch tubes rectify the voltage; the third and fourth switch tubes in the battery side bridge are always turned off, and the current flows through the freewheeling diodes of the third and fourth switch tubes during the entire charging process, thereby reducing the switching losses of the third and fourth switch tubes; The first and second switching tubes are alternately turned on briefly according to the polarity of the voltage. When the first switching tube or the second switching tube is turned on, the energy in the parasitic capacitance of the power device is completely released into the leakage inductance of the transformer, causing the intermediate bus voltage to drop to zero; ultimately, a zero voltage zone is generated in the intermediate bus voltage, creating conditions for achieving zero voltage switching for the subsequent circuit; before the intermediate bus voltage rises, the first switching tube or the second switching tube is turned off to ensure smooth commutation of the previous circuit.

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